Ivan Madjarov
email: ivan.madjarov@lsis.org

An Adaptive Approach to Designing Responsive m-Learning

Keywords: Responsive m-Learning Design, Responsive Web design, Web services, Content adaptation

The ubiquitous availability of a wide range of Internet connected mobile devices makes possible to dedicate small amounts of spare time for mobile learning. The limited screen size makes current mobile devices quite hard to visualize multifaceted web pages, so this kind of content could be adjusted to meet the device needs. CMS-based websites, in general, are designed for large screens and when users go online with mobile phones, smartphones and tablets they feel some discomfort. The usability and user experience may be reduced when interacting with learning platforms that are not optimized for such features. A possible solution has arisen with recent advances in Responsive Web Design specifications. This adaptive web design technique advocates a single self-adaptive interface and avoids thereby multiplication of information sources and technical supports and promotes the possibility to design adaptive e-Learning courses and adaptive e-Learning mobile applications. In this paper we discuss about the possibilities of Responsive Web Design technique to adapt a learning content compared to an adaptive service-based approach to design responsive learning contents for m-Learning usage.

INTRODUCTION

With the advent of connected phones, smartphones and tablets web users have gradually changed behavior. Website developers in majority have reacted by duplicating the content to make one version suitable for the small screen of mobile units and another standardized version for wide screens of desktops and laptops. It is obvious that this approach is impractical and does not really fit the needs of mobile web browsers users. On the other hand, web browsers in mobile devices typically render a given web page without taking into account the difference in the user's visual requirements [START_REF] Sunkara | Responsive, adaptive and user personalized rendering on mobile browsers[END_REF].

The Responsive Web Design (RWD) is a solution for content adaptation on different connected mobiles units [START_REF] Frain | Responsive Web Designe avec HTML5 et CSS3[END_REF]. RWD essentially means that the content presented on a device will adjust to fit the screen size it detects [START_REF] Marcotte | Responsive Web Design, A Book Apart[END_REF]. This new technology is changing the way to design a website. User interfaces are no longer fixed and they are adapted automatically to any medium (desktops, laptops, smartphones, tablets, TV, etc.) without any zooming. Zooming, in particular, significantly accentuates the flawed ergonomics sensation especially on touch devices.

Our motivation for this paper is inspired by two factors: (1) Our university curriculum in computer network and telecommunication offers students modules that handle web development problems, design and administration. In these modules the competencies for RWD implementing are directly targeted. (2) The pedagogical policy of our university is to provide in the near future more than 30% of online courses (MOOC). This has an impact on our courses, not only about the content, but also for the form if one wishes to facilitate the "mobile" access to our pedagogy and encourage students to enjoy at their leisure, without limits, even moving. Today, students are rarely limited on desktop stations because they often revise in transit or elsewhere using their smartphone or tablet. It is therefore crucial that RWD and e-Learning become interrelated so that students can learn their lessons in the most ergonomic way.

Following tests, carried out as part of tutored project, we found that the implementation of RWD in Moodle or Dokeos Learning Content Management Systems (LCMS) for course authors is a difficult task and vary from one system version to another. So, authors are pushed towards a noncommercial and practical solution that is to manually tag the web page and the style file or to use an HTML-compatible editor that better respects the break-points defined by CSS.

In this paper, we discuss and compare some solutions. The paper is organized as follows: first in section 2 we present the state of the art for a responsive e-Learning and related work. In section 3 we present a use case scenario for responsive e-Learning design and a services-based approach. Finally, in section 4 we conclude with some comments about future developments related to this work.

BACKGROUND AND MOTIVATION

An obstacle to open and ubiquitous learning is that mobile devices may have several different hardware and software features. The usability and the user experience may be reduced when interacting with learning platforms that are not optimized for such features. This heterogeneity of resources and devices makes the design of e-Learning content a relevant issue. So, approaches to design websites and applications that are responsive and adaptive to different kinds of devices in different contexts becomes an important task.

Solutions based on applications for mobile devices (native applications) exploit to the full the device's features, but they are platform specific and must be developed, updated and managed separately.

Web-based solutions, on the contrary, follow the principles of open Web and run in standardscompatible web browser. As a consequence, being not developed for a specific device, web-based solutions are typically less powerful and with a poorer look and feel when accessed through a mobile device. An answer to this problem has been the use of techniques borrowed from adaptive hypermedia systems. Device feature include browser version and platform, availability of plug-ins, band-width, display size, input size. Adaptive techniques are an effective solution but their design and implementation is costly and time consuming.

An alternative solution is RWD. This paradigm exploits new specifications for web technologies in particular HTML5 and CSS3 to deploy websites that respond to the user's media environment. RWD enables web applications to have a better look and feel and enables user interfaces to dynamically suite the user device without the implementation of specialized adaptive modules.

Related Work

In the field of adaptive web design we can enumerate three categories of approaches [START_REF] Koehl | Site: Efficient Content Adaptation for Mobile Devices[END_REF]:

 Platform-specific mobile design includes mobile mark-up languages (XHTML Mobile Profile; WML) and various graphical authoring tools to facilitate the creation of a mobile website. The platform-specific mobile design needs mobile designers to invest a significant amount of manual efforts on implementing a mobile website, which is especially optimized to fit the distinct features of mobile devices.  Web page restructuring supports browsing a desktop webpage on a mobile device by adapting the original layout. It first segments a desktop webpage into semantically related information blocks (i.e., page segmentation) and then provides a semantics-directed adaptation (Madjarov, 2012).  Zooming based interaction avoids adaptation, and provides an alternative solution to support browsing desktop webpages on mobile devices. It displays a desktop webpage within a mobile screen as an overview, and provides a mobilefriendly navigation facility to efficiently zoom in on any portion for a detailed reading. In [START_REF] Koehl | Site: Efficient Content Adaptation for Mobile Devices[END_REF] a complex work is presented. A site administrator visually selects objects within a web page, and assigns one or more attributes to page objects from a rich collection of predefined page modifications. The proposed system then generates code for a multi-session, php-based proxy server to provide dynamic mobile content adaptations based on the selected attributes. The proposed adaptation approach cannot be computed on demand or in an independent manner.

Responsive m-Learning

For an adaptive m-Learning it is important to have visuals that communicate effectively on all display sizes and so to deliver user-friendly learning content on multiple devices. To achieve this goal, as mentioned above, we can proceed either by developing a different version for every target device and browser or by the implementation of the RWD technique for our courses. To apply this technique to responsive e-Learning (m-Learning) we can define three key technical features (Baturay, 2013):

Screen Resolutions with CSS3 Media Queries

Flexibility is the key factor to take in consideration when designing responsive e-Learning courses so that the learning content can be restructured and adjusted to fit different screen sizes and resolutions. Media queries, as part of CSS3 recommended standard, return information on devices physical specifications (width, height, orientation, resolution, etc.). So, different styles can be delivered to different devices in order to offer the best visual experience to each user. With Media Queries we can build multiple layouts and direct the fluid grid and flexible images to be aligned to the device dimensions.

A best practice is to define breakpoints for each layout to cover the largest, medium and smallest screen sizes e.g. a desktop, a portrait tablet, and a portrait smartphone. A breakpoint defines the size at which the style rules change and the elements in the layouts are dynamically resized and readjusted to fit screen sizes. As suggested in (Baturay, 2013) and shown in Figure 1, we can base our responsive design on three breakpoints, e.g. 1024px for largest screen, 768px for medium screen size and 320px for smallest screen size.

Fluid Grid Layouts

A traditional e-Learning course design consists of texts, tables, frames and images. For each element, stored in e-Learning system, is assigned a fixed or absolute value in pixels, which specifies exactly his size when displayed on the required web page. With a fluid grid design, the size of each cell in the grid and its contents is defined in relative terms, in proportion to its container. Containers, fonts and images are assigned relative rather than absolute values, expressed as percentages or ems.

The "em" is defined as a size relative to its parent unit, i.e. the web browser. So, we can set 1em to be equal to the browser's default font size which in general is 16px [START_REF] Marcotte | Responsive Web Design, A Book Apart[END_REF]. From there on, we can structure learning content and set relative font sizes in "ems" for texts in different layouts.

The font type is another not insignificant detail. Serif fonts are more difficult to read on digital devices, so it is recommended to select a sans serif font [START_REF] Rasmussen | Learner Engagement: Tips for Responsive Design[END_REF].

Flexible and Scalable Images and Media

Most of e-Learning courses include either raster images or vector graphics or animations. Image flexibility is another important point to consider in the responsive context. This point allows images to be moved and be scaled proportionally on the screen size. Scaling or cropping eliminates the need to upload multiple versions (resolutions) of the same image on the server. Scaling operation resizes image to screen size. For this it is recommended to use bigger images and dynamically scale them down based on information about device size. Scaling smaller image up may lead to quality loss.

Cropping operation means using a larger image and then reducing its size by cutting the image around a focus area that holds the essential area. This operation is preferred when images are scaled to a very small size and their meaning gets lost because one no longer sees the details.

A combination of these operations is always possible to dynamically crop and then scale image to device size.

A flexible solution that bypasses the problem of screen resolution is to use or to convert animations and schemas to Scalable Vector Graphics (SVG). The SVG format is XML based and can be manipulated and styled just like any other HTML element. This format is natively recognized by most Web browsers, (for details see Figure 2).

Responsive e-Learning Design Limitations

The RWD approach (1) is not presented as a panacea and cannot settle all issues related to a complex pedagogical content, (2) it is not straightforward to adapt an existing course, (3) a new course requires careful preparation, (4) there are few technical things to consider when implement:

 Not all mobile browsers and devices support all CSS3 features. For some former models pronounced incompatibilities are observed.  The responsive technique aims to resize content for any device, but in practice there will always be some mobile devices with unusual resolutions.  The responsive technique aims also to resize bitmap image. This process may impact the mobile unit's performance because the full image is downloaded on a user's device and then scaled and cropped to fit the screen.  An e-Learning course that presents multiple multimedia concepts is less easy to adapt for scaling down.

SERVICES-BASED APPROACH FOR RESPONSIVE E-LEARNING DESIGN

Use Case Scenario

In content adaptation, one challenge is to group semantically related information since the HTML specification does not reveal an information organization (Roudaki, 2015). To cover all device capabilities and to lead an efficient course adaptation process we advocate for a semantically clear format for learning content repository and content mining.

To deliver a single version of a course to multiple devices we adopt service oriented approach. So, we can show different content on different devices and we can adapt content depending on the context, e.g. provide learners a highlighted summary of a larger course on mobile devices as a refresher. The full content is downloaded if requested in adapted format. We favor the scroll instead of single page style by learning activity. This technique is more suited to mobile units without zooming or horizontal scrolling.

In case we have some complex multimedia elements on a desktop version, we may decide depends on the mobiles characteristics to substitute this content with an alternative text or audio description. Because of these difficulties and limits, we think that in certain cases making the entire course available on mobile devices may not be the best approach. The main idea here is to make e-Learning content independent of his presentation by adopting a unified XML based format for learning content storage, editing and searching. This approach greatly facilitates content adaptation, (data) extraction and content (data) integration in a composite course on demand (mining).

Mobile Browsers Responsive Compatibility

A study conducted in late 2013 shows that 50% of mobile phone users use mobile as their primary Internet source (Roudaki, 2015). However, the development of mobile websites and especially for LCMS lags behind mobile devices.

We have created courses for some informatics modules using authoring tools accessible in LCMS Moodle and Dokeos and we have had multiple requests from students for options that they could run on Smartphones to fit the content on the screen size to obtain better legibility.

To proof the content portability for mobile browsers, we have processed by several tests. We put a representative selection of mobile browsers through a series of web pages under mentored projects. For this experience a common type of web pages were designed by students, not especially to work on small screens. The results of conducted tests are shown in Figure 2. The results analysis confirms that a multimedia pedagogical content, which is especially appreciated by students, is suitable on mobile browsers. The essential problem to manage here does not focus on the complexity of pedagogical hypermedia content, but its responsive presentation on smallscreen devices. This process include the development of a suitable page-adaptation technique that analyzes course structure and generates pages into logically related units to be fitted to mobile device browser (Madjarov, 2012).

MathML is part of HTML5 standard and it is supported natively or via plug-ins by the majority of modern browsers. A scientific content can serve in standards-compliant HTML pages some MathML equations. But this is not necessarily true for the mobile versions of these browsers. Our tests, conducted with some basic MathML elements, prove that only the mobile version of Firefox web browser provides complete satisfaction. In Figure 3 and Figure 4 the MathML script is natively interpreted by the Firefox browser mobile version. Other tested browsers required for the same task additional resources as shown in Figure 2 for each tested mobile browser.

Tests (Figure 3 and Figure 4) also show the natively compatibility of SVG scripts for vector graphics visualization for all tested browsers. This point is important and may replace in certain cases the unwieldy raster graphics. The SVG script is XMLbased and fully integrated in the web page in the same manner as MathML.

In this HTML5 collection, we can insert a bitmap image or any other binary data using Base64 encoding method to produce an ASCII sequence natively interpreted by latest versions of mobile browsers. So, we obtain a smart script for a responsive pedagogical content compatible with any existing LCMS. The raster image of Pythagoras in Figure 3 is Base64 encoded in Figure 4.

The latest W3C-HTML5 recommendation remains semantically poor and limits the possibility for a learning content adaptation on the fly. The adaptation process for a mobile unit is associated with complex algorithms and techniques for splitting, restructuring, extraction, and summarizing the learning content (Madjarov, 2012). Taking into account the previously mentioned limits of RWD this stage can be simplified if we use a semantically clear and adapted course structure as presented in Figure 5.

The course semantics schema shows common elements of a scientific learning content (Fig. 5). A science course contains chapters, sections, text paragraphs, bitmap images, scalable vector graphics, and mathematical equations, programming code, charts and tables. This XML schema defines the course grammar for a XML-based description of a learning content with a wide range of scientific elements. By using XML-based content, we can create course collection and then extract and dispatch selected parts of learning content in an appropriate (responsive) format by using simple transformation. This allows producing a responsive learning content on demand in function of mobile unit's characteristics.

Figure 5: The course semantics schema.

In our concept we promote the use of XML as a medium-neutral data format for data storage and processing that allows the learning contents to be classified hierarchically, to be structured at the desired level of granularity and to be adjusted to different contexts, situations and devices. In addition, we propose a method for external applications integration through Web services (Madjarov, 2012). This allows extending existing LCMS without modifications of their own source code. The feasibility of this concept was tested during the academic year as part of AMETICE system (Moodle based e-Learning system).

CONCLUSIONS

The benefits of building responsive design into e-Learning courses are obvious and many. While the extra time needed to create these courses can't be argued, it is well worth the effort because responsive e-Learning delivers single version of a course to multiple devices. In our concept an adaptive m-Learning system is based on responsive course design that combines wide range of multimedia components and put scrolling as principle of page (chapter) design. New mobile media are very well suited to eBook style. Like an eBook, a course chapter retains unlocked the scroll bar, but keep a page to a single learning objective or activity, i.e. context oriented since its pedagogical objectives remain intact.

Future research efforts will be focused on: (1) the integration of HTML5 applications built with Apache-Cordova technology in our Web servicebased concept. When done properly we can create multi-platform applications that is just as responsive to the user as a native application; (2) the feasibility of personalizing and adapting the m-Learning applications and contents to learner preferences, in regard to the increasing capabilities of mobile devices; and (3) the development of an adapted learning-podcasting service that will be used in a mobile pedagogical (social) networks.

Figure 1 :

 1 Figure 1: The main Responsive Web Design task: reshaping depending on screen sizes and resolutions.

Figure 2 :

 2 Figure 2: Mobile browsers tests results for Web options support: Θ partially supported; √ supported;  none supported; p plug-in needed.

  Figure 3: HTML5 page with CSS3 styled text, MathML, SVG and raster image in Firefox mobile browser (available at: http://139.124.26.245/tbrowser/).

Figure 4 :

 4 Figure 4: Fragment of source code from Web page in Figure 3 with stylesheet breakpoints, MathML, SVG and Base64encoded image.

√ √ √ √  MathML 

ACID3 √ √ √ √ Θ √ Θ


Remy Vauche

Eloi Muhr

Olivier Fourquin

Sylvain Bourdel

Jean Gaubert

Nicolas Dehaese

Stephane Meillere

Herve Barthelemy

Laurent Ouvry

A 100 MHz PRF IR-UWB CMOS Transceiver With Pulse Shaping Capabilities and Peak Voltage Detector

published or not. The documents may come

Abstract-This work presents a high rate IR-UWB transceiver chipset implemented in a 130 nm CMOS technology for WBAN and biomedical applications in the 3.1GHz-4.9GHz band. The transmitter is based on a pulse synthesizer and an analytical upconverted Gaussian pulse is used to predict its settings. Its measured peak to peak output voltage is equal to 0.9Vpp on a 100 load for a central frequency of 4GHz, and a supply voltage of 1.2V, which gives an emitted energy per pulse of 0.64pJ. The receiver is a non-coherent architecture based on a LNA followed by a peak voltage detector. A BER of 10 -3 is measured for a 3.1GHz-4.9GHz input peak-to-peak pulse amplitude of 1.1mV which corresponds to a sensitivity of -85.8dBm at 1Mbps and gives a communication range estimated to 1.9m.

Index Terms-IR-UWB, Ultra-Wideband CMOS transceiver, WBAN, non-coherent receiver, Pulse shaping Capabilities

I. INTRODUCTION

Ultra-WideBand Impulse Radio (IR-UWB) is a well-known technique based on the transmission of short duration pulses. Such approach is well suited for low range, medium rate and low power applications such as Body Area Network (BAN), Wireless Sensor Network (WSN) or ranging [START_REF] Dokania | A Low Power Impulse Radio Design for Body-Area-Networks[END_REF] [START_REF] Morche | A Double Quadrature UWB Receiver for wide Range Localization Applications with sub-cm Precision[END_REF]. Since the Federal Communications Commission (FCC) has approved UWB communications in the 3.1-10.6GHz bandwidth, many successful works have been reported showing the ease of making energy efficient systems with IR-UWB.

Recently, the research field has moved towards new challenges such as system reliability, integration and cost issues or high energy efficiency. In this context, a few meter range 100Mbps transceiver designed for BAN applications is presented. Here, the approach is to alleviate as much as possible the system constraints while allowing high tuning capabilities to ensure a successful design.

High energy efficiency is intrinsically based on the gated nature of the IR-UWB signal. To fully exploit this property, duty-cycled systems can be used. A first approach consists in duty-cycling the system at the bit rate [START_REF] Dokania | A Low Power Impulse Radio Design for Body-Area-Networks[END_REF] which improves the system immunity to narrow band interferers and reduces power consumption. For low bit rates (less or equal to 1Mbps), an on-time less than ten percent of the bit time can be reached with fast turn-on devices and pulse coupled oscillator based architectures for synchronization [START_REF] Wang | A Crystal-Less Self-Synchronized Bit-Level Duty-Cycled IR-UWB Transceiver System[END_REF]. A lower on-time (close to 30ns) has been reached with a 100MHz All Digital -Clock and Data Recovery (AD-CDR) having 10ns jitter [START_REF] Vigraham | A Self-Duty-Cycled and Synchronized UWB Pulse-Radio Receiver SoC With Automatic Threshold-Recovery Based Demodulation[END_REF]. However, when a medium rate (e.g., [START_REF] Dokania | An Ultralow-Power Dual-Band UWB Impulse Radio[END_REF].6Mbps) is considered, the Pulse Repetition Period (PRP) is around a few tens of nanoseconds (e.g., 66.7ns) and the precision of the dutycycling clock must be high enough (a few nanoseconds) in order to reduce efficiently the power consumption. In this case, the clock frequency of the AD-CDR must be increased up to 1GHz which impacts the system power consumption [START_REF] Vigraham | A Self-Duty-Cycled and Synchronized UWB Pulse-Radio Receiver SoC With Automatic Threshold-Recovery Based Demodulation[END_REF]. Solutions based on synchronized modulation schemes [START_REF] Crepaldi | A 0.07~mm Asynchronous Logic CMOS Pulsed Receiver Based on Radio Events Self-Synchronization[END_REF] have also been proposed. However, they require the transmission of additional pulses for synchronization which reduces the energy efficiency and the payload bit rate of the system.

For medium rate, a solution to reduce power consumption is to achieve a duty-cycling at burst rate. As presented in [START_REF] Sparrow | High rate UWB CMOS transceiver chipset for WBAN and biomedical applications[END_REF], this approach can reach the same energy efficiency as in a bit rate duty-cycled system, while the duty-cycling clock precision is moved towards a few tens of microseconds which releases constraints on the CDR jitter characteristics. In this case, the required idle time T IDLE to maintain constant the Power Spectral Density (PSD) is equal to T SWEEP (1 -PRF MEAN / PRF BURST ) where T SWEEP is the sweep time imposed by FCC to measure the PSD (1ms), PRF BURST the Pulse Repetition Frequency (PRF) during the burst, and PRF MEAN the mean PRF averaged on T SWEEP . By considering PRF MEAN (resp. PRF BURST ) is equal to 15.6MHz (resp. 100MHz), the ideal ontime T ON is 156s, the ideal idle time T IDLE is 844s, and the number of pulses per burst is 15.6k. In the context of a dutycycled at burst level, the synchronization scheme is no more an issue and is not addressed in this paper.

However, a high rate capability (around 100MHz) is decisive to minimize the ratio PRF MEAN / PRF BURST and to maximize the power reduction. Such approach also requires a high data rate CDR which can be duty cycled at the burst rate like the receiver. Moreover, the acquisition time of an AD-CDR will not impact the energy efficiency because the number of pulses required for the acquisition (32 in [START_REF] Vigraham | A Self-Duty-Cycled and Synchronized UWB Pulse-Radio Receiver SoC With Automatic Threshold-Recovery Based Demodulation[END_REF]) is negligible regarding the number of pulses in the burst.

Regarding the transmitter, controlling the radiated spectrum is necessary to comply with standard masks or to reject spectrum side-lobes. Passive solutions based on external filters or integrated filters can be used [START_REF] Bourdel | A 9pJ/Pulse 1.42Vpp OOK CMOS UWB Pulse Generator for the 3[END_REF] to shape the pulse but they dramatically impact the size and the cost. Solutions based on active devices [START_REF] Kim | Fifth-Derivative Gaussian Pulse Generator for UWB System[END_REF][9] suite better the integration issues but need high tuning capabilities to compensate Process Voltage and Temperature (PVT) variations. In this context, several solutions have been proposed to control the shape of the A 100MHz PRF IR-UWB CMOS Transceiver with Pulse Shaping Capabilities and Peak Voltage Detector R. Vauche 1 , E. Muhr 1 , O. Fourquin 1 , S. Bourdel 2 , J. Gaubert 1 , N. Dehaese 1 , S Meillere 1 , H. Barthelemy 1 , L. Ouvry 3 1 Aix Marseille Univ, Univ Toulon, CNRS, IM2NP, Marseille, France 2 Université Grenoble-Alpes, CNRS, IMEP-LAHC UMR 5130, F-38000 Grenoble, France 3 Université Grenoble Alpes, F-38000 Grenoble, France, CEA, LETI, MINATEC Campus, F-38054 Grenoble, France pulses [START_REF] Norimatsu | A UWB-IR Transmitter With Digitally Controlled Pulse Generator[END_REF][11] [START_REF] Bourdel | Low-cost CMOS pulse generator for UWB systems[END_REF]. Among these solutions, digital synthesis [START_REF] Bourdel | Low-cost CMOS pulse generator for UWB systems[END_REF] is a promising approach because it operates without inductor and allows a fine tuning of the pulse shape to be obtained.

On the receiver side, Non-Coherent (NC) detection is often used to reduce complexity and save power in IR-UWB receivers. Among the different architectures, energy detectors based on super-regenerative or self-mixing coupled to integrators achieve good sensitivity performances [START_REF] Pelissier | Super-Regenerative Architecture for UWB PulseDetection: From Theory to RF Front-End esign[END_REF] [START_REF] Lee | A 2.5 nJ/bit 0[END_REF]. However, the need for synchronization schemes operating at symbol rate increases the integration issues and reduces the maximum achievable bit rate. Another way of designing NC receivers is to use Peak Voltage Detectors (PVD). On the one hand, PVDs suffer from a lower theoretical sensitivity since the signal is not collected all along the symbol duration time as it is in energy detectors. However, using PVDs can be justified in case of a short-range transceiver. On the other hand, PVDs are intrinsically asynchronous and are well suited for a burst duty-cycled receiver.

In this paper, a high rate 3.1GHz-4.9GHz IR-UWB transceiver for WBAN applications is presented. It uses a pulse synthesizer with pulse shaping capabilities on the emitter side and a PVD at the receiver side. Section II gives a system overview and analyses the theoretical performance in terms of Bit Error Rate (BER). Since the transmitter architecture is based on pulse synthesis, an analytical pulse model is proposed to predict the required pulse shape. Next, this shape is used to estimate BER performance of the peak voltage detector implemented in the receiver. Section III and section IV present respectively the transmitter and the receiver parts of the transceiver. In Section V, measurement results obtained with the proposed transceiver are presented and compared with previous published works.

II. SYSTEM OVERVIEW

In this section, the proposed IR-UWB transmitter, which is based on pulse synthesis technique, is presented and an analytical model of Gaussian up-converted pulse is proposed to predict the waveform to generate according to the targeted bandwidth. Next, after a presentation of the proposed NC IR-UWB receiver based on a peak voltage detector, its BER performance is estimated with the help of the presented pulse model.

A. Proposed IR-UWB Transmitter

To communicate in the allocated FCC band, an IR-UWB transmitter requires a pulse generator to transmit data. Lots of approaches have been developed to generate UWB signals but they have generally few capabilities regarding spectrum shapes programmability.

Solutions based on the modulation of a carrier by an analog signal, such as a triangular waveform or a raised cosine, have been proposed [15][16] in order to shape the envelop of the emitted pulse, to better fit standard masks, and to reduce side lobes. This approach is efficient in terms of side lobe rejection but lacks of reconfiguration and tuning capabilities. All digital architectures, which are more flexible than analog ones, have been proposed. They allow the pulse shape to be tuned in case of PVT variations [START_REF] Norimatsu | A UWB-IR Transmitter With Digitally Controlled Pulse Generator[END_REF] [START_REF] Park | An All-Digital 12pJ/Pulse IR-UWB Transmitter Synthesized from a Standard Cell Library[END_REF]. All-digital pulse generators are generally based on Digitally Controlled Oscillator (DCO) [START_REF] Mercier | An Energy-Efficient All-Digital UWB Transmitter Employing Dual Capacitively-Coupled Pulse-Shaping Drivers[END_REF], Digital Delay Line (DDL) [START_REF] Norimatsu | A UWB-IR Transmitter With Digitally Controlled Pulse Generator[END_REF], or both [START_REF] Park | An All-Digital 12pJ/Pulse IR-UWB Transmitter Synthesized from a Standard Cell Library[END_REF]. Compared to a DCO based generator, a DDL based pulse generator can be easily duty-cycled to reduce the power consumption since it requires shorter start-up time. Moreover, it is possible to adjust each sub-lob of the pulse to accurately tune the envelop and control the side-lobe level.

As shown in Fig. 1, an UWB pulse p(t) can be seen as a linear combination of N baseband pulses (g n (t)) as follows:

 1 1 1 () N n n n p n p p t A g t                 (1)
where A n and τ n are respectively magnitude and width of the n th baseband pulse. To synthesize p(t), two circuits can be associated, a N baseband pulses generator and a N baseband pulses combiner, as shown in Fig. 1. The baseband pulses generator allows each baseband pulse (from n=1 to N) to be generated according to the targeted width (τ n from n=1 to N).

Next, baseband pulses are combined by the baseband pulses combiner according to the targeted magnitude (A n from n=1 to N). The proposed IR-UWB transmitter is based on this approach.

To optimize the bit rate and/or the communication range with a NC receiver while the maximum mean power given by FCC is respected, it is proposed here to use a Random Alternate On-Off Keying modulation (RA-OOK) [START_REF] Sparrow | High rate UWB CMOS transceiver chipset for WBAN and biomedical applications[END_REF]. This modulation does not have discrete spectrum in its PSD but is similar to an OOK modulation from the receiver point of view. Indeed, to send a binary '0', no pulse is emitted. However, to send a binary '1', a positive pulse or a negative pulse is randomly sent. To generate these positive and negative pulses, a Delay-Based Binary Phase Shift Keying (DB-BPSK) modulator can be used [START_REF] Bourdel | A 9pJ/Pulse 1.42Vpp OOK CMOS UWB Pulse Generator for the 3[END_REF] [START_REF] Wentzloff | Delay-based BPSK for pulsed-UWB communication[END_REF]. It is based on a delay cell which will softly delay or not the CLOCK triggered signal according to the DATA logical state indicated in Fig. 1. This allows positive and negative pulses of a BPSK modulation to be approximated and also allows the discrete spectrum of the emitted PSD to be suppressed which reduces the mean emitted power. To obtain the best results, the CLOCK delay ( DB-BPSK ) must be equal to 1/(2•f M ) with f M , the frequency where the emitted power is maximum [START_REF] Wentzloff | Delay-based BPSK for pulsed-UWB communication[END_REF]. In this case, the mean emitted power can be increased until the authorized limit by increasing the bit rate and/or the energy of each pulse [START_REF] Bourdel | A 9pJ/Pulse 1.42Vpp OOK CMOS UWB Pulse Generator for the 3[END_REF].

This DB-BPSK modulator has been implemented in the proposed IR-UWB transmitter and it is also used to obtain a Random Alternate On-Off Keying modulation [START_REF] Sparrow | High rate UWB CMOS transceiver chipset for WBAN and biomedical applications[END_REF] by using the following rules. To send a binary '0', no rising edge is provided on CLOCK and no pulse is emitted. To send a binary '1', a rising edge is provided on CLOCK and a positive or negative pulse is emitted according to the state of DATA.

B. UWB pulse waveform estimation

The use of pulse synthesizers as IR-UWB transmitters implies the definition of an UWB pulse which can match with the targeted bandwidth, in order to get an ad hoc configuration for A n and τ n control inputs. To define this pulse, it is possible to compute the inverse Fourier transform of the targeted frequency mask, or to analyze the impulse response of a filter which has a compliant frequency response. The fifth [START_REF] Kim | Fifth-Derivative Gaussian Pulse Generator for UWB System[END_REF][20] and the seventh [START_REF] Low | Novel Low Cost Higher Order Derivitive Gaussian Pulse Generator Circuit[END_REF] [START_REF] Phan | 4.7pj/pulse 7th Derivative Gaussian Pulse Generator for Impulse Radio UWB[END_REF] Gaussian derivatives can also be used especially to match with FCC mask. However, these methods lead to a high number N of couples (A n , τ n ) to synthesize when a 3.1GHz -4.9GHz bandwidth is targeted. To address this bandwidth with a few couples (A n , τ n ), triangular or Gaussian up-converted pulses have to be preferred. However, the Gaussian up-converted pulse has a better spectral efficiency η SE than the triangular one. This spectral efficiency η SE can be defined as follows:

 2 2 2 1 ( ). ( ). L XdB SE REF p t dt Z BW p t dt E A        (2)
where Z L is the impedance load of the antenna connected to the pulse generator output, and E REF is the energy of a reference pulse which is here an up-converted cardinal sine having the same magnitude (A) of the emitted pulse and occupying the same bandwidth (BW -XdB ). In this case, it appears that the Gaussian up-converted pulse has an efficiency of 86% instead of 56% for the triangular one. Thus, to keep N low and a high emitted energy distribution, an odd Gaussian up-converted pulse is proposed here. It only requires N/2 values of A n in addition to one τ n value. Moreover, it can be easily tuned to match with different spectrum masks since the Fourier transform of a Gaussian function is another analytically computable Gaussian function.

The proposed odd Gaussian up-converted pulse can be written in time domain as follows:

     2 ( ) exp sin 2 G M p t A t f t      (3)
where A is the maximum of the Gaussian envelope and α, a parameter which sets the pulse bandwidth. In this case, the width τ n of every baseband pulses is also equal to the optimum value of τ DB-BPSK which is 1/(2•f M ). The maximum of the Gaussian envelope A can be related to the peak to peak magnitude A PP of the pulse as follows:

   2 exp . 2 4 PP M A A f            (4)
The magnitude of the single side Fourier transform of p G (t) can be written as follows:

  2 2 ˆ( ) exp 2 4 M G ff p f A                    (5)
and allows the α parameter to be computed as a function of the targeted bandwidth BW -XdB defined at -X dB as follows:

   2 2 dB dB/10 . 2 ln 10 X X BW      (6)
Moreover, the energy emitted per pulse E P-G can be computed from (4) and is equal to:

    2 2 4 1 exp 2 2 8 M PG L f A E Z                          (7)
This result can be approximated with an error of 5% with the following expression: and enables energy estimation for other works comparison with only few parameters [START_REF] Sparrow | High rate UWB CMOS transceiver chipset for WBAN and biomedical applications[END_REF]. An example of this Gaussian up-converted pulse is shown in Fig. 2 for the targeted 3-5GHz bandwidth. However, as the Gaussian envelope defined here by A•exp(-α•t 2 ) is never equal to zero and that an infinite duration pulse cannot be synthesized, it is necessary to approximate p G (t). In Fig. 2, p G (t) is approximated by the height higher monocycle of the Gaussian pulse. This approximation shows a loss of 2dB in frequency domain and an unwanted side lobe around 1.5GHz. The both can be corrected thanks to pulse synthesis by using a fine tuning of A n values.

   dB/10 2 dB ln 10 2 X PG L X A E Z BW     (8)

C. Proposed Non-Coherent Receiver

Usually Non-Coherent UWB receivers use energy detection for pulse detection. In these architectures an image of the pulse energy is obtained by squaring and integrating the incoming signal during the symbol duration time. The signal amplitude at the squarer output, which varies as the square of the amplitude of the input signal, quickly becomes very small and unusable when the receiver moves away from the transmitter. Consequently, in order to obtain signals which exceed the noise floor of the stages downstream of the squarer, a large amount of amplification must be provided by the Low Noise Amplifier (LNA),, whereas achieving high amplification gain with large bandwidth is difficult in low cost CMOS technologies. Although it is less efficient from a theoretical point of view, a NC receiver based on peak voltage detection could provide an interesting alternative which relaxes design constraints regarding the LNA gain, while being compatible with the requirements of many short range applications.

The proposed NC receiver architecture which uses peak voltage detection is described in Fig. 3. It is composed of a band pass type LNA that drives an asynchronous peak detector based on envelope detection. The peak detector provides a short duration (1.5ns) baseband pulse for each incoming UWB pulse. The single ended numeric signal is then converted into a Low Voltage Differential Signaling (LVDS) signal in order to provide standardized signals to the board interface.

D. Performance analysis of peak voltage detection

In this section, an estimation of the BER which can be achieved with peak voltage detection is presented by considering an OOK coding, an AWGN channel, and no multipath propagation.

As shown in [START_REF] Van Trees | Detection, Estimation, and Modulation Theory, part I[END_REF][24], the dimensionality of the space of finite energy signals is about 2 • B • T + 1 with a bandwidth B and a time spread T. From its equivalent formula in the discrete domain, it is possible to perform the BER calculation. In this work, it is then assumed that, during the symbol duration time T S , the continuous time domain signal at the receiver input can be replaced by P discrete samples of the received signal obtained at the Nyquist rate:

 2 1. MAX S P f T      (9)
Given that, for the 3.1GHz-4.9GHz bandwidth UWB signal considered here, the noise bandwidth (1.8GHz) is close to the maximum spectrum frequency f MAX (4.9GHz), we also assume that the noise samples are independent. The discrete samples of the received signal are the sum of the samples of the unnoisy signal and of the noise. Assuming an AWGN channel and the independence of the noise samples, the probability density function of the noise voltage for each sample is given by:

 2 2 2 1 ( ) exp . 2 2 N N N v pv         (10)
Assuming that the noise bandwidth is the same that the bandwidth (BW) of the signal, the noise variance  N at the LNA output is given by:    2 10 log 10 log

N LNA LNA LNA K T BW G NF R           (11)
where K is the Boltzmann constant, T the absolute temperature, R LNA , G LNA and NF LNA the input resistor, the gain, and the noise figure of the LNA.

For an OOK coding, the error probability is given by:

 Pr 0.5 Pr 0.5 Pr E FA ND      (12)
where Pr FA is the probability of a false alarm and Pr ND is the probability of a non-detection. A false alarm occurs when the detector output switches to the high level while no pulse has been sent during the symbol duration time. So a false alarm occurs when the noise level crosses the value of the decision threshold S D for at least one of the noisy samples during the symbol duration time. So, the probability of a false alarm is given by: where P is the number of samples during the symbol duration time T S given by [START_REF] Dokania | A 6μW ,100Kbps, 3-5GHz, UWB impulse radio transmitter[END_REF]. A non-detection occurs when a pulse is sent and the magnitude of the noisy signal remains lower than the decision threshold during the symbol duration time. This corresponds for the equivalent discrete signal when no noisy signal sample crosses the value of the decision threshold S D during the symbol duration time. So the probability of a non-detection is given by:

 2 2 2 1 Pr 1 1 exp 2 2 D P FA N S N v dv                     (13)
   2 2 2 0 1 Pr exp 2 2 D S i S V t P ND i N N v dv                      (14)
where V S (t i ) is the magnitude of the discrete samples of the unnoisy signal at the peak voltage detector input. The shape of V S (t i ) depends on the signal bandwidth and its magnitude depends on the received power and the LNA gain. In our case V S (t i ) is computed from (3) knowing the emitted power at the emitter side, the communication range, and the LNA gain. For a given received power, a given data rate and a given decision threshold, the BER value is obtained with the equations ( 12), ( 13) and ( 14). For a given BER, there is an optimal decision threshold S D that minimizes the received power and balances Pr FA and Pr ND . So, by using equations ( 12), ( 13), ( 14) and an iterative algorithm, the receiver sensitivity can be obtained for a given BER. Finally, the use of the Friis formula with this minimum received input power gives the achievable range for a given emitted power. By using the UWB pulse defined by (3) for a data rate of 100Mbps with an OOK encoding and a 1.8GHz -10dB bandwidth centered at 4GHz, the results plotted in Fig. 4 are obtained. The same analysis has been done for an ideal Energy Detector (ED) in the appendix. Results are also shown in Fig. 4 for an integration time equal to the pulse width (2ns) when a 100Mbps OOK is considered. Results are also shown in Fig. 4 and even if the peak voltage detection shows a lower theoretical sensitivity than energy detection, it is sufficient for many short range applications. Indeed, it is possible to observe in Fig. 4 that a theoretical range of 8.28m is achievable with a peak voltage detection for a BER value of 10 -3 when a power of -78dBm is received. This range is obtained when a 0.9Vpp pulse on a 2×50differential antenna is considered on the emitter side, which is achievable in CMOS technologies [START_REF] Bourdel | A 9pJ/Pulse 1.42Vpp OOK CMOS UWB Pulse Generator for the 3[END_REF].

III. TRANSMITTER DESIGN

The presented pulse synthesizer shown in Fig. 1 has been designed in order to synthesize UWB pulses for bandwidths upper than 1.5GHz centered on a frequency between 3GHz and 5GHz. To satisfy the targeted bandwidth, the designed pulse synthesizer is able to generate UWB pulses composed of 1 to 8 baseband pulses. All the baseband pulse widths  n are controlled by the same biasing voltage except the first and the last in order to compensate for their slower responses due to their positions. Finally, a DB-BPSK pulse manager is integrated in order to implement RA-OOK modulation which allows bit rate and/or pulse energy to be increased while legal maximum powers are respected. The design of each part shown in Fig. 1 is now presented.

A. Baseband Pulses Generator Design

The baseband pulses generator is composed of two parts, a Voltage Controlled Delay Line (VCDL), and a fast logic stage as shown in Fig. 5.

The VCDL is here a delay line where a rising edge is propagated through tunable delay cells. Each delay cell is built of two buffered stages based on a CMOS inverter loaded by a capacitor (D-C1) in series with a NMOS analog switch (D-M3) which gives a large delay tuning range. The width of each baseband pulse (τ n ) depends on the propagation delay time of the n th stage of the VCDL which is set by D-M3, and D-C1 sizes and V d-n value. In this implementation, all V d-n are connected to the same control voltage since the proposed odd up-converted Gaussian pulse requires similar τ n , except for the first (V d-1 ) and the last (V d-N ) delay cells which are slower.

Next, the fast logic stage combines rising edges at the output of the n th cell (A) and at the output of the n+1 th cell (B) using a fast logic gate to achieve the n th baseband pulse (g n (t)). To increase the switching speed and also the maximum pulse central frequency, the fast logic stage has been modified from the architecture presented in [START_REF] Vauche | Fully tunable UWB pulse generator with zero DC power consumption[END_REF]. Then, the number of nodes has been reduced in order to limit the parasitic capacitances due to interconnections. However, this fast logic stage is able to generate the complementary baseband pulses required by the baseband pulses combiner (EPn and ENn).

B. Baseband Pulses Combiner Design

To generate UWB pulses with 1 to 8 baseband pulses, the baseband pulses combiner is based on 8 H-bridge paths going through the load as shown in Fig. 6. Each H-bridge path is composed of two transmission gates (S-M1 and S-M2) and is driven by the balanced baseband pulses on both sides of the load. The magnitude of the recombined baseband pulses (A n ) is controlled by applying different control voltages (V a-n ) at the top of each H-bridge path. An advantage of this structure is that the obtained A n is quasi-proportional to V a-n and also can be easily controlled. To alternate the sign of the baseband pulses, V a-n are alternatively applied to the top and to the bottom of the paths. The baseband pulses combiner has been designed in order to be able to drive a 100Ω differential antenna with a peak-to-peak magnitude equals to the supply voltage in post-layout simulations (S-M1 and S-M2 widths are respectively equal to 200µm and 480µm).

C. DB-BPSK Manager Design

As shown in Fig. 7, the DB-BPSK manager consists of two paths, one where CLOCK is delayed, and another one where CLOCK is not delayed. It uses standard CMOS logic gates as buffers, transmission gates, and an OR gate. However, it is equally composed of one delay cell which allows CLOCK rising edge to be delayed when DATA is equal to '1'. As the delay cell is the same as the one used in the VCDL, and that all baseband pulses of the synthesized pulse have the same time duration ( n ), V d-DATA has to be connected to V d-n to obtain  DB-BPSK equal to 1/(2•f M ). In this configuration, the DB-BPSK modulation is close as much as possible to the BPSK one in terms of spectrum. To ensure the realization of this configuration, buffers are placed on the both sides of the delay cell, and the same buffers are equally placed on the no delayed path.

IV. NON-COHERENT RECEIVER DESIGN

The presented NC IR-UWB receiver shown in Fig. 3 has been designed to be used with 3GHz-5GHz pulses. It consists of a LNA, a voltage peak detector, and a LVDS driver, which are described in this section.

A. LNA

The LNA architecture is depicted in Fig. 8. The LNA includes three stages: a LC matched input stage and two stages with active load. The LC matched input stage (M 1 , M 2 ) achieves a bandpass response in the 3-5GHz bandwidth [START_REF] Battista | 6-10 GHz Ultra Wide-Band CMOS LNA[END_REF]. In a low-cost perspective, the use of an off-chip filter should be avoided, so the LNA bandpass response reduces both parasitic signals and equivalent noise bandwidth before the pulse detector input. The two active load stages (M 3 , M 4 and M 5 , M 6 ) provide an additional gain controlled by the voltage V GAIN . The LNA sizing process is done to minimize the noise figure and to maximize the voltage gain as presented in [START_REF] Battista | 6-10 GHz Ultra Wide-Band CMOS LNA[END_REF]. For such a high gain amplifier at high frequency, signal integrity and stability are important issues. The power supply pads (V DD ) are decoupled by using several MIM and MOS capacitors (C DEC in Fig. 8) in order to get a large on chip capacitor value (a few hundred of pF) leading to low return path ground impedance at high frequencies.

B. Peak detector

The proposed peak detector architecture is given in Fig. 9. The incoming UWB pulse is first rectified and a low-pass filter is then used to extract the pulse envelope and to reduce the noise bandwidth. Finally a decision stage based on a comparator followed by a shaping circuit provides a rectangular pulse when pulse envelope crosses the decision voltage V DET .

1) Rectifier stage

The rectifier is the key element of the detector architecture. A wide bandwidth rectification is achieved by using a biased CMOS inverters chain. Part (b) of Fig. 10 depicts the operation of a biased CMOS inverter. If the biasing voltage V BIAS is set to V P , the CMOS inverter operates close to the point A of its characteristic. By using this operating point, the positive part of the input signal is amplified and inverted while the negative part of the input signal is suppressed. Conversely, if V BIAS is set to V N the point B, the CMOS inverter operates close to the point B of its characteristic. In this case, the negative part of the input signal is amplified and inverted while the positive part of the input signal is suppressed. By cascading CMOS inverters alternatively biased with V P and V N voltages, the input signal is rectified and also amplified. The rectifier stage uses five AC coupled CMOS inverters alternatively biased with voltage V P and V N . Part (c) of Fig. 10 shows the simplified waveforms obtained in the case of an even number of rectifier cells.

2) Low-pass filter

The low-pass filter consists of three cascaded AC coupled first-order filters with a cutoff frequency of 2GHz. As shown in Fig. 11, the first order low-pass cell is achieved by a singleended transconductance loaded by a capacitor and a resistor. This simple architecture does not allow the filter cutoff frequency to be adjusted, but avoids the implementation of a common mode feedback circuit commonly used in gm-c filters.

3) Decision stage

As shown in Fig. 12, the decision stage circuit is built from CMOS inverters. The gate of the first inverter is externally biased by a control voltage V CTRL which allows the noise amplitude at its input to be maintained up the switching threshold of the inverter. A set of two other CMOS inverters follows the first one in order to generate a clean logic signal. The amplitude of the UWB pulse envelope is compared to a threshold voltage V DET at a distance S D of V CTRL . If the pulse envelope crosses the threshold V DET , a pulse is detected and the output signal goes to V DD long enough to enable the postdetection circuits to properly process the signal. The decision threshold value is set to optimize the trade-off between the false alarm rate, due to the influence of noise, and the nondetection rate and so the receiver sensitivity.

C. LVDS transmitter

The LVDS interface allows high bit rates (up to 100Mbs -1 ) to be achieved for chip to board interface with a low sensitivity to parasitic capacitors and inductors. The implemented LVDS transmitter circuit uses the typical architecture based on four MOS switches in a bridge configuration as shown in Fig. 13. When the pair (M1, M3) is active and the pair (M2, M4) is off, the polarity of the output current is positive together with the differential output voltage. Conversely, if the pair (M1, M3) is off and the pair (M2, M4) is active, the polarities of the output current and of the differential output voltage are reversed. The required termination resistors are provided by an off-chip standard LVDS receiver chips. With a nominal 100 load on the LVDS receiver side, the differential swing at the output should fall within the LVDS standard specifications (350mV). Nevertheless, since the supply voltage of the integrated circuit is 1.2V, the common mode voltage does not match with the LVDS standard specifications (1.25V). The common mode voltage is set here at V DD /2 (0.6V) by the common mode feedback block and an external chip receiver supporting such a common mode voltage must be used.

V. MEASUREMENT RESULT

The emitter and the receiver of the presented transceiver have been realized in two separated chips using the same 130nm CMOS technology from STMicroelectronics and have a supply voltage of 1.2V. Even if the transmitter could have better performances with a more recent process due to its alldigital architecture, this technological node achieves good performances up to 5GHz and is a good tradeoff between cost and performances [START_REF] Morche | A Double Quadrature UWB Receiver for wide Range Localization Applications with sub-cm Precision[END_REF]. Emitter and receiver have been integrated on two separated chips, and measured with two different setups for a better test convenience.

A. Transmitter

As shown on part (a) of Fig. 14, the pulse synthesizer has a die area of 1.4mm 2 and a core area of only 0.11mm 2 . To measure the pulse synthesizer, it has been packaged into a QFN32 package and three boards (represented in parts (b) of Fig. 17) have been realized: one RF-board to receive the chip, another one to generate the required supply and control voltages and an USB-board to connect with a computer. The test setup principle is given in part (a) of Fig. 17.

To generate a pulse for a given bandwidth with a pulse synthesizer, available values of A n and  n have been characterized for the tuning ranges of V a-n and V d-n . Next, the analytical pulse model has been used to set all V a-n and V d-n voltages according to the targeted central frequency f M and bandwidth BW -XdB . Finally, all V a-n are adjusted in order to optimize rejection and also respect FCC requirements. To verify that emitted pulses are well-suited to the targeted bandwidth, or to compensate for spectrum variations generally due to slow phenomenon such as temperature variations, the synthesizer can be controlled by a calibration system as described in [START_REF] Goavec | An Efficient Method of Power Spectral Density Estimation for On-Chip IR-UWB Transmitter Self-Calibration[END_REF].

From a Gaussian up-converted pulse with f M and BW -10dB respectively equal to 4GHz and 1.8GHz, an UWB pulse has been synthesized by adjusting the control voltages and is shown in part (a) of Fig. 15. Its peak to peak voltage is about 0.9Vpp and its energy is equal to 0.64pJ on a 100 load (which is greater than the 0.52pJ estimated from (8) since some parts of the pulse have a magnitude greater than the model). The delayed pulses needed for RA-OOK signaling are represented in part (b) of Fig. 15.

Due to the gated nature of IR-UWB signals, the mean consumed power can be written as a function of the Pulse Repetition Frequency (PRF) as follows [START_REF] Vauche | A remotely UHF powered UWB transmitter for high precision localization of RFID tag[END_REF] :

   C AC DC P PRF E PRF P     ( 15
)
where P DC is the DC consumed power when no pulse is emitted and E AC is the additional consumed energy when a pulse is emitted. These parameters can be extracted from two powers measured at different PRF which are here for the worst configuration 115µW@100kHz and 14.7mW@100MHz, which give a P DC equals to 100µW and an E AC equals to 146pJ per pulse. At 1MHz, the pulse generator consumes 250µW which gives a 250pJ energy consumed per pulses. Finally, the pulse shaping capabilities of the pulse generator are presented in Fig. 16. The targeted -10dB bandwidths, 3.5GHz (a) and 2.4GHz (b), are obtained in measurement thanks to the analytical pulse model given in (3). It appears that it is possible to change the central frequency from 3GHz to 5GHz by varying the delay  in the VCDL. For a pulse centered at 4GHz, the fine tuning of V a-n voltages allows A n to be linearly controlled between 0V and 0.45V when V a-n is between 0V and the supply voltage 1.2V.

The pulse generator is also able to address several bandwidths and several center frequencies with a high output dynamic as shown in Table I which gives a summary of different reported pulse generators with pulse shaping capabilities. The presented pulse generator achieves a quite good output dynamic (A =0.45V) compared to other reported works. The tradeoff between output dynamic and power consumption is comparable to other all-digital synthesizers, which generally consume more power than analog pulse generators as it can be observed in Table I.

B. Receiver

The non-coherent receiver core area is 1.7 mm 2 as indicated in part (b) of Fig. 14. The chip where the receiver has been integrated includes other circuits which are out of the scope of this paper. It has been packaged in a QFN88 package. The DC power consumption of the non-coherent receiver is 30.5mW. From post-layout simulations, the LNA achieves for the 3.1GHz-4.9GHz band, a minimum voltage gain of 28dB, a maximum noise figure of 3dB, a maximum S11 of -11.5dB, and a current consumption of 17mA.

The performance of the non-coherent receiver has been measured through the BER characteristic. As it can be seen in part (c) of Fig. 17, three boards have been realized, one RFboard to receive the chip, another one to generate all the required supply voltages and control signals and a PXI-board for connection with a PC. A 2GHz bandwidth pulse centered at 4GHz is used for the measurement setup. The measurement has been done using the same considerations than in the theoretical BER calculus of the section II.D: a sequence of successive '1' is sent to measure the probability of a nondetection and a sequence of successive '0' is sent to measure the probability of a false-alarm. The detection threshold is set to obtain the same number of false-alarms and non-detections. The number of successive '1' and successive '0' used for this measurement is 10 6 .The measurement has been done for a rate of 15.6Mbps and the receiver exhibits a 10 -3 BER for a pulse energy of 5.35 aJ. This corresponds to a -85.8dBm sensitivity at a 1Mbps rate for OOK modulations. This measured sensitivity value is much lower than the theoretical one presented in Fig. 4 (-98 dBm if downscaled to 1Mbps) since Fig. 4 gives the upper performance limit for a receiver using ideal blocks with no noise, no loss, and no bandwidth limitation. However, this sensitivity is comparable to the performances of other receivers based on energy detection.

C. Overview

The final budget link based on measurement results is presented in Table II and introduces A RX (resp. E RX ) which is the peak magnitude (resp. energy) of the received pulse, and S RX which is the receiver sensitivity for a given bit rate. By using the emitted pulse indicated in Fig. 15 and the measured receiver sensitivity, the range d is computed from Friis formula. Thus, the maximum communication range is estimated to 1.9m with the radiated pulse energy on a 100 differential load and the receiver sensitivity on a 50 load. In this budget link, no antenna gain and no fade margin are taken into account. This budget link shows that peak voltage detection is suitable for short range communications. The transceiver performances are compared with previous works in table III. This shows that peak voltage detection exhibits performances comparable to other type of detectors. The high output voltage of the transmitter combined with the receiver sensitivity finally gives a 1.9m communication range which is at the state of the art.

VI. CONCLUSION

A 3-5 GHz peak voltage detector based IR-UWB CMOS transceiver with pulse shaping capabilities has been presented in a 130nm CMOS technology. A non-coherent receiver based on a peak power detector has been presented. The detection is fully asynchronous. The receiver has a sensitivity of -85.8dBm for a rate of 1Mbps. This sensitivity allows detecting an input signal with a magnitude of 1.1mV which leads to a communication range of 1.9m. The receiver power consumption is about 30.5mW@100Mbps. The transmitter emits pulses with a high pulse shaping capabilities for a power consumption of 14.7mW@100Mbps.

APPENDIX

In this appendix, an estimation of the BER which can be achieved with an UWB energy detector is presented by considering an OOK coding, an AWGN channel, and no multipath propagation. The considered receiver architecture is similar to the one depicted in Fig. 3 but the peak voltage detector has been replaced by a perfect energy detector. This energy detector collects without any loss the signal energy available at the LNA output during the integration window. A perfect synchronization is assumed for the integration window.

A. Probability of error

During the integration window T S , the continuous time domain signal at the receiver input is replaced by P discrete samples of the received signal obtained at the Nyquist rate: 2

1.

MAX S P f T     (9)
Given that, for the 3.1GHz-4.9GHz signal bandwidth considered here, the noise bandwidth (1.8GHz) is close to the maximum spectrum frequency f MAX (4.9GHz), it is assumed that the noise samples are independent. Consequently, the discrete samples of the received signal are the sum of samples of the unnoisy signal and samples of the noise. Assuming an AWGN channel and the independence of the noise samples, the probability density function of the noise voltage for each sample is given by:

2 2 2 1 ( ) exp . 2 2 N N N v pv        (10)
Assuming that the noise bandwidth is the same that the bandwidth (BW) of the signal, the noise variance  N at the LNA output is given by:    2 10 log 10 log

N LNA LNA LNA K T BW G NF R           (11)
where K is the Boltzmann constant, T the absolute temperature, R LNA , G LNA , and NF LNA the input resistor, the gain and the noise figure of the LNA respectively.

For an OOK coding, the probability of error is given by:  Pr 0.5 Pr 0.5 Pr

E FA ND      (12)
where Pr FA is the probability of a false alarm and Pr ND is the probability of a non-detection.

B. Probability of non-detection

A non-detection occurs when the energy of the noisy signal integrated during the integration window T S remains lower than the decision threshold. The energy collected by the energy detector is given by:  

2 S i i P T W S N RP     (16)
where S i are the discrete samples of the unnoisy signal, N i the discrete samples of the noise, and R is the input impedance of the energy detector which is assumed to be real.

The sum of the discrete sample of the signal S i and of the noise N i is a random variable X i :

i i i X S N  . (17)
The P values of X i are P (independent) normally distributed random variables with a mean equal to the corresponding unnoisy signal sample and a variance equal to the noise variance  N . Consequently the random variable Y i defined as:

i i N X Y   (18)
has a unit variance and:

P P i i i i N X YY       (19)
is a random variable which is distributed according to the noncentral chi-squared distribution. The relation between Y and the energy (W) collected by the energy detector during T S can be written as follows:

2 SN T W Y RP     . ( 20
)
Y has two parameters: P the number of samples which specifies the number of degrees of freedom, and  which is related to the mean of the random variables by: where Q(x,k) is the cumulative distribution function of the central chi-squared distribution with k degrees of freedom, which is given by:

      2 , 2 , 2 kx Q x k k    (27)
where γ(z,a) is the lower incomplete Gamma function defined as follows:

 

) 30
So the probability of a non-detection can be obtained by using the cumulative distribution of Y:

C. Probability of false alarm

A false alarm occurs when the energy collected is greater than the decision threshold W D while no pulse has been sent during the integration window. The mathematical expressions given in the previous part are not valid in this case because when no pulse has been sent, the parameter defined in ( 21) is equal to zero. In the case where is equal to zero, the random variable Y defined in [START_REF] Wentzloff | Delay-based BPSK for pulsed-UWB communication[END_REF] becomes a normal law which is called Y' in the following. Thus, Y' is a random variable which is distributed according to the central chisquared distribution with P degrees of freedom and with a probability density function given by:    

) 32
The cumulative distribution function of Y' is also given by:

      ' 2 , 2 , 2 Y Px F x P P    . ( 33
)
The energy collected W by the energy detector during the integration window can be computed and is given by:

A false-alarm occurs when the energy collected by the energy detector given by W is greater than the decision threshold W D . It occurs when:

D. BER Calculation

Knowing the probability of a non-detection given by (31) and the probability of a false-alarm given by (37), [START_REF] Bourdel | Low-cost CMOS pulse generator for UWB systems[END_REF] gives the probability of error for a given received power, a given integration window and a given decision threshold. Similarly as in the case of a peak voltage detection, there is an optimal decision threshold W D that minimizes the received power and balances Pr FA and Pr ND . So, by using relations ( 12), ( 31), (37) and an iterative algorithm, the receiver sensitivity can be obtained for a given BER. Finally, the use of Friis formula with this minimum received input power gives the achievable range for a given emitted power. By using the UWB pulse defined by (3) with a -10dB bandwidth of 1.8GHz centered at 4GHz, the theoretical BER obtained with an energy detector is given in Fig. 4 for a date rate of 100Mbps with OOK encoding, an integration window of 2ns, and a LNA noise figure of 3dB.

Fig. 1 .

 1 Fig. 1. Proposed IR-UWB transmitter based on pulse synthesizing technique.

Fig. 2 .

 2 Fig. 2. Example of transient and frequency responses of the proposed Gaussian up-converted pulses with APP=2V, fM=4GHz, and BW-10dB=1.8GHz.

Fig. 3 .

 3 Fig. 3. Proposed IR-UWB NC receiver based on peak voltage detection.

Fig. 4 .

 4 Fig. 4. Bit error rate versus the received power for OOK modulated 3.1GHz-4.9GHz pulses at a rate of 100Mbps and with NFLNA = 3dB for a Peak Voltage Detector (PVD) and an Energy Detector (ED) (a). Theoretical range with PVD as a function of the receiver sensitivity when a 0.52pJ (0.45V peak magnitude) pulse is emitted on a 2×50Ω differential antenna (b).

Fig. 5 .

 5 Fig. 5. Design of the proposed baseband pulses generator.

Fig. 6 .

 6 Fig. 6. Design of the proposed baseband pulses combiner.

Fig. 7 .

 7 Fig. 7. Design of the proposed DB-BPSK manager.

Fig. 8 .Fig. 9 .

 89 Fig. 8. Schematic of the low noise amplifier circuit.

Fig. 10 .

 10 Fig. 10. Rectifier stage: (a) Schematic, (b) CMOS inverter based rectifier cell principle, (c) Simplified waveforms obtained at the rectifier input/output.

Fig. 11 .

 11 Fig. 11. First order low-pass filter schematic.

Fig. 12 .

 12 Fig. 12. Decision stage circuit.

Fig. 14 .Fig. 15 .

 1415 Fig. 15. Comparison of the ideal Gaussian up-converted pulse with measured transient and frequency responses of a pulse with A=0.45V, fM=4GHz, and BW-10dB=1.8GHz for DATA = '0' (a). Comparison between pulses required for RA-OOK (DATA = '1' and DATA = '0') (b).

Fig. 16 .

 16 Fig. 16. Comparison of ideal Gaussian up-converted pulses with measured transient and frequency responses of a 4.85GHz centered pulse with BW-10dB = 3.5GHz (a) and of a 5GHz centered pulse with BW-10dB = 2.4GHz (b).

Fig. 17 .

 17 Fig. 17. Measurement setup principle (a), emitter measurement setup (b) receiver measurement setup (c).

( 22 )

 22 The probability density function of Y is given by: where I  (y) is order modified first kind Bessel function which is given by:       Finally the cumulative distribution function of Y is given by:

       A non-detection occurs when the energy collected by the energy detector is lower than the decision threshold W D

  of a false alarm can be obtained by using the cumulative distribution of Y':

TABLE I :

 I COMPARISON OF RECENT PULSE GENERATORS WITH PULSE SHAPING CAPABILITIES

	[REF]	[9]	[11]	[16]	[17]	[32]	[33]	This Work
	VDD (V)	0.9	1	1	0.9	1.2	1.2	1.2
	fM (GHz)	3.5-4.5	3-5	4	3.1-5	3.1-6	4.9	3-5
	BW-10dB(GHz)	0.5	0.5	2	0.5-1.4	1-1.86	3.5	1.8-3.5
	A (V)	0.25	0.305	0.25	0.048-0.063	0.2	0.11	0.45
	EAC (pJ/pulse)	N/A	2	N/A	8-12	N/A	N/A	146
	PDC (µW)	2.8	105	28	170	N/A	N/A	100
	PC@PRF (mW)	5.7	4.36	N/A	0.57	N/A	87.6	14.7
	PRF (MHz)	0.1	15.6	32	50	<500	800	100
	EC (nJ/pulses)	0.057	0.28	0.008	0.012	0.0036-0.02	0.1	0.147
	Tech. (CMOS)	90nm	90nm	65nm	90nm	90nm	130nm	130nm
	**on a 100 ohm
	s impedance

TABLE II :

 II

		FINAL LINK BUDGET
	PRFMEAN(MHz)	0.5
	Db (Mbps)	1
	BW-10dB(GHz)	1.8
	EP (pJ)	0.52***
	A (V)**	0.45
	ARX (mV)*	1.1
	ERX (pJ)	5.35•10 -6
	SRX@1Mbps(dBm)	-85.8
	d(m)	1.9
	*on a 50Ω impedance
	**on a 100Ω impedance
	*** estimated from (8)

TABLE III

 III

						PERFORMANCE COMPARISON WITH PREVIOUS WORKS
	Ref	Year	Tech. (CMOS)	size (mm 2 )	Pulse BW (GHz)	fM (GHz)	d (m)	Data Rate (bps)	Tx Power Cons. at Data Rate (mW)	Tx cons. nrj/b (pJ/b)	Rx Sensi. at Data Rate (dBm)	Rx Sensi. Scaled @ 1Mbps (dBm)	Rx Power Cons. at Data Rate (mW)	Rx cons. nrj/b (nJ/b)
	This Work		130nm	1.31	0.5-2	3.5-5	1.9	100M	14.7	147	-65.8	-85.8	30.5	0.305
	[3]	2013	90nm	1.7	0.5	3.5-4.5	0.5	100k	0.0085	60.7	-85	-75	0.11	0.8
	[16]	2015	65nm	2.25	0.5	7.5-9.5	1.2	500M	6.9	13.8	-60	-87	5.9	0.0118
	[29]	2011	90nm	0.6	3.6-4.3	2.9-3.8	1.5	1M	0.258	258	-60/-66	-60/-66	1.64-2.18	1.64
	[30]	2014	180nm	4.4	1.36	3-5	N/A	20M	13.4	670	-81	-94	3.54*	3.54
	[31]	2016	65nm	4.6**	0.5	3.1-10.6	1	1G	31.9***	31.9	-74	-104	27.8***	0.0278
	[34]	2008	180nm	0.36	1	3	0.04	500M	0.28	0.56	N/A	N/A	11	0.022
	[35]	2013	180nm	2.73	N/A	9-12	N/A	30k	0.022	747	-77	-61.7	0.037	1.2

*at 1Mbps, **with pads, ***with PLL


A Daumas

J Magalon

R Truillet

D Casanova

L Giraudo

J Veran

A Benyamine

F Dignat-George

G Magalon

F Sabatier

MD Brigitte Granel
email: bgranel@ap-hm.fr

Long-term follow-up after autologous adipose-derived stromal vascular fraction injection into fingers in systemic sclerosis patients

Long-term follow-up after autologous adipose-derived stromal vascular fraction injection into fingers in systemic sclerosis patients

ABSTRACT Introduction

Hand involvement confers a substantial handicap in work and daily activities in patients with Systemic sclerosis (SSc). Autologous adipose-derived stromal vascular fraction is as an easily accessible source of cells with regenerative effects. We previously performed a phase I openlabel clinical trial (NTC01813279) assessing the safety of subcutaneous injection of autologous adipose-derived stromal vascular fraction. Six and 12 month data have been reported. As patients were followed in our medical centre, we report their longer term outcome beyond the end of the trial.

Patients and method

Twelve females, mean age 54.5±10.3 years, initially enrolled in the clinical trial were assessed during a scheduled medical care, which took place between 22 and 30 months after treatment.

Results

Multiple patient-reported outcomes showed sustained improvement, in comparison with the assessment performed just before surgery: 62.5% in the Cochin Hand Function Scale, 51.1% in the Scleroderma Health Assessment Questionnaire, 33.1% in hand pain, and 88.3% in the Raynaud Condition Score. A decrease in the number of digital ulcers number was noted.

Mobility, strength and fibrosis of the hand also showed improvement. None of the 8 patients who had previously received iloprost infusion required new infusion.

Conclusion

Despite the limits of an open label study, the data are in favour of the long-term safety of the adipose-derived stromal vascular fraction injection. Two randomized double blind, placebocontrolled trials of this therapeutic agent are ongoing in the USA (NCT02396238) and in France (NCT02558543) and will help determine the place of this innovative therapy for SSc patients.

Introduction

Hand involvement is frequent in Systemic sclerosis (SSc) and confers a substantial handicap in work and daily activities [START_REF] Guillevin | Functional impairment of systemic scleroderma patients with digital ulcerations: results from the DUO Registry[END_REF]. To date, clinical care for the hand relies on vasodilators, cold and trauma protection, and regular physiotherapy. No antifibrotic therapy has proven effective.

The regenerative properties of cells derived from adipose tissue have been explored for over a decade. In 2002, Zuk identified and described a putative population of multipotent stem and progenitor cells within the stromal vascular fraction (SVF) cell population derived by enzymatic digestion of adipose tissue [START_REF] Zuk | Human adipose tissue is a source of multipotent stem cells[END_REF]. The SVF is composed of blood cells, fibroblasts, endothelial cells and their progenitors, pericytes, adipose stromal/stem cells (ASC) and preadipocytes. This population has been reported to possess multiple angiogenic, antiinflammatory, immunomodulatory and regenerative properties [START_REF] Daumas | State of the art. Autologous fat graft and adipose tissue-derived stromal vascular fraction injection for hand therapy in systemic sclerosis patients[END_REF].

We previously performed a phase I open-label single center clinical trial, called SCLERADEC (NTC01813279) assessing the safety and efficacy of adipose-derived stromal vascular fraction (ADSVF) in 12 SSc patients followed for 6 months [START_REF] Granel | Safety, tolerability and potential efficacy of injection of autologous adipose-derived stromal vascular fraction in the fingers of patients with systemic sclerosis: an open-label phase I trial[END_REF], and later reported their extending outcome at 12 months [START_REF] Daumas | Autologous adipose-derived stromal vascular fraction in patients with systemic sclerosis: 12month follow-up[END_REF]. We took advantage of routine medical follow-up of these patients to assess their longer term outcome. Procedures: Adipose tissue collection, ADSVF extraction and quality controls have previously been described [START_REF] Granel | Safety, tolerability and potential efficacy of injection of autologous adipose-derived stromal vascular fraction in the fingers of patients with systemic sclerosis: an open-label phase I trial[END_REF][START_REF] Daumas | Autologous adipose-derived stromal vascular fraction in patients with systemic sclerosis: 12month follow-up[END_REF][6]. ADSVF was obtained within 2 hours after lipoaspiration using the automated processing Celution800/CRS system (Cytori Therapeutics, San Diego, USA). An average of 3.76±1.85 x10 6 cells was injected into each finger as previously described [3-5, with video of the surgical technique in 6].

Patients and methods

Patients

Outcome: After completion of the trial patients continued their routine medical appointments in the internal medicine department. This allowed us to assess their hand function and other parameters using the same evaluators and same tools as those applied within the trial [START_REF] Granel | Safety, tolerability and potential efficacy of injection of autologous adipose-derived stromal vascular fraction in the fingers of patients with systemic sclerosis: an open-label phase I trial[END_REF]. This was performed during a single visit occurring within the range of 22 to 30 months from the date of the ADSVF injection.

Data analysis: Continuous data were summarized by mean±standard deviation and median [minimum-maximum]. CHFS, Scleroderma Health Assessment Questionnaire (SHAQ), Raynaud's Condition Score (RCS) and hand pain visual analogue scale (VAS) were analyzed as co-primary outcomes. Mean changes from baseline were analyzed using mixed model with cutaneous form (diffuse or limited) as fixed effect and time (2, 6, 12 and 24 months) as repeated effect. Least square mean differences form baseline were tested with Tukey adjustment. Significance was set at p<0.05 level after adjustment.

Results

Consistent with prior reports [START_REF] Granel | Safety, tolerability and potential efficacy of injection of autologous adipose-derived stromal vascular fraction in the fingers of patients with systemic sclerosis: an open-label phase I trial[END_REF][START_REF] Daumas | Autologous adipose-derived stromal vascular fraction in patients with systemic sclerosis: 12month follow-up[END_REF], no evidence of treatment-related adverse events was noted in any patient. Results of functional assessments are shown in Table 1 and Figure 1. Results show that the benefit reported at the 6 month time point of the trial is sustained at 22-30 months. For example, the long-term follow-up data for CHFS, SHAQ, and RCS endpoints showed 62.5%, 51.1% and 88.3% improvement over baseline respectively. It is worth noting that the decrease of the VAS for hand pain which had lost statistical significance at 12 months post-surgery [START_REF] Daumas | Autologous adipose-derived stromal vascular fraction in patients with systemic sclerosis: 12month follow-up[END_REF] regained significance at 24 months (33.1% decrease from baseline) (Figure 1). Improvement in objective endpoints such as Jamar grip strength reported at 6 months was sustained in the current assessment while pinch strength showed continued improvement. The modified Rodnan Skin Score focused on hands decreased from 10.9±4.9 at baseline to 8.8±5.9 although this difference did not achieve statistical significance. Lateral range of motion of the fingers assessed with the first corner distance and the sum of the 2-4th corner distances did not change significantly. The sum of pad to distal palmar line distances decreased over time for both hands without reaching significance p-value. The Kapandji score was high at baseline and did not change over time. The total number of digital ulcers (DU) number was 15 at entry and 6 at last visit.

Mean global disease severity score (Medsger scale) remained stable: 2.5 [START_REF] Guillevin | Functional impairment of systemic scleroderma patients with digital ulcerations: results from the DUO Registry[END_REF][START_REF] Zuk | Human adipose tissue is a source of multipotent stem cells[END_REF][START_REF] Daumas | State of the art. Autologous fat graft and adipose tissue-derived stromal vascular fraction injection for hand therapy in systemic sclerosis patients[END_REF] at baseline and 2.5 [0-3] at last clinical assessment. Comparison of changes in medical drugs from baseline to last visit showed that none of the 8 patients who had previously received iloprost infusion required new infusion. One patient with diffuse cutaneous SSc was prescribed an immunosuppressive medication due to progression of skin fibrosis and another patient with diffuse cutaneous SSc received sildenafil for recurrent ischemic digital ulcer. Three patients previously treated with a calcium channel blocker stopped this treatment due to side effect without worsening of Raynaud's phenomenon.

Discussion

While the current report is limited by the repetition of hand assessment in an extended followup from an original phase I open label study and the potential bias related to natural course of the disease and the seasonal effect, the long-term assessment is in favour of safety and durability of the effect. Given the costs associated with preparation of an autologous autologous cell therapy product to standards required by local, national, and European regulatory authorities it is important to assess the apparent durability of the response in order to begin to build a pharmaco-economic argument for or against this therapy. The current study, with evidence of apparently sustained-possibly progressive-improvement and the reduction in need for costly medications such as iloprost, represents the first step in building the data set needed for a cost-benefit assessment. Due to the lack of a placebo group, we are aware that we cannot be definitive about the causal link between the long-term hand improvement and ADSVF injection. However, it appears somewhat unlikely that the substantial magnitude of change over such a prolonged period could be attributed to a placebo effect and is unlikely to represent spontaneous sustained positive changes in the natural history of the disease. Rather, from a physiopathological point of view, we believe that the combined and synergistic pro-angiogenic, anti-inflammatory, anti-fibrotic and immunomodulatory effects possessed by the ASC and other progenitor cells inside the ADSVF [START_REF] Daumas | State of the art. Autologous fat graft and adipose tissue-derived stromal vascular fraction injection for hand therapy in systemic sclerosis patients[END_REF] could explain these clinical results. Of course, the final determination of this question will depend upon the results of the robust randomized, placebo-controlled trials such as the two trials currently underway in France and in the USA.

Conclusion

Although interpretation of these results is hampered by the lack of a control group, the use of autologous ADSVF is an innovative therapy which appears to provide benefit for patients suffering from SSc who have hand dysfunction. Two randomized double blind, placebocontrolled clinical trials with this form of ADSVF are ongoing: Scleroderma Treatment with Celution Processed Adipose Derived Regenerative Cells (STAR) in USA and SCLERADEC 2 in France. Such evidence will help guide health care professionals on the place of this therapy for SSc patients.

:

  Twelve SSc patients, all female, were enrolled from December 2012 through May 2013. The population was composed of 8 subjects with limited cutaneous SSc and 3 with diffuse disease. Three patients were classified as early SSc disease (< 4 years). Subjects had mean age of 54.5±10.3 years and body mass index of 22.0±2.1 kg/m 2 . As part of the inclusion criteria all subjects had hand disability of at least 20 points using the Cochin Hand Function Scale (CHFS).

Table 1 :

 1 Outcome of assessed parameters from baseline to last complete

Figure 1 :

 1 Figure 1: Evolution of the 4 main parameters after 2 years of the procedure

All authors participated in the writing of the manuscript and the interpretation of the data. They all approved the manuscript.

Acknowledgements:

We thank the French Scleroderma Research Group (GFRS), the Centre d'Investigation Clinique (CIC) for having welcomed the patients, Joelle Micallef and Estelle Charles-Baumel from the Centre d'Investigation Clinique -Centre de Pharmacologie Clinique et d'Evaluations Thérapeutiques (CIC-CPCET).

Funding: We acknowledge support from Cytori for having given free devices and consumables for ADSVF surgical procedure.

Disclosure statement: Prof Magalon has received honoraria for consulting services for Cytori Therapeutics. The others authors declared no conflicts of interest.

KEYWORDS: systemic sclerosis, stromal vascular fraction, adipose tissue


Audrey Benyamine
email: audrey.benyamine@ap-hm.fr

Daniel Bertin

Pharm.D Xavier Heim

Pharm.D Brigitte Granel

Nathalie Bardin

Should we look for anti-RNA polymerase III antibodies in Systemic Sclerosis patients with anti-centromere or anti-topoisomerase I antibodies? Running title: anti-RNA polymerase III in Systemic Sclerosis

Keywords: Systemic Sclerosis, Anti-RNA Polymerase III antibodies, Anti-centromere Antibodies, Anti-topoisomerase I antibodies, Immunofluorescence pattern

Systemic sclerosis (SSc) is a chronic autoimmune disease characterised by skin and internal organs fibrosis, vascular damage and positive antinuclear autoantibodies (ANAs). The classical antigenic specificities of ANAs are anti-centromere (ACA), anti-topoisomerase I antibodies (anti-topo I) and the more recently described anti-RNA polymerase III antibodies (anti-RNAPIII).

Anti-RNAPIII, firstly described in SSc in 1993 [START_REF] Kuwana | Autoantibody reactive with three classes of RNA polymerases in sera from patients with systemic sclerosis[END_REF], are currently admitted as specific SScrelated autoantibodies and have been incorporated into the 2013 ACR/EULAR classification criteria for the disease [START_REF] Van Den Hoogen | classification criteria for systemic sclerosis: an American College of Rheumatology/European League against Rheumatism collaborative initiative[END_REF]. The prevalence of anti-RNAPIII partially depends on the geographic origin of patients [START_REF] Sobanski | Prevalence of anti-RNA polymerase III antibodies in systemic sclerosis: New data from a French cohort and a systematic review and meta-analysis[END_REF]. In France, the prevalence is low and ranges from 3 to 9% whereas it can reach 14% in North America and 41% in South America [START_REF] Sobanski | Prevalence of anti-RNA polymerase III antibodies in systemic sclerosis: New data from a French cohort and a systematic review and meta-analysis[END_REF][START_REF] Faucher | Low prevalence of anti-RNA polymerase III antibodies in a French scleroderma population: anti-RNA polymerase III scleroderma[END_REF]. The ANA pattern associated with anti-RNAPIII was described as a fine-speckled nuclear stain with additional occasional bright dots, with or without concurrent punctate nucleolar staining but no typical pattern was proposed [START_REF] Parker | Anti-RNA polymerase III antibodies in patients with systemic sclerosis detected by indirect immunofluorescence and ELISA[END_REF]. Anti-RNAPIII are mainly associated with a diffuse cutaneous subtype, scleroderma renal crisis and a risk of cancer in close temporal relationship to SSc onset [START_REF] Sobanski | Prevalence of anti-RNA polymerase III antibodies in systemic sclerosis: New data from a French cohort and a systematic review and meta-analysis[END_REF][START_REF] Faucher | Low prevalence of anti-RNA polymerase III antibodies in a French scleroderma population: anti-RNA polymerase III scleroderma[END_REF][START_REF] Emilie | Anti-RNA polymerase III antibodies are associated with scleroderma renal crisis in a French cohort[END_REF][START_REF] Meyer | Anti-RNA polymerase III antibody prevalence and associated clinical manifestations in a large series of French patients with systemic sclerosis: a cross-sectional study[END_REF]. As the specific method for their identification was the radioimmunoprecipitation assay, a cumbersome method not suitable for routine practice, these autoantibodies were not routinely looked for. More recently immunoenzymatic methods have been developed, but, the detection strategy was not clearly established for routine practice [START_REF] Parker | Anti-RNA polymerase III antibodies in patients with systemic sclerosis detected by indirect immunofluorescence and ELISA[END_REF].

Although the positivity of ACA and anti-topo I is considered to be exclusive, the co-positivity of anti-RNAPIII with these SSc-specific auto-antibodies remains to be clarified. In order to design the best strategy for anti-RNAPIII detection, we aimed to 1/ evaluate the co-positivity of anti-RNAPIII in SSc patients positive for ACA or anti-topo I and to 2/ analyse immunofluorescence patterns and clinical characteristics of anti-RNAPIII positive patients.

Firstly, 76 sera from SSc patients (9 men, 67 women) from Marseilles (South of France) positive for either ACA or anti-topo I antibodies were tested for the presence of anti-RNAPIII. All patients fulfilled the 2013 ACR/EULAR criteria and were further classified as having diffuse or limited cutaneous SSc [START_REF] Van Den Hoogen | classification criteria for systemic sclerosis: an American College of Rheumatology/European League against Rheumatism collaborative initiative[END_REF]. All the sera were collected from 2012 to 2016 and were issued from a Biobank (DC 2012-1704) with respect of ethical directives.

Antinuclear antibodies (ANAs) were detected by indirect immunofluorescence on HEp-2 cells (Bio-Rad Laboratories, Hercules, CA) at a screening dilution of 1:160. ACA, anti-topo I and anti-RNAPIII were detected by commercially kits (EliA Thermo Fisher). The cutoff value for anti-RNAPIII positivity was 10 Arbitrary Unit/ml (AU).

Secondly, the ANAs immunofluorescence patterns and clinical data of 8 SSc patients positive for anti-RNAPIII were collected from 2012 to 2016 and compared to anti-RNAPIII negative SSc patients (<10 AU/ml). Results were expressed as median +/-interquartile range or as frequencies (fq). Medians were compared using Mann Whitney U Test. Frequencies were compared using Chi 2 Test. Among the 76 selected sera of SSc patients, 33 patients (43%) were positive for ACA and 43 (57%) for anti-topo I antibodies. Immunofluorescence nuclear patterns were: centromeric (n=32), nucleolar homogeneous (n=41), speckled-centromeric (n=1), speckled-homogeneous (n=1) and mixed speckled-nucleolar-centromeric (n=1). Anti-RNAPIII were investigated in these SSc patients: only one ACA-positive serum (1.3%) was found also positive for anti-RNAPIII with a titer of 192 AU/ml. This serum corresponded to the mixed specklednucleolar-centromeric immunofluorescence nuclear pattern (Figure 1). No other specificities associated with a speckled pattern (anti-Ro/SSA anti-La/SSB, anti-Sm, anti-RNP) were detected. The patient, a 47-year-old female, had a diffuse cutaneous subset, with digital ulcers, joint contractures, a reduced diffuse lung capacity for carbon monoxide and oesophageal reflux disorder. Her medical history was remarkable for an ovarian adenocarcinoma that was diagnosed 6 years before SSc, and considered in remission after surgery and radio-chemotherapy.

Then, we retrospectively analysed immunofluorescence pattern and clinical characteristics of 8 SSc patients found positive for anti-RNAPIII in our laboratory. The various immunofluorescence aspects were nuclear speckled (n=5), nucleolar (n=1), nucleolarspeckled (n=1) (Figure 2) and mixed speckled-nucleolar-centromeric (n=1) (Figure 1). The median titer of anti-RNAPIII was 54 [18-298] AU/ml. Table 1 illustrates the comparison between anti-RNAPIII positive and negative patients. Sex ratio did not differ between the two groups. A trend to a higher frequency of the diffuse cutaneous form was observed in anti-RNAPIII positive SSc patients. The two patients with sine scleroderma belonged to the group of negative anti-RNAPIII. Scleroderma renal crisis was solely documented in patients with positive anti-RNAPIII. Other variables were not significantly associated with anti-RNAPIII positivity.

The present study extends the previously published data performed in South of France [START_REF] Faucher | Low prevalence of anti-RNA polymerase III antibodies in a French scleroderma population: anti-RNA polymerase III scleroderma[END_REF] about the anti-RNAPIII screening strategy. This study highlights that anti-RNAPIII are rarely encountered in SSc patients already positive for ACA or anti-topo I antibodies. The case of the patient with coexisting ACA and anti-RNAPIII was interesting regarding two aspects.

First, the clinical feature was closer to the one described in patients with anti-RNAPIII with a diffuse cutaneous form and a concomitant cancer [START_REF] Lazzaroni | Malignancies in Patients with Anti-RNA Polymerase III Antibodies and Systemic Sclerosis: Analysis of the EULAR Scleroderma Trials and Research Cohort and Possible Recommendations for Screening[END_REF]. Second, the observed ANAs immunofluorescence pattern was highly remarkable due to the peculiar aspect of immunofluorescence. Therefore, in ACA or anti-topo I positive sera, the search for anti-RNAPIII can be recommended faced to a mixed fluorescence pattern. Conversely, in case of typical fluorescence aspects related to ACA or anti-topo I positivity, the systematic search for anti-RNAPIII is not mandatory. As observed herein, different immunofluorescence pattern can be associated with anti-RNAPIII [START_REF] Codullo | Validation of a new immunoenzymatic method to detect antibodies to RNA polymerase III in systemic sclerosis[END_REF]. Therefore the search for this auto-antibody should not be restrained to a nucleolar immunofluorescence pattern of ANAs [START_REF] Yamasaki | Nucleolar staining cannot be used as a screening test for the scleroderma marker anti-RNA polymerase I/III antibodies[END_REF].

Regarding the phenotype of anti-RNAPIII positive patients, a higher frequency of scleroderma renal crisis and diffuse cutaneous form was observed. The low number of patients and our geographic location might be a limiting factor to evidence any association with other anti-RNAPIII features such as the frequency of cancer [START_REF] Lazzaroni | Malignancies in Patients with Anti-RNA Polymerase III Antibodies and Systemic Sclerosis: Analysis of the EULAR Scleroderma Trials and Research Cohort and Possible Recommendations for Screening[END_REF].

Although SSc-related autoantibodies are exclusive markers, co-positivity can exist in rare cases. The immunofluorescence reading step appears crucial to focus the screening. In order to improve the benefit cost ratio, anti-RNAPIII should be searched with respect to ANAs fluorescence pattern and clinical characteristics of the patients.

Legends of the figures

Figure 1 :

 1 Figure 1: The mixed fluorescence pattern (speckled-nucleolar-centromeric) of the ACA and

Figure 2 :

 2 Figure 2: Various immunofluorescence patterns obtained from sera positive for anti-RNAPIII

ACKNOWLEDGEMENT

This research did not receive any specific grant from funding agencies in the public, commercial, or not-for-profit sectors.

We thank internal medicine physicians for their help in collecting the clinical data (M.Ebbo, N.Schleinitz, G.Kaplanski, C.Gomez).