Design of an energy detector for heartbeat localization in ECG signals

Z. Benjelloun, R. Vauche, W. Rahajandraibe, R.

Abstract-Cardiovascular Diseases (CVD) causes as many deaths as cancer in Europe. A method to detect them consists in tracking the heartbeat rate from ECG signal. In this case, QRS complexes have to be localized and it is proposed here to use an energy detector to allow this. It can be integrated in an analog circuits and allows low power and low cost applications to be addressed such as consumer embedded electronics. Using 0.35µm CMOS technology from AMS, the design of the analog multiplier cell required by the energy detector is also presented here. It shows a voltage gain upper than 18 with enough linearity for the targeted applications. Preliminary results of the full energy detector which mix Eldo and Matlab simulations show an accuracy upper than 99% for SNR upper than 0dB. Index Terms-Electrocardiography, Energy Detector, Heart Rate Variability, Multiplier, QRS detection.

I. INTRODUCTION

The Heart Rate Variability (HRV) represents the variation of the time intervals between each heartbeat. HRV is normally low for healthy people and increases with the risk of cardiac events such as fibrillation, arrhythmia, and Premature Ventricular Contraction (PVC) equally known as ventricular extrasystole. To know the moments when a heartbeat occurs, QRS complexes can be extracted from an Electrocardiogram (ECG) recording. These QRS complexes, synchronous with heartbeats and which correspond to the depolarization of the right and left ventricles of the human heart, are generally the most visible pulses in a clean ECG signal between P-waves and T-waves as indicated in Fig. 1.

The time intervals between QRS complexes, and also heartbeats, are known in medical papers as RR signal when they are plotted in time domain. Then, to estimate correctly HRV, it is necessary to know this RR signal with the best possible accuracy which is only possible when all QRS complexes are detected [START_REF] Bernston | ECG artifacts and heart period variability: Don't miss a beat![END_REF]. However, ECG recordings may contain distorted episodes due to various extrinsic sources of interferences such as electrical activity produced by skeletal muscles, power-line, motion or electrode artifacts, etc. Moreover, as indicated in [START_REF] Arzeno | Analysis of First-Derivative Based QRS Detection Algorithms[END_REF], other intrinsic reasons can break QRS detection such as wide PVCs which occur before regular heartbeats, high-amplitude of T-waves, low-amplitude QRS complexes, negative QRS complexes, low Signal-to-Noise Ratio (SNR), etc.

To detect QRS complexes as well as possible, several algorithms have been designed. A comparative study of these algorithms has been done in [START_REF] Bernston | ECG artifacts and heart period variability: Don't miss a beat![END_REF] and shows that firstderivative-based methods used in real-time analysis have the best accuracy. The most popular of these QRS detection methods is the Hamilton-Tompkins algorithm [START_REF] Hamilton | Quantitative Investigation of QRS Detection Rules Using the MIT/BIH Arrhythmia Database[END_REF], which is an improvement of the Pan and Tompkins algorithm proposed in [START_REF] Pan | A Real-Time QRS Detection Algorithm[END_REF]. This algorithm uses a specific adaptive threshold after filtering, differentiation, squaring and integration of the ECG signal. Other algorithms exist as the ones based on Hilbert transform. An example of this type of algorithm is proposed in [START_REF] Benitez | A new QRS detection algorithm based on the Hilbert transform[END_REF] and uses a threshold which does not need to be experimentally fixed. Finally, Discrete Wavelet Transform (DWT) can be used for QRS detection as proposed in [START_REF] Szilagyi | Wavelet transform and neural-networkbased adaptive filtering for QRS detection[END_REF] and is based on a multi-resolution sup-band decomposition process and an adaptive threshold depending on the output signal of the moving window integrator.

All the methods mentioned previously are digital solutions which are easily implementable on a microcontroller or digital signal processor. However, even if the used algorithm can be easily changed in this type of systems, they integrate lots of functions which consume power and silicon area. Thus, to address low cost and low power applications such as consumer embedded electronics, it is proposed here to design an analog Application Specific Integrated Circuit (ASIC) dedicated to the QRS complexes detection. Moreover, since QRS complexes are similar to the pulses used in IR-UWB in terms of waveforms, a solution inspired from previous works on non-coherent Ultra-Wide-Band Impulse Radio (IR-UWB) receivers [START_REF] Sparrow | High rate UWB CMOS transceiver chipset for WBAN and biomedical applications[END_REF] is proposed here.

In IR-UWB, it is possible to find in literature two families of non-coherent receivers which are known for their simplicity; the first is based on peak voltage detection and the second one on energy detection. In [START_REF] Sparrow | High rate UWB CMOS transceiver chipset for WBAN and biomedical applications[END_REF], a non-coherent IR-UWB receiver based on energy detection is presented. It consists in a Low Noise Amplifier (LNA), a squarer, an envelope detector and a decision stage. This structure is here the starting point of the presented study and the aim of this paper is also to provide the necessary elements to transform this UWB pulse detector to a QRS detector, and to provide a squarer compatible with QRS detection. In this paper, to adapt the structure presented in [START_REF] Sparrow | High rate UWB CMOS transceiver chipset for WBAN and biomedical applications[END_REF] to QRS complexes detection, a preliminary study on QRS spectrum is done in Section II before the sizing of the detection chain. Next, Section III focuses on the design of the squarer which is the key element of the proposed QRS detector since it has to work on a sufficient input voltage range and consumes as few as possible. Section IV presents the first simulations results of the full chain where Eldo simulation results of the squarer have been mixed to Matlab computing. Finally, conclusions and prospects are presented in Section V.

II. ARCHITECTURE OF THE QRS COMPLEXES DETECTOR

As seen previously, heartbeat rate can be obtained by computing the time interval between two consecutive QRS complexes. An efficient real time processing is also needed to detect them even if the ECG signal is distorted. As low power and low cost applications are targeted, the processing must be achievable in an analog ASIC and it is proposed here to adapt the IR-UWB pulse receiver presented in [START_REF] Sparrow | High rate UWB CMOS transceiver chipset for WBAN and biomedical applications[END_REF]. For these reasons, QRS complexes are characterized in frequency domain which allows the processing chain to be sized.

A. Characterization of QRS Complexes in Frequency Domain

The relative power spectrum averaged on 150 beats of QRS complexes and its interference sources are provided in [START_REF] Valtino | ECG QRS detection[END_REF] and shown in Fig. 2. It is possible to see that the central frequency of QRS complex is about 10Hz and its -10dB bandwidth to 10Hz. To confirm that, the distribution of the central frequency of QRS complexes has been studied with the help of the public ECG database Physionet. Results are given in Fig. 3 and shows that the central frequency is uniformly spread in the 8-12Hz range. It is also confirmed that the typical central frequency of QRS complexes is 10Hz.

B. QRS detectors processing chain

QRS detection algorithm typically includes four signal processing steps: (i) a bandpass filter, (ii) a nonlinear transformation, (iii) a moving window integrator and (iv) a decision rule algorithm. The architecture of the IR-UWB noncoherent receiver based on energy detection presented in [START_REF] Sparrow | High rate UWB CMOS transceiver chipset for WBAN and biomedical applications[END_REF] has a similar approach where, the non-linear transformation is a squaring function and the moving window integrator is replaced by a low-pass filter (which allows envelope of squared pulses to be extracted too). As QRS complexes and IR-UWB pulses have closed waveforms, the QRS detection chain based on energy detection proposed here consists in a bandpass filter (a), a squarer (b), an integrator (c) and an adaptive threshold comparator (d) as shown in Fig. 4. The Matlab simulation results of this structure are given in Fig. 5.

The choice of characteristics of the bandpass filter (a) is a critical step in the processing chain design since it should eradicate the noise interfering with the targeted signal without filtering it. From results of the previous study, it has been chosen with a -10dB bandwidth of 8Hz centered on 10Hz. It allows then the baseline (DC block) and the main (50Hz or 60Hz) to be filtered. Moreover, noise from skeletal muscles, T-waves and motion artifacts are reduced too since their power spectrum is concentrated below 5Hz as shown in Fig. 2. Finally, as this stage works as an LNA, it should have a low noise figure especially to minimize 1/f noise and a sufficient gain to get a peak signal arbitrary fixed to 100mV to obtain usable signals after squaring operation.

As the proposed QRS complexes detector is based on an energy detector, the second block is a squarer (b). This allows to stronger signals to be more amplified than weak signal. However, although this operator is easy to implement in a digital system, it is more difficult to integrate in an analog ASIC when a large signal has to be squared. For these reasons, the design of a low power and low cost multiplier is proposed in the following section. It is now necessary to extract envelope of the squared signal which is done here by a sliding window integrator (c).

The key parameter to adjust in this block is the window width. If the window is too wide, two QRS complexes can be interpreted as only one and if the window is too narrow, there is a risk of missing the QRS peaks. From experiments, the width has been chosen equal to 150ms which correspond to the double of the founded average QRS width.

Finally, an adaptive threshold comparator (d) has to be used to keep only the larger pulses normally due to QRS complexes. This threshold is fixed equal to 60% of the maximum value of the input signal during the last 5s as proposed in [START_REF] Valtino | ECG QRS detection[END_REF]. Moreover, to filter pulses due to interferences, this adaptive threshold comparator is not retriggerable during 366ms since the maximum heartbeat is equal to 220bpm for a human. To improve accuracy of this block, a QRS complex is considered detected when its input is higher than the threshold, and when its first derivative is equal to 0 which indicates a local maximum.

III. SQUARER DESIGN

In the case of an analog ASIC, a squaring stage can be obtained by using an analog voltage multiplier. This multiplier has to perform a linear products between its two input voltages 1 ,v 2 ), and also provides on its output a voltage which can be expressed as K.v 1 .v 2 , where K is a constant gain factor. The evaluation of this multiplier has also to include linearity, power consumption, and especially input range and gain. This is due to the fact that stronger signals are be more amplified than weak signal. In this section, a differential multiplier is designed and integrated in an unbalanced squarer using a 0.35µm CMOS design kit from AMS.

A. Differential Multiplier Design

In [START_REF] Han | CMOS transconductance multipliers: a tutorial[END_REF], a comparison between different CMOS transconductance multipliers structures is done. It appears that they can be categorized into two groups below MOS operation region: linear and saturation region. In this paper, the designed multiplier operates in saturation region in order to obtain sufficient input range and gain according to applications requirements. The proposed design shown in Fig. 6 is based on a Gilbert mixer with balanced inputs (V1 and V2) and unbalanced output (VOP). To drive correctly the multiplier, balanced inputs VxP and VxN have to be symmetrical with respect to VDD/2. Finally, it can be turned-off when it is necessary by disabling the current source since PDB is a logic signal equal to '0' (GND) or '1' (VDD).

B. Integration in an Unbalanced Squarer

To drive the proposed multiplier with an unbalanced filtered ECG signal, it is necessary to generate symmetrical inputs signals with respect to VDD/2. A solution based on operational amplifier is proposed in Fig. 7 and allows symmetrical filtered ECG signals to be obtained with respect to VOFFSET. It consists in a voltage follower connected to two substractors, the first being used to provide high impedance to ECG source since the equivalent input impedance of the both connected substractors is R. To limit the power consumption of operational amplifiers and make a compromise with silicon area, R has been fixed to 100kΩ.

A. Multiplier characterization

To characterize the multiplier, the common mode voltage of V1 and V2 which is equal to (VxP+VxN)/2 is set to VDD/2 (1.65V). The differential mode voltage of V1 (resp. V2) which is equal to (VxP-VxN)/2 is called v1 (resp. v2). The results of the multiplier DC analysis for an input range of [-200mV; 200mV] is given in Fig. 9. It appears that (VOP -VDD/2) can be identified to K.v1.v2 where K is equal to 18.3 (resp. 18.5) with a standard derivation of 0.67 (resp. 1.03) for [-100mV; 100mV] (resp. [-200mV; 200mV]) vx voltage range. Finally, its power consumption is 0.94mW when it is turned-on (PDB = VDD) and about 17pW when it is turned-off (PDB = GND).

B. Squarer simulation

To simulate the presented energy detector in the worst case, an ECG signal containing many irregularities from Physionet database has been used. The blocks (a), (c) and (d) which can be seen on Fig. 4 have been simulated with Matlab and the block (b) has been post-layout simulated with Eldo. The gain of the bandpass filter (a) has been tuned to have a maximum peak voltage of 100mV on its output. Simulation results of the processing chain are shown on Fig. 10. They show the filtered ECG voltage B applied to the squarer and the voltage C obtained before the integration stage, the both for an infinite (a) and 0dB (b) SNR using an Additive White Gaussian Noise (AWGN) to the ECG input voltage. Finally, simulations of the full energy detector show that 99.2% (resp. 100%) of QRS peaks are well detected for a 0 dB (resp. > 3dB) SNR.

V. CONCLUSION To extract heartbeat rate from ECG signal, a QRS detector has been proposed. It is based on energy detection and requires also a squarer function. To implement it in a low cost and low power analog ASIC, an analog multiplier cell has been designed and post-layout simulated with Eldo. It shows a voltage gain upper than 18 with enough linearity. To estimate performances of the full processing chain, the other required functions have been simulated with Matlab. Preliminary results which mix Eldo and Matlab simulations show a good accuracy for SNR upper than 0dB, which allows the use of the presented QRS detector to be expected for cardiac monitoring [START_REF] Lynn | Arm and writ surface potential mapping for wearable ECG rhythm recording devices: a pilot clinical study[END_REF] or other physiological parameters by modifying characteristics (frequency band, etc.) of the energy detector.

Fig. 1 .

 1 Fig. 1. Waveform of a QRS complex.

Fig. 2 .Fig. 3 .

 23 Fig.2. Relative power spectra of QRS complex, P and T waves, muscle noise and motion artifacts based on an average of 150 beats[START_REF] Valtino | ECG QRS detection[END_REF].

Fig. 4 .

 4 Fig. 4. Block diagram of the processing chain.

Fig. 5 .

 5 Fig. 5. Illustration of the different processing steps.

Fig. 6 .

 6 Fig. 6. Design of the presented balanced multiplier.

Fig. 7 .Fig. 8 .

 78 Fig. 7. Integration of the balanced multiplier in a unbalanced squarer. IV. SIMULATION RESULTS The presented energy detector for heartbeat localization in ECG signals has been post-layout simulated using a 0.35µm CMOS design kit from AMS with a supply voltage of 3.3V. The operational amplifiers required by the squarer are taken from AMS IP library. The layouts of the designed multiplier and of the full squarer are shown in Fig. 8 and respectively have a core area of 20µm by 31µm and 1mm by 0.9mm. The following part gives multiplier characteristics. Next, examples of energy detector output signal obtained by Cadence and Matlab mixed simulations are provided.

Fig. 9 .

 9 Fig. 9. DC analysis of the multiplier: VOP vs (v1, v2).

Fig. 10 .

 10 Fig. 10. Simulations results of the filtered ECG voltage B applied to the squarer and the voltage C obtained before the integration stage for an infinite (a) and 0dB (b) SNR.

ACKNOWLEDGMENT This work, which is part of the APPRISE/HIT project, was performed with the support of the A*MIDEX project (n° ANR-11-IDEX-0001-02) funded by the «Investissements d'Avenir» program of the French Government, which is managed by the French National Research Agency (ANR).


Abstract-This paper presents some techniques to synthesize pulses in the context of IR-UWB. Pulse synthesizers are good candidates when different kinds of pulses are needed to comply with different standards or when the shape of the pulses needs to be modified. Two different approaches are described here. Voltage Controlled Delay Line (VCDL) based pulsers are first described. For this family, two kinds of combiners are presented : differential pairs based and H-bridge based. The second one achieves a larger output dynamic but is limited by the use of the VCDL. A second type of H-bridge based synthesizer using VCO is presented. Using both a VCO and a H-Bridge offers the largest synthesis capabilities. It allows large voltage pulses to be synthesized and the pulse duration is not limited. All proposed architectures has been implemented and their measured performances are given for comparison.

Index Terms-Pulse generation, FCC, Transmitters, Impulse Radio, Ultra-wideband (UWB), Pulse shaping circuit.

I. INTRODUCTION

ommunication in the UWB band is regulated by the FCC since it has been authorized in 2002 in the 3.1GHz to 10.6GHz frequency band. Several standards have emerged such as the 802.15.4a for PAN applications, the 802.15.4f for asset tracking or the 802.15.6 for WBAN. Dedicated systems have also been developed for localization or biomedical applications but most of these applications needs low power communication systems to operate.

A general trend in communications systems is the convergence of several applications in an unique device. This allows cost reductions and increase the functionality of the devices. To reduce power consumption, functions must be shared by several sub-systems. To do so, these functions must be reconfigurable to meet various requirements. This is especially a bottleneck for RF devices which are often based on passive elements which are not tunable such as inductors. In IR-UWB, reconfigurable front-end means to be able to generate and detect pulses having different shapes, durations and central frequencies. The very wide bandwidth of the systems can be an advantage, especially on the receiver side. Non-coherent receivers are able to detect a large variety of pulses in different frequency bands. By varying the integration time of the detector; Non-Coherent receivers can detect pulses with different bandwidths and shapes. In coherent receivers, since the correlation is little affected by the pulse shape [START_REF] Vereecken | An I/Q based CMOS Pulsed Ultra Wideband Receiver Front End for the 3.1 to 10.6 GHz Band[END_REF], the most critical point is to control the central frequency and the width of the local pulse used for correlating the received signal. However, this ability is inherent of a correlation receiver and is not a bottleneck. On the transmitter side, the generation of a large variety of pulses is more difficult. The basic parameters to be controlled are the central frequency and the pulse width. To do so, LO switching techniques can be used. It allows the width and the central frequency of the pulse to be controlled. However, adjusting the pulse shape is more difficult whereas it highly affects the spectrum and especially the side lobs which reduces the Adjacent Chanel Power Ratio (ACPR). Several solutions have been proposed to shape the pulse by modulating the pulse magnitude with a shape function [START_REF] Dokania | An Ultralow-Power Dual-Band UWB Impulse Radio[END_REF]. These techniques suffer from a lake of tuning abilities since they are designed to achieve a given pulse shape. Moreover, the performances of the modulator depend on the operating frequency and it is difficult to maintain a constant shape over a very wide frequency band of operation. To overcome this drawback, pulse synthesizing techniques have emerged [START_REF] Bourdel | Low-Cost CMOS Pulse Generator for UWB Systems[END_REF]. By using a pulse synthesizer, it is possible to change the shape of a pulse to comply with different standards which have different requirement regarding the pulse spectrum and the ACPR as shown in Fig. 1. The principle of pulse synthesizing is to control the width (p) and the magnitude (gn) of each elementary pulse (en(t)) that makes the emitted pulse s(t) has given by (1) and represented in Fig. 3.

1 1 0 1 ( ) . N n n n p n p s t g e t  - - = =   = -       (1)
In this paper we present several implementations of pulse synthesizers and we discuss their performances and their limitations. In the first section, synthesizers based on Voltage Controlled Delay Line (VCDL) are presented. In this family of synthesizer, the maximum number (N) of elementary pulses is fixed. This limitation has encouraged the emergence of VCO based pulse synthesizer especially to address lower bandwidths which require a large number of elementary pulses, especially in high frequency bands. These synthesizers will be discussed in section III.

II. VCDL BASED PULSE SYNTHESIZER

The first reported work on pulse combining was based on fixed magnitude and width elementary pulse [START_REF] Kim | All-digital low-power CMOS pulse generator for UWB system[END_REF]. This preliminary work paves the road to the pulse synthesizing technique.

A. Differential pulse combiner based synthesizer

Based on pulse combining techniques, a pulse synthesizer has been proposed [START_REF] Bourdel1 | An Inductorless CMOS UWB Pulse Generator with Active Pulse Shaping Circuit[END_REF]. The synthesizer uses a Digital Delay Line (DDL) to propagate a rising edge. The outputs of the delay line are combined to produce delayed complementary square pulses as shown in Fig. 4. The width of the n th digital pulse can be modified by changing the propagation delay in the n th inverter. The pulses are then combining using current summing in N differential pairs. The outputs of the differential pairs are alternately crossed to achieve positive and negative combination of the elementary pulses. The magnitudes of the elementary pulses are fixed by the DC current in the differential pair. The pulses are then amplified and matched to the antenna with a common source amplifier.

A seven stages synthesizer has been implemented in 130nm CMOS technology. The circuit is represented in Fig. 10.a). The active area is only 0,06mm 2 , it consumes 87.6mW@800MHz and it can achieve several pulse shapes as shown in Fig. 2. This first work demonstrates the possibility of synthesizing different pulse shape but suffer from many drawbacks. The DC power consumption is quite heavy due to the use of differential pair. The performances of the output amplifier highly limit the overall performances. Its limited bandwidth produces distortion and affects the pulse shape. This can be compensated with predistortion methods but the tuning of the synthesizer is not straightforward.

B. H-Bridge pulse combiner based synthesizer

To overcome the limitation of the differential pulse combiner, an H-Bridge is proposed to combine the elementary pulses [START_REF] Vauche | Fully tunable UWB pulse generator with zero DC power consumption[END_REF]. The principle is presented in Fig. 5. The n th elementary pulse (EPn) is used to control the n th branch of the bridge. The sign of the current flowing into the charge determines the polarity of the n th elementary pulse and the voltage applied to the n th branch set the magnitude. The same pulse digital generator is used.

An eight stages synthesizer has been implemented in 130nm CMOS technology. The circuit is represented in Fig. 10.b). The core size is 700um*400um. Thanks to the H-Bridge architecture it consumes only 250uW@1MHz and 100uW@0Hz. It can achieves several pulse shapes as shown in Fig. 10 and several frequency bands can be addressed by varying the propagation delay in the VCDL. The maximum output magnitude is 1.8V (resp. 0.9V) when pulses centered on 2.5GHz (resp. 5GHz) are generated. However this structure is limited by the size of interconnection line required by the H-Bridge output because its length grows with the number of stages which is set by the number of elementary pulses to generate. For the high frequency band, the number of elementary pulses needed to generate a 500MHz pulse can be greater than 60 which makes this architecture unsuitable for the high band of the IEEE 802.15.4a and 802.15.6 standards.

III. VCO BASED PULSE SYNTHESIZER

To overcome the limitations of the VCDL based architectures when a high number of elementary pulses is needed, the implementation of a VCO based pulse synthesizers is necessary since it relaxes constraints on the size of the delay line and on the number of the required branches inside the output combining stage. However, this requires at first sight a magnitude control of each elementary pulse. Fortunately, based on an analytical analysis of UWB pulses in time and in frequency domains, it is demonstrated that a discrete symmetrical Gaussian envelope with only 5 samples (N=5) is enough to respect all IEEE frequency masks of 500MHz channels if a margin around 2dB is considered for ACPR [START_REF] Muhr | Design of VCO based CMOS pulse generator for IR-UWB applications[END_REF]. Moreover, as the pulse bandwidth is independent on the central frequency of the targeted channel, it is only necessary to synchronise the sampled envelope generator and the VCO with a programmable frequency divider where the division factor is set according to the targeted channels.

An implementation of this circuit is proposed on Fig. 7. A VCO generates a sinusoidal waveform whose the frequency is equal to the central frequency of the targeted channel. A sampled envelope is synchronously generated according this channel, each sample has also a length multiple of the sinus half-period. Finally, an H-bridge is used to mix the sinusoidal waveform with the envelope.

To be able to address each channel of IEEE standards, the proposed architecture requires also a VCO which is able to cover all the 3.1GHz-10.6GHz band. Due to the width of the frequency band to address, it is usual to implement two oscillators, one for low band channels and another one for high band channels. However, we demonstrate in [START_REF] Vauche | Ultra-WideBand Voltage Controlled Oscillator with Commutable Phases for BPSK Implementation[END_REF] that only one VCO designed with a standard 65nm CMOS technology can cover all channels. It is based on a three stages ring oscillator architecture where Current Mode Logic (CML) gates are used instead of usual CMOS gates. The oscillations frequency is also controlled by adjusting the gate source voltages of their active loads and can be started and stopped quasi instantaneously which allow OOK modulation to be obtained. Moreover, it can be put in a low power mode when oscillations are stopped, and is able to directly act as a BPSK modulator. To keep simple the control of UWB pulses shape, a sampled envelope generator, which allows each step to be tuned around their nominal values, is proposed here. To set these values, an analytical model of UWB pulses as the Gaussian up-converted pulse introduced in [START_REF] Sparrow | High rate UWB CMOS transceiver chipset for WBAN and biomedical applications[END_REF] pulse can be used after considering the required bandwidth. However, a degree of freedom on these values must be kept in order to ensure the respect of the targeted frequency masks in case of process, voltage or temperature variations. The proposed implementation is shown in Fig. 8 and used CMOS transmission gates assembled in step tuners to supply a standard H-bridge. As the Gaussian up-converted pulse is symmetrical, only 3 step tuners are required. However, an additional step tuner, which can be turned-off, has been implemented to have a better envelope control abilities. Digital control signals required by these transmission gates are set by a custom serial interface and enable by the frequency divider. Finally, particular values for step magnitude are fixed by adjusting the size of the transistors of the step tuner.

A prototype of the architecture has been implemented in 65nm CMOS technology from STMicroelectronics. The circuit is represented in Fig. 10.c). Its core area is 220µm x 250µm and its supply voltage is 1.2V. The VCO based pulse synthesizer has been measured and it shows a peak to peak voltage of 1.1V for low band IEEE channels in the worst case. In this case, its power consumption is between 600µW and 700µW@1MHz, depending on the configured envelope, when a power gating cycle (turns-on, generates the pulse, and turnsoff) of 10ns. Finally, two different pulses for channels 2 and 4 of IEEE 802.15.4 standard are shown in Fig. 9.

IV. CONCLUSION

Three pulse synthesizers for IR applications have been presented. The first one uses VCDL and differential pair based combiner. The combiner limits the output dynamic of this structure and the H-Bridge based synthesizer has been proposed. The H-Bridge allows high voltage pulses up to 1,9V to be synthesized but the VCDL avoid the generation of pulses having 500MHz bandwidth. A H-bridge pulse synthesizer using a VCO instead of the VCDL has been proposed to overcome this limitation.

Fig. 3 .Fig. 2 .

 32 Fig. 3. Principle of pulse synthesizing

Fig. 6 .

 6 Fig. 6. Synthesized pulse shape with H-Bridge based synthesizer

Fig. 7 .

 7 Fig. 7. Principle of the proposed VCO based pulse synthesizer

Fig. 10 .Fig. 9 .

 109 Fig. 10. Fabricated Pulse Synthesizer: a) differential pair based, b) VCDL and H-bridge Based, c) VCO and H-bridge based.

  Aix Marseille Université, CNRS, Université de Toulon, IM2NP UMR 7334 Marseille, France **Université Grenoble Alpes, F-38000 Grenoble, France, CEA, LETI, MINATEC Campus, F-38054 Grenoble, France ***Université Grenoble-Alpes, F-38000 Grenoble, France, IMEP-LAHC, Minatec Campus, 3 Parvis Louis Neel, F-38016 Grenoble, France

Design of Pulse Synthesizers for the Convergence of IR-UWB Solutions R. Vauché*, S. Bourdel***, E. Muhr*, Member, IEEE, J. Gaubert*, N. Dehaese*, F. Hameau**, and H. Barthelemy*, Member, IEEE * C Fig. 1. Pulse Shaping for 802.15.4a and 802.15.6


Abderrahmane Haloua
email: abderrahmane.haloua@im2np.fr

Rémy Vauche
email: rémy.vauché@im2np.fr

Nicolas Dehaese
email: nicolas.dehaese@im2np.fr

Jean Gaubert
email: jean.gaubert@im2np.fr

A Fully Digital Ultra-wide Band Sub-GHz Pulse Generator

Keywords: CMOS, UWB, Fully digital, sub-GHz, pulse generator I

The design of a fully digital tunable sub-GHz Ultra-wide band (UWB) pulse generator is presented. The pulse generator has been implemented in a 0.13µm CMOS technology. Post-layout simulation shows that the proposed pulse generator achieves UWB pulses with a 900mVpp magnitude while consuming 27pJ by pulse under 1.2V supply voltage. The tuning capabilities allow the center frequency setting between 400MHz and 610MHz.

INTRODUCTION

Ultra-Wide Band (UWB) communications are regulated by the Federal Communications Commission (FCC) Since 2002 [START_REF]FCC rules and regulations[END_REF]. Ultra-Wide-Band Impulse Radio (IR-UWB) is a well-known technique based on the transmission of short duration pulses which demonstrated great potential for high and low data rates in short range wireless communications in the 3.1-10.6GHz bandwidth [START_REF] Vauche | A 100MHz PRF IR-UWB CMOS transceiver with pulse shaping capabilities and peak voltage detector[END_REF]. The IEEE 802.15.4a standard [START_REF]Part 15.4: Low-Rate Wireless Personal Area Networks (LR-WPANs)[END_REF] also allows the use of the sub-GHz UWB bandwidth. In addition of the IR-UWB advantages which have been demonstrated in the 3.1-10.6GHz bandwidth such as low power consumption and high data rate capabilities, the use of the sub-GHz UWB band allows low transmission path loss and potentially better performance regarding the power consumption because active components are working far away from their cutoff frequency even when low cost technologies are used.

Very few works regarding pulse generators have been published in the UWB sub-GHz band. In [START_REF] Stoopman | A Sub-GHz UWB Pulse Generator for Wireless Implantable Medical Devices[END_REF] a relaxation oscillator followed by a power amplifier is used to generate the pulses. The pulse generator is tunable and shows a power consumption of 78pJ/bit. In [START_REF] Liu | An Energy-Efficient Ultra-Wideband transmitter with an FIR Pulse-Shaping Filter[END_REF], a pulse generator, which uses a finite impulse response (FIR) digital filter, demonstrates a very low power consumption of 30pJ/bit but no solution is implemented to compensate Process Voltage and Temperature (PVT) variation. In [START_REF] Joo | A tunable pulse generator for Sub-GHz UWB systems[END_REF] a tunable architecture based on a delay line and an elementary pulse combiner is proposed, but the output magnitude (337mVpp) does not exploit the full range of the allowed peak power limitation of the sub-GHz band.

In this paper, we present a fully digital and tunable pulse generator for the sub-GHz UWB channel of IEEE 802.15.4a. Post-Layout simulations in a 0.13µm CMOS technology show that IEEE 802.15.4a compliant pulses of 900mVpp magnitude with a center frequency of 500MHz and a 650MHz -10dB bandwidth could be generated. The power consumption is 2.7mW at 100MHz that gives an energy consumption of 27pJ by pulse. The power consumption reduces at 192µW for 100 kHz data rate. The simulated center frequency tuning range is 200 MHz allowing PVT compensation. This paper is organized as follows. In section II an analysis of Ultra-Wide Band communications regulations in the UWB sub-GHz band of the 802.15.4a standard gives relationship between data rate and the maximum allowed pulse magnitude. Section III presents the architecture and the design of the fully digital tunable generator in a 0.13µm CMOS technology. Post layout simulation results are given in section IV.

II. ULTRA-WIDE BAND COMMUNICATIONS REGULATIONS

ANALYSIS IN THE UWB SUB-GHZ BAND The IEEE 802.15.4a standard [START_REF]Part 15.4: Low-Rate Wireless Personal Area Networks (LR-WPANs)[END_REF] defines an UWB sub-GHz channel of 499.9MHz -3dB bandwidth centered on 499.9MHz. Its -10dB (resp. -18dB) bandwidth is about 650MHz (resp. 800MHz) as shown in Fig. 1. As shown in [START_REF] Vauche | Fully tunable UWB pulse generator with zero DC power consumption[END_REF], and presented in Fig. 2 in the case of a 5.3 GHz bandwidth UWB pulse, a pulse s(t) could be approximated by a combination of elementary pulses:

1 11 ( ) ( ) Nn n n p np s t A p t      (1)
where pn(t) is the mathematical expression of the elementary pulse which has a magnitude An and a duration n (with τ0 = 0). In [START_REF] Vauche | Fully tunable UWB pulse generator with zero DC power consumption[END_REF], a method to approximate Gaussian-like pulse is presented. This method uses N sinusoidal elementary pulses pn(t) with a constant duration .

The pulse length  in this simplified case is:

0 1 2. f   (2)
where f0 is the central frequency of the channel.

In order to define N, the number of elementary pluses, the following equation of Gaussian pulse is used : 0 ( ) .exp( ).sin(2 . ) 2

s t A t f t m      ( 3
)
where A is the magnitude of the Gaussian pulse and α is related to the -XdB bandwidth as follows: 10 .

2.ln(10 )

XdB X BW              (4)
In (3) m=1 for an odd oscillation number and m=0 otherwise. Because a Gaussian pulse has an infinite duration, for practical design consideration s(t) is bounded in this work to oscillations having a magnitude greater than 10% of the Gaussian pulse magnitude. Using this approximation, ( 3) and ( 4) the minimum number of elementary pulses N defined in (1) is 3 if m=0 and 4 otherwise for the sub-GHz UWB IEEE 802.15.4a channel.

The magnitudes of elementary pulses and their durations are given in Table I for the sub-GHz UWB IEEE 802.15.4a channel for N=3 and N=4. The corresponding time domain waveforms are given in Fig. 3.

Peak power magnitude and average power magnitude of UWB communications are regulated by FCC which sets a peak value of -7.9dBm and an average value of -49.1dBm/MHz for the PSD in the sub-GHz band [START_REF] Bourdel | A 9pJ/ Pulse 1.42Vpp OOK CMOS UWB pulse generator for the 3[END_REF]. While the peak PSD value depends only on the UWB pulse shape and magnitude, the mean PSD value depends also on the data rate. Fig. 4 gives the allowed maximum magnitude as a function of the data rate for the Gaussian pulse defined in (3) to comply with FCC regulation in the sub-GHz UWB band. Fig. 4 shows that the UWB pulse peak to peak magnitude should not exceed 0.9V even at low data rates.

In this work, we choose to design a pulse generator using four elementary pulses. Indeed using four elementary pulses instead of three enables more setting capabilities in the architecture which allow an easier control of the pulse shape in order to be compliant with the IEEE802.15.4a standard. The UWB pulse generator architecture is given in Fig. 5. It is composed of two main functions: an elementary pulse generator and a pulse combiner. The elementary pulse generator provides the four elementary square pulses psqn(t) when it receives a trigger edge. From the outputs of the elementary pulse generator, the pulse combiner gives the four oscillations pn(t)with the right magnitude of the UWB pulse and drives the 50  antenna.

A. Elementary pulses generator

The elementary pulses generator is composed of a voltage control delay line (VCDL) followed by a logic edge combiner. The VCDL is based on tunable delay buffers as shown in Fig. 5. The delay cells are built up of two CMOS inverters loaded by a capacitor C in series with a NMOS as shown in Fig. 5. The widths of elementary pulses could be modified thanks to the variation of the voltage VCTRL applied on the MOS in series with the capacitor. C is sized to achieve a delay of 500ps (C=850fF) when VCTRL is equal to its nominal value 0.7V. The other part of the VCDL is built up with high speed inverters, which are used as buffers for each output. Phantom cells are inserted to keep a constant load between each elementary delay cells.  (ns)

1 1 1 1 n A (N=3) -0.537 1 -0.537 / n A (N=4) 0.260 -1 1 -0.260
The second part of the elementary pulses generator is the logic edge combiner circuit, which combines the five edges given by the VCDL. As shown in Fig. 5, the negative pulses psq1(t) and psq3(t) are generated thanks to a NXOR-function. The edges OUT1 and OUT2 form psq1(t) while psq3(t) is generated by the combination of the edges OUT3 and OUT4. The positive pulses psq2(t) and psq4 (t) are generated by a XOR function. The edges OUT2 and OUT3 will form psq2(t) while psq4 (t) is generated by combining the edges OUT4, OUT5. Each logic gate is followed by a buffer to reduce the rise time.

B. Pulses combination circuit

In order to generate a Gaussian-like UWB pulse, elementary pulses should have a sinusoidal shape. Thus the square pulses psqn(t) must be transformed into sinusoidal ones. To achieve this transformation, the drive capacities of the pulses combination circuit inputs buffers have been reduced by choosing a W/L ratio smaller than the one of the output buffers of the logic edge combiner (W1/L). Finally each elementary pulse is injected to a common source transistor in order to set its magnitude as shown in Fig. 5.

Transistors sizes ratios are chosen in a way to achieve the magnitude to comply with the FCC IEEE mask and to be able to drive the 50Ω antenna. A capacitor of 5pF is added at the end of the pulse combination circuit for the DC isolation of the antenna.

IV. POST-LAYOUT SIMULATION AND COMPARISON

The UWB pulse generator has been implemented in a 0.13µm STMicroelectronics technology. The layout, given in Fig. 6, shows a core size of 390µm by 370µm. Post-layout simulations have been made using the software Cadence for a supply voltage of 1.2V. V. CONCLUSION

In this paper, a fully tunable UWB pulse generator has been designed and implemented in a 0.13µm CMOS technology. Post-Layout simulation results show its ability to generate sub-GHz UWB pulses with a peak to peak magnitude voltage of 0.9V. The pulse generator consumes only 27pJ/pulse and allows a central frequency tuning range from 400MHz to 610MHz.

Fig. 1 .

 1 Fig. 1. IEEE 802.15.4a standard UWB sub-GHz channel.

Fig. 2 .

 2 Fig. 2. Example of Gaussian 5.3 GHz UWB pulse formed by N=6 elementary pulses[7].

Fig. 3 .Fig. 4 .

 34 Fig. 3. Time domain waveform for a 3 and 4 elementary pulses Gaussian approximation

Fig. 5 .

 5 Fig. 5. Pulse generator architecture

Fig. 7 .

 7 Fig. 7.a represents the time domain UWB pulse generated using nominal settings (VCTRL=700mV). The peak to peak voltage magnitude which is about 900mV reach the maximum allowed peak value as indicated in Fig 4.Fig. 7.b shows the spectrum of the UWB pulse generated. The central frequency is 496MHz and the -10dB and -18dB bandwidths are 629.76MHz and 792.5MHz respectively. The average current consumption of the generator, simulated for a PRF of 100MHz, is 2.24mA with a supply voltage of 1.2V. That gives a 2.7mW average power consumption and a 27pJ energy

Fig. 7 .

 7 Fig. 7.a represents the time domain UWB pulse generated using nominal settings (VCTRL=700mV). The peak to peak voltage magnitude which is about 900mV reach the maximum allowed peak value as indicated in Fig 4.Fig. 7.b shows the spectrum of the UWB pulse generated. The central frequency is 496MHz and the -10dB and -18dB bandwidths are 629.76MHz and 792.5MHz respectively. The average current consumption of the generator, simulated for a PRF of 100MHz, is 2.24mA with a supply voltage of 1.2V. That gives a 2.7mW average power consumption and a 27pJ energy consumption by pulse for this data rate. A figure of merit  is introduced to achieve a comparison of the energy consumed by pulse regarding the pulse magnitude:

Fig. 6 .Fig. 7 .

 67 Fig. 6. Pulse generator layout

TABLE I .

 I ELEMENTARY PULSES PARAMETERS FOR A GAUSSIAN PULSE APPROXIMATION IN THE SUB-GHZ IEEE 802.14.4A BAND.

TABLE II .

 II SUMMARY AND COMPARAISON WITH STATE OF ART

	Reference	[4]	[5]	[6]	This work
	Technology (µm)	0.13	0.18	0.5	0.13
	VDD (V)	2.5	1	3	1.2
	Vpp (mV)	120	-	372	900
	E (pJ/pulse)	78	30	130	27
	BW-10dB (MHz)	550	150	500	629.76
	Consumption (mW)	0.078	1.5	1.3	2.7
	@	@	@	@	@
	PRF(MHz)	1	50	10	100
	 (pJ/pulse/V)	650	-	349	30


Y Gan

R Vauche

J-F Pons

W Rahajandraibe

Dry Electrode Materials for Electrocardiographic Monitoring

Keywords: ECG, Dry electrode materials, Electrical Double Layer Impedance I

In this paper, a preliminary study about the most suitable dry electrode materials is done in the context of electrocardiogram (ECG) measurement systems used at home. Therefore, the electrical equivalent circuit between two electrodes through the body is developed as well as the ECG quality evaluation factors. Finally, an initial series of impedance and ECG quality measurements is presented which led to the conclusion that dry stainless steel electrodes seem to provide the best ECG signal quality.

INTRODUCTION

To diagnose heart disease which remains the most common death cause in the world with 17.3 million deaths each year [START_REF] Benjamin | Heart Disease and Stroke Statistics-2017 Update[END_REF], the 12-lead electrocardiogram (ECG) monitoring (which requires 12 wet Ag/AgCl electrodes) is the most common used ECG acquisition method in a hospital environment. However, with this technique, the patient must stay still and electrodes must be placed by medical staff, which is hardly possible at home. Thereby, to diagnose heart disease at home, simple ECG monitoring systems must be developed. The one developed by WitMonki is a credit card format device with two electrodes where left and right thumbs have to be placed. To make the equipment widely accepted, dry electrodes without electrolytic gel have been adopted. Nevertheless, ECG signal acquisition is more sensitive to motion when dry electrodes are used. For this reason, this paper presents the on-going research works concerning the materials of the electrodes. The main objectives are 1) selecting the material which allows ECG to be as few motion sensitive as possible while keeping its dynamic high, 2) characterizing its electrical parameters, and 3) relating ECG quality and electrical characteristics as future work.

To do so, an electrical model of the impedance between two electrodes through the body is presented in Section II. Next, Section III develops the model fitting procedure and measurement results for several materials. Finally, ECG quality evaluation factors are given in Section IV, and a conclusion presents the direction for the future researches.

II. ELECTRICAL MODELING BETWEEN ELECTRODES

When a finger touches an electrode, a spatial region of finite thickness, usually on the order of angstroms, is formed at the surface between the material and the body fluid. It can be seen as a parallel arrangement of opposite charges called Electrical Double Layer (EDL) [START_REF] Wang | Accurate simulations of electric double layer capacitance of ultramicroelectrodes[END_REF]. To model the electrical behavior of the EDL, the Constant Phase Element (CPE) [START_REF] Jorcin | CPE analysis by local electrochemical impedance spectroscopy[END_REF] has been introduced and its admittance has been defined as follows:

where n is a coefficient between 0 and 1 and Q is a constant. Thus, when n is equal to 0, the CPE is equivalent to a resistor R (R=1/Q) whereas when n is equal to 1, the CPE is equivalent to a capacitor C (C=Q). Using the CPE, the impedance Z EE seen between both electrodes can be modeled as shown in Fig. 1 where R P /2 in parallel to the CPE models the thumb-electrode interface and R S the human body resistance. Hence, the impedance Z EE is defined as follows:

The real part and the imaginary part of Z EE can also be formally written as: which allows Z EE to be represented in the Nyquist plot as shown in Fig. 2.

III. FITTING PROCEDURE AND MEASUREMENT RESULTS

To extract the values of R P , R S , Q, and n for different materials, a first set of complex impedance measurements was obtained with the impedance tester from Metrohm France SAS between 0.1 Hz and 1 kHz for several brass electrodes with a deposited layer of stainless steel (SST), platinum (Pt), chromium (Cr), palladium (Pd), and gold (Au) respectively or

( ) ( ) n CPE j Q Y ω ω ⋅ = , (1)
( ) ( ) n P P S EE j Q R R R Z ω ω + + = 1 . ( 2
) ( ) [ ] ( ) (
)

n P n P n P P S EE Q R n Q R n Q R R R Z 2 2 2 2 2 cos 2 1 2 cos Re ω π ω π ω ω + + + + = , ( 3
) ( ) [ ] ( ) (
)

n P n P n P EE Q R n Q R n Q R Z 2 2 2 2 2 cos 2 1 2 sin Im ω π ω π ω ω + + - = , (4)
without layer (BS). The modulus of the measured impedance Z EE is shown in Fig. 3 for each material and it appears that the higher the overall material impedance is, the noisier the measured impedances between 1Hz and 100Hz are. The extracted values of each component of the electrical equivalent circuit seen between both electrodes are shown in Table I for each material. To obtain them, the Scilab "datafit" function has been used on the imaginary part of Z EE using the quasi-Newton algorithm for several initial values of n (from 0 to 1 with a step of 0. The solution which gives the lowest error has been chosen. Finally, to validate the accuracy of the model, the coefficient of determination R 2 computed from the measured and the modeled Z EE (ω) is also given in Table I. The model can be accepted since the values of R 2 were all close to 1.

IV. ECG QUALITY EVALUATION FACTORS

An ECG signal obtained through clinical wet electrodes with electrolytic gel was used as a reference to compare the ECG signal quality from the dry electrodes. These two signals were recorded simultaneously as shown in Fig. 4. A radar chart was then produced by gathering several performance metrics for each material as seen in Fig. 5. The metrics were normalized to guarantee that their values lay within the same range. In this chart, the smaller a polygon area is, the higher the ECG signal quality is. The metrics used can be divided into two groups; one considering the comparison between the ECG signal obtained with experimental electrodes (denoted x), and the reference one (denoted y), another only considering the experimental signal itself. In the first group, the reciprocal of the cross-correlation coefficient and the Root-Mean-Square-Error (RMSE) were defined as follows:

The RMSE was calculated for two frequency bands (5Hz-25Hz, and 25Hz-40Hz). The second group consists of the standard deviation in low-frequency range (less than 2Hz) and the Power Spectrum Density (PSD) for three frequency bands (45Hz-55Hz, 40-150Hz, and >150Hz). Standard deviation, RMSE and cross-correlation mainly represent the motion sensitivity of the electrode, while PSD quantify noise. Finally, the radar chart indicated that the stainless steel is the bestsuited material for ECG signal acquisition. In this paper, the on-going research works related to the selection of the most suitable dry electrodes material for ECG monitoring systems used at home have been presented. In addition, to identify the materials and the associated electrical characteristics which allow high-quality ECGs to be obtained, an equivalent electrical circuit of the impedance between two electrodes through the human body using the CPE has been presented. Next, the series of impedance measurements allowed the parameters to be computed for each material and the accuracy of the model to be evaluated. Finally, several performance metrics for the chosen materials were defined and plotted in a radar chart to visualize the ECG signal quality, from which the stainless steel has been designated as the bestsuited electrode for ECG signal acquisition, unlike the gold. Finally, complementary study showed that the ECG signal quality could be related to the Q value of the CPE. Therefore, an in-deep measurement campaign must be done in the future to confirm these observations.

Fig. 1 .

 1 Fig. 1. The equivalent electrical circuit between the left and right electrodes.

Fig. 2 .

 2 Fig. 2. Nyquist diagram of the impedance between both electrodes.

  01). The other initial values were R S = Re[Z EE (2π0.1)], R P = Re[Z EE (2π1000)] -Re[Z EE (2π0.1)], and Q = 0.05 µΩ -1 .s n .

Fig. 3 .

 3 Fig. 3. Measured modulus values of ZEE.

Fig. 4 .

 4 Fig. 4. Reference (black line) and experimental (gray line) ECG signals obtained with Au electrodes (a) and SST electrodes (b).

Fig. 5 .

 5 Fig. 5. Radar chart of ECG quality indicators V. CONCLUSION

TABLE I .

 I SUMMARY OF THE FITTED PARAMETER RESULTS

		BS	SST	Pt	Cr	Pd	Au
	Rs (Ω)	5542	4325	6450	10300	1000	33900
	Rp (MΩ)	1.00	1.16	1.12	1.35	1.18	1.44
	n	0.62	0.58	0.66	0.55	0.68	0.62
	Q (µΩ -1 .s n )	0.20	0.25	0.16	0.24	0.12	0.09
		0.997	0.994	0.998	0.999	0.987	0.970


R Vauche

H Bounaceur

S Zerenini

N Dehaese

J Gaubert

H Barthelemy

V Gies

Implementation of the Standardized Human Body Communications -A Feasibility Study

Keywords: Human Body Communications, Capacitive Coupling, Human Body Path Loss I

Due to the disparity of the measured human body path losses published in the literature, it is proposed here to make a feasibility study about the implementation of the Human Body Communications (HBC) defined in the IEEE 802.15.6 standard. In this way, the path loss required by the standard is firstly computed. Next, a measurement setup of the human body path loss is presented. Finally, measurement results are compared to the required one and pave the way to studies closer to real use cases.

INTRODUCTION

In 2012, the IEEE 802.15.6 standard [1] dedicated to Wireless Body Area Networks (WBAN) has introduced the specifications of 3 physical layers (PHY): (i) a NarrowBand (NB) PHY specification using the ISM frequency bands, (ii) an Ultra-WideBand (UWB) PHY specification using the 3.1GHz-10GHz frequency band, and (iii) a Human Body Communications (HBC) PHY specification using a channel around 21MHz.

On the contrary of standard wireless communications which use for example the air or the vacuum as propagation medium [START_REF] Shannon | A Mathematical Theory of Communication[END_REF], the HBC use the human body as propagation medium and antennas are replaced by electrodes. In this way, HBC can be associated to a particular type of Body Coupled Communications (BCC) which uses the capacitive coupling [START_REF] Callejón | Distributed Circuit Modeling of Galvanic and Capacitive Coupling for Intrabody Communication[END_REF] where only one signal electrode is used on the emitter side and on the receiver side as shown in Fig. 1.

Since HBC have been standardized, lots of simulated or measured values of the human body path loss have been published as shown in Fig. 2 [START_REF] Maity | Characterization of Human Body Forward Path Loss and Variability Effects in Voltage-Mode HBC[END_REF]. However, the human body path loss around 21MHz is at best between 20dB and 60dB which leads to a high disparity even if a given configuration (coupling type, electrodes position, electrodes type, etc.) is considered. Moreover, it appears that according to the effective value of the human body path loss, the implementation of standardized HBC could be impossible with the specifications (transmitted power, receiver sensitivity, bit rate) defined in the IEEE 802.15.6 standard.

From this analysis, it is proposed in this paper to review the situation by computing in section II the human body path loss required by the HBC PHY specification defined in the IEEE 802.15.6 standard. Next, a setup for human body path loss measurement is proposed in section III. Finally, the measured values of human body path loss are compared to the published ones in section IV and a conclusion on the feasibility of the HBC implementation is given.

II. COMPUTATION OF THE HUMAN BODY PATH LOSS

According to the IEEE 802.15.6 standard, the electric field produced by a HBC device shall never exceed 30µV/m when it is measured at a distance of 30m in free space, in addition to be compliant with local regulations. This electric field strength E can be converted in a surface power density SPD with the help of Poynting theorem as follows:

η 2 E H E SPD = × = ( 1
)
where H is the magnetic field strength and η is the wave impedance of a plane wave in free space which is equal to the square root of the free space permeability-permittivity ratio (η can be approximated by 120π for air or vacuum). Next, with the help of the Friis formula, it is possible to write the surface power density as a function of the power emitted by the transmitter electrode P TX as follows:

( )

2 4 d P SPD TX × × = π (2)
where d is the distance between the transmitter electrode and the electric field probe. From ( 1) and (2), it is now possible to directly compute the power emitted by the transmitter electrode from the electric field strength as follows:

η π 2 2 4 E d P TX × × × = (3)
which gives a maximum emitted power of 27nW (-45.7dBm). Finally, it is possible to compute the maximum human body path loss (dB) permitted for HBC by subtracting the maximum authorized value of P TX (dBm) by the minimum receiver sensitivity (dBm) specified by the standard. The minimum receiver sensitivity, the maximum emitted power, and the maximum human body path loss permitted by HBC are summarized in Table I for the different standardized bit rates.

III. MEASUREMENT OF THE HUMAN BODY PATH LOSS

To measure the human body path loss, two measurement methods have been in used in the literature. The first one uses a radiofrequency Signal Generator (SG) and a Power Spectrum Analyzer (PSA) [START_REF] Bae | The Signal Transmission Mechanism on the Surface of Human Body for Body Channel Communication[END_REF] whereas the second one uses a Vector Network Analyzer (VNA) [START_REF] Sayedi | A novel intrabody communication transceiver for biomedical applications[END_REF]. In this paper, the first method using a SG SMBV100A and a PSA FSH4 from Rhode & Schwarz is considered as shown in Fig. 3. To properly connect the TX & RX electrodes, SMA cables have been used. Finally, the used test electrodes are dry electrodes for electrical muscle stimulation from Comepa which consist of a 14mm-diameter disc of silver alloy where the protection sheet has not been removed to avoid oxidization. The values of the measured human body path loss have been summarized in table II for various electrodes position (averaged on three persons). They have been obtained using a 21MHz continuous wave with a level of -46dBm in order to be compliant with the maximum allowed emitted power. Finally, the PSA has been configured with a span of 10kHz around 21MHz, a Resolution BandWidth (RBW) of 300Hz, and an attenuation of 0dB to reduce its noise floor to -100dBm. IV. CONCLUSION

The measured human body path loss obtained with the presented setup is between 32dB and 38dB according to the electrodes position, which is between the minimum (20dB) and the maximum (60dB) published in the literature review [START_REF] Maity | Characterization of Human Body Forward Path Loss and Variability Effects in Voltage-Mode HBC[END_REF]. It also appears that in these conditions the human body path losses are always lower than the maximum one permitted by the standard which is between 42.62dB and 51.65dB according to the targeted bit rate. Thus, these measured values of the human body path loss show that standardized HBC can be implemented. However, to confirm this with a configuration closer to a real use case, a basic TX patch, which will generate a 21MHz continuous wave with a level of -46dBm, will be designed in addition to a RX patch which will be able to estimate the power received around 21MHz. These patches, which include each one an electrode, will be placed on the skin for different configurations and will make possible the measurement of the human body path loss during motions in order to verify its compatibility with standardized HBC.

Fig. 1 .

 1 Fig. 1. The two types of body-coupled communications: a) the galvanic coupling, b) the capacitive coupling [3].

Fig. 2 .

 2 Fig. 2. Attenuation of the human body previously published [4].

Fig. 3 .

 3 Fig. 3. Measurment setup of the human body path loss (a), electrodes localizations (b).

TABLE I .

 I MAXIMUM HUMAN BODY PATH LOSS ALLOWED BY HBC.

	Bit rate	Minimum Receiver	Maximum Emitted	Maximum Path
	(kbps)	Sensitivity (dBm)	Power (dBm)	Loss (dB)
	164.1	-97.35	-45.7	51.65
	328.1	-94.34	-45.7	48.64
	656.3	-91.33	-45.7	45.63
	1312.5	-88.32	-45.7	42.62

TABLE II .

 II MEASURED HUMAN BODY PATH LOSS.

	Electrodes position	Range (cm)	Measured Path Loss (dB)
	A-B	20	32.8
	A-C	40	35.8
	A-D	60	36.4
	A-E	120	37.2


M Egels

R Vauche

E Bergeret

Antennas Design for the Characterization of an UHF Remotely Powered UWB Pulse Generator

In the context of an UHF remotely powered UWB tag, this paper presents the design of the UHF antenna used to harvest energy on the tag side, and the design of the antenna used to receive emitted UWB pulses on the reader side. They have been used to characterize an UHF remotely powered UWB pulse generator whose the measurement results conclude this article.

INTRODUCTION

With the grow of the internet of things and its billions of smart devices, battery less radio links become an attractive concept. In this context, the combination of the Ultra-WideBand (UWB) impulse radio and the UHF Radio Frequency IDentification (RFID) can become a major competitor for short range communications. Indeed, it has been demonstrated with help of simulations that the power consumption of an UWB pulse generator can be low enough to be remotely powered, and thus advantages of the both wireless technologies to be associated [1]. However, due to the high power levels used in UHF-RFID, it is difficult for a remote reader to efficiently detect low energy UWB pulses without a specific UWB antenna which filters enough UHF frequencies. Moreover, cohabitation of an UWB antenna and an UHF antenna on a PCB small enough to be used as a tag is a challenge.

Based on the UWB pulse generator presented in [START_REF] Vauche | High efficiency uwb pulse generator for ultra-low-power applications[END_REF] and the UWB antenna presented in [START_REF] Chami | Miniaturisation et intégration d'antennes imprimées pour systèmes communicants ULB pulsés[END_REF], the system presented in [1] has been implemented, and measured in this paper. It focuses on the design of the UHF antenna used to harvest energy, and the design of the antenna used to receive emitted UWB pulses. This paper is organized as follows. In section 2, the system and its PCB is presented. Next, the design of the UHF antenna used to harvest energy required by the pulse generator is presented in section 3 and the antenna used on the reader side to receive emitted UWB pulses while filtering UHF frequencies is presented in section 4. Finally, section 5 presents the measurement results of the UHF remotely powered UWB pulse generator obtained with the designed antennas.

SYSTEME ARCHITECTURE AND PCB DESIGN

Systeme architecture

As shown in Fig. 1 (a), the system is an UHF-UWB transmitter tag which consists in an UHF harvesting unit and an UWB transmitter. Both have been implemented in the same chip using a 130 nm CMOS technology from STMicroelectronics. A top view of the designed UHF-UWB transmitter tag prototype can be seen in Fig. 1 (b).

The UWB transmitter has been designed to generate 3.1-10.6 GHz pulses and implements the filter excitation technique which allows output dynamic voltage to be higher than the voltage supply while keeping low the power consumption [START_REF] Bourdel | A 9-pj/pulse 1.42-vpp ook cmos uwb pulse generator for the 3.1 2013;10.6-ghz fcc band[END_REF]. Its power consumption P U W B can be written as follows:

P U W B = P DC + E AC • P RF mean (1)
where P DC is the power consumed when no pulse is emitted, E AC is the additional energy consumed when a pulse is emitted, and P RF mean is the mean Pulse Repetition Frequency (PRF) of the emitted pulses. To supply the UWB transmitter with an UHF rectifier, the main issue comes from P DC which has to be reduced as far as possible to optimize the efficiency of the system. Thanks to the filter generation technique whose the used architecture is described on Fig. 2 (a), P DC is here only due to leakage currents since the baseband pulse generator uses only CMOS logic gates. Moreover, to reduce leakage currents, the UWB transmitter takes advantage of the used 130 nm technology by using a multi-Vt architecture [START_REF] Mutoh | 1-v power supply high-speed digital circuit technology with multithreshold-voltage cmos[END_REF][6] based on the High Speed (HS) and Low Leakage (LL) MOS transistors provided by the technology. For a supply voltage of 1.2 V and a load impedance of 50 Ω, this leads to a measured P DC of 3.91 µW, a measured E AC of 68 pJ per pulse (which corresponds to 34 pJ per bit for an OOK modulation), a measured peak to peak output voltage of 2 V, and an emitted energy of 2.34 pJ per pulse [START_REF] Vauche | High efficiency uwb pulse generator for ultra-low-power applications[END_REF]. As shown in Fig. 2 (b), the UHF harvesting unit is built around a rectifier which is based on a three stages multiplier. It has been interconnected with the UWB pulse transmitter to work as indicated in Fig. 3. Firstly, V DD increases when an UHF carrier is sensed by the rectifier. Next, a rising edge occurs on CLK when V DD is higher than a threshold voltage fixed by the threshold detector circuit. This CLK rising edge triggers the UWB transmitter wake-up and the generation of one UWB pulse. One nanosecond later, the UWB pulse generator goes into a deep sleep mode and a rising edge occurs on RESET V DD which turns V DD to 0 V. Finally, a falling edge occurs on CLK followed by another one on RESET V DD which allows V DD to increase again. Thus, this UHF-UWB tag continuously emits UWB pulses with a PRF which is directly proportional to the power available on the UHF antenna output.

PCB design

The chip described previously is connected with the help of bondwires to the two required antennas. The first one is used to emit UWB pulses through the air and the second one is used to receive power from an UHF carrier for supplying the tag. These two antennas are placed on the same PCB. In order to limit losses in the UWB frequency band, a low loss Roger 4003 substrate has been chosen ( r = 3.3 and tan δ = 2 , 10 -3 ). As the UWB antenna operates in the 3.1-10.6 GHz frequency band, the inlayer has been chosen in order to have mono modal Grounded Coplanar Wave Guide (G-CPW) and rigid board (see Tab. 1). The G-CPW is designed using the top first gold layers (layer 1 and 3). Layer 3 is also dedicated to the G-CPW ground and the layer 1 to the signal and lateral grounds of the G-CPW. Widths and gaps of signal lines are set to have a 50 Ω characteristic impedance and vias are added to connect lateral grounds to the internal ground in GHz UWB antenna are not described in this paper but are fully presented in [START_REF] Chami | Miniaturisation et intégration d'antennes imprimées pour systèmes communicants ULB pulsés[END_REF] and [START_REF] Chami | Design and integration of uwb antennas for high data rate miniature impulse radio transmitter[END_REF] and the next section deals with the design of the UHF antenna. The UHF tag antenna has been designed to supply the tag from an UHF carrier. It has also been directly matched to the input UHF rectifier in order to cancel the losses added by a matching circuit [START_REF] Marrocco | The art of uhf rfid antenna design: impedance matching and size-reduction techniques[END_REF]. To achieve it in a limited area (i.e. a credit card 87 × 50 mm 2 ), a slot folded dipole [START_REF] Marrocco | The art of uhf rfid antenna design: impedance matching and size-reduction techniques[END_REF] feeded by a buried micro strip line is chosen. This topology provides a maximum peak gain in the zero azimuth direction and a quasi isotropic gain in the zero azimuth plan (see Fig. 4).

The length of the antenna is set to have a maximum radiated gain to maximize the wireless power transfer range and an inductive input impedance to match with the rectifier IC input. The final match is achieved with the optimization of the length and the width of the open ended buried microstrip line. Figures 4 (a can not be directly measured for a non 50 Ω antenna). Finally, figure 4 (c) shows the simulated reflection coefficient between the antenna and the input UHF rectifier computed as follows :

S 11 = Z ant -Z * ship Z ant + Z ship . ( 2
)

UHF rectifier characterization

A manufactured prototype has been fully measured in the anechoic chamber of the lab. The measurements have been performed with a vector signal generator connected to a 40 dB power amplifier, itself connected to a 7 dBi realized gain antenna through a 10 m long low loss cable. Finally, the rectifier has been placed at 2.5 m of the phase center of the test antenna. With this setup, the incident power on the rectifier antenna can be computed from the Friss formula as usually. However, only the output rectifier voltage can be measured here to characterize the performances of the UHF rectifier and the UHF antenna. Indeed, the efficiency and the output power of the rectifier cannot be computed since the current consummed by the UWB pulse generator cannot be measured. The output voltage has also been measured for several incident power (P in ) in the 780-880 MHz frequency band.

As shown in Fig. 5 (a), the maximum output rectifier voltage is obtained at 815 MHz which represents a shift down of 53 MHz compared to simulations (i.e. 6%). This can be explained by the deviation of the input transistor behavior from the large signal model used to design the rectifier, in addition to the length of the bondwires used to connect the IC to the antenna. An other reason can be the frequency shift between the maximum efficiency of the UHF rectifier and the minimum matching of the antenna. In addition, a voltage step can be observed in the output voltage between 0.15 and 0.6 V. It is due to the leakage current of the pulse generator which has a peak value for this voltage range since leakage current versus supply voltage is not a monotonic function. This effect can be seen in Fig. 5 (b) too since it shows the output voltage of the rectifier as a function of the incident power at 815MHz. Finally, the last performed measurement is the output voltage for several orientations of the UHF antenna. They have been performed in the E and V polarization planes. The results plotted on Fig 5 (c) confirm the simulated shape of the radiation pattern.

READER ANTENNA

As the maximum gain direction is the same for the UWB antenna and the UHF antenna which have been implemented on the PCB tag, a specific UWB antenna which rejects the UHF frequency band and operates in 3.1-10.6GHz frequency band has been designed.

Design

The main goal of this antenna is also to receive the 3.1-10.6 GHz UWB pulses emited by the pulse transmitter and to reject the UHF frequencies. To achieve it, an Archimedes spiral complimentary antenna is chosen. This antenna pattern has tree major advantages that can be exploited in the particular context of the targeted application. Firstly, it has a frequency independent behavior for the frequency band where its greater diameter is greater than λ/π, which means that its gain and its impedance are constant for these frequencies. Next, the radiated electromagnetic field is circular for this frequency band and linear for lower frequencies. Finally, the antenna is totally mismatched for lower frequencies.

By taking into account these considerations, a spiral antenna with a flat gain and an axial ratio lower than 3dB in the 3.1-10.6GHz frequency band can be obtained using the following steps:

1. Its diameter is set to satisfy the condition D max > λ 3.1 GHz /π.

2.

As the spiral must be truncated in its center for manufacturing reasons, its minimal diameter is set to satisfy D min < λ 11 GHz /π.

3. Width and gap between spiral harms are set to obtain complementary spiral antennas. This provides a constant gain and a constant input impedance of 120 Ω in the 3.1-10.6 GHz frequency band.

4. A ring is added and connected to the antenna in order to miniaturize and make constant the axial ratio and the radiated gain for the lower frequencies [START_REF] Ripoche | Spiral antenna miniaturization in very high frequency band[END_REF].

5.

A balun is added in order to match the antenna to 50 Ω in the UWB frequency band [10][11].

6. As a final step, its length is set to maximize the mismatch antenna in the UHF frequency band.

A prototype of this antenna has been fabricated on a low loss substrate (Taconic TLY with r = 2.2 and tan δ = 1.9 10 -4 ) with a laser engraving process. The Fig. 6 shows the top view of the antenna prototype and the values of the geometric parameters obtained with the design method previously described.

Measurements

The designed spiral antenna has been measured in the anechoic chamber of the lab. All measurements have been performed in the 0.5-12 GHz frequency band using a 24GHz Rhode & Schwarz vector network analyzer and a 18 GHz wideband test antenna. The matching is plotted on Fig. 7 (a). The comparison between measured and simulated results shows a good accuracy up to 8 GHz. However, a mismatch at frequencies higher than 10 GHz is observed but does not affect the antenna performances. As expected, the antenna is fully mismatched for the sub-GHz frequency band.

The circular realized gain and Axial Ratio (AR) have been computed by measuring the linear realized gain in several polarization planes. Results plotted on Fig. 7 (b) and (c) show that the AR is increased up to 3 dB for frequencies higher than 8 GHz. This causes the lower circular realized gain in the upper part of the UWB frequency band. It can be due to the use of a SMA connector to connect the antenna to the VNA but does not affect the antenna performances in the UWB frequency band.

UHF REMOTELY POWERED UWB PULSE GENERATOR MEASUREMENT

In this section, the characterization of the UWB pulses emited by the UHF remotely powered UWB pulse generator is done. As a reminder, it uses as UWB antenna the one presented in [START_REF] Chami | Miniaturisation et intégration d'antennes imprimées pour systèmes communicants ULB pulsés[END_REF], and as UHF antenna the one presented in section 3 of this article. In addition, it emits UWB pulses with a PRF directly proportional to the power available on the UHF antenna output since a pulse is generated when the rectifier ouput voltage is higher than a fixed threshold as shown in Fig. 3. Thus, the potential bit rate available for the UWB radio link can be measured in this configuration by the mesurement of the pulse repetition frequency.

To measure the continuously emitted pulses, an UWB receiver is emulated in the anechoic chamber by a real time 12 GHz oscilloscope, the UWB reader antenna presented in section 4, and a 26 GHz LNA which allows UWB pulses to be highlighted from oscilloscope noise floor. The LNA gain which has been measured with the cables used to capture the UWB pulses is equal to 20 dB and the UWB (resp. UHF) reader antenna has been placed at 0.2m (resp. 2.5m) of the UHF-UWB tag. As at 6.5GHz the UWB tag (resp. reader) antenna has a gain of 0 dBi (resp. 4dBi), and the waves attenuation is equal to 43dB for a range of 0.2m, the power attenuation of UWB pulses is equal to 19dB. This is confirmed by the UWB pulse shown in Fig. 8 (a) which has been measured on the oscilloscope since its energy can be estimated to 0.2pJ [START_REF] Vauche | High efficiency uwb pulse generator for ultra-low-power applications[END_REF] whereas the energy of the emitted UWB pulse can be estimated to 2.37pJ [START_REF] Vauche | High efficiency uwb pulse generator for ultra-low-power applications[END_REF]. Finally, as shown in Fig. 8 (b), UWB pulses are continuously emitted up to a PRF of 165 kpulse/s for an incident power between -10dBm to 5 dBm. By considering an OOK modulation, the tag can also potentially send data to the reader with a bit rate up to 330 kbps if an incident power of 5 dBm is received from an UHF carrier. energy from an UHF carrier and the UWB reader antenna used to detect UWB pulses emitted by the tag. The main constraints were for the first one the cohabitation with the UWB tag antenna and the limited available area, and for the second one a sufficient rejection of UHF frequencies which can make difficult the detection of the emitted pulses. Finally, presented UHF-UWB tag has been characterized with the help of the both designed antennas. Measurement show that it a minimum incident power -10dBm and can communicate with a reader at a bit rate of 330kbps for an incident power of 5dBm.

Figure 1 :

 1 Figure 1: Bloc diagram (a) and top view photography (b) of the UHF-UWB tag.

Figure 2 :

 2 Figure 2: Bloc diagram of the UWB pulse generator (a) and bloc diagram of the UHF harvesting unit (b).

Figure 3 :

 3 Figure 3: Main signals of the UHF-UWB transmitter tag

Figure 4 :

 4 Figure 4: Top view photography (a), simulated radiation patern at 868 MHz (b), and matching (c) of the UHF harvesting antenna.

Figure 5 :

 5 Figure 5: Output rectifier voltage as a function of the frequency for varius incident power (a), output rectifier voltage at 815 MHz as a function of the power at 815 MHz (b), output rectifier voltage in azimutal and elevation planes at 815 MHz and 0 dBm incident power (c).

Figure 6 :

 6 Figure 6: Top view photography and geometrical parameters description.

Figure 7 :

 7 Figure 7: Simulated and measured matching of the UWB reader antenna for the 0.5-12 GHz frequency band.

Figure 8 :

 8 Figure 8: Example of a received pulse (a) and pulse repetition frequency of the system (b) for a distance tag-reader of 0.2 m.

Table 1 :

 1 Inlay of the UHF-UWB tag

	Layer n	material	use	thickness
	1	copper	UWB + RFID antennas	30 µm
	2	Roger 4003	dielectric	200 µm
	3	copper	UWB ground and RFID antenna feed	30 µm
	4	Roger 4003	dielectric	500 µm
	5	copper	UWB ground	30 µm
	6	Roger 4003	dielectric	200 µm
	7	copper	UWB ground	30 µm
	3. UHF ANTENNA DESIGN AND UHF RECTIFIER CHARACTERIZATION
	3.1. UHF antenna design


R Assila

Belhadj Mefteh

R Vauche

N Dehaese

J Gaubert

H Barthelemy

Design of a Battery-powered Emitter dedicated to the Characterization of Human Body Communications

Keywords: Goertzel algorithm, HBC, under-sampling processing, microcontroller-based emitter, power estimation

Human Body Communications (HBC), which has been standardized around 21MHz, potentially offers more favourable propagation conditions than usual wireless technologies such as ZigBee, or Bluetooth. However, due their sensitivity to the close environment, the signal attenuation cannot be accurately determined using lab equipment. For this reason, a battery-powered emitter with emitted power estimation capability is fully described in this paper. The prototype implemented in a 5x5 cm PCB is supplied with a 3.3V battery and has been built around a microcontroller (MCU). The MCU has been programmed to generate a 21MHz square carrier and to under-sample it at 480ksps using the MCU embedded Analog-to-Digital Converter (ADC). To enable the carrier power estimation with this constraint, the Goertzel light discrete Fourier transform algorithm has been implemented using particular parameters values. Finally, to overcome the bandwidth limitations of the ADC MCU pin, an external sample & hold circuit driven by the MCU have been added and optimized.

I. INTRODUCTION

Human Body Communications (HBC) are the third physical (PHY) layer defined by the IEEE 802.15.6 standard dedicated to Wireless Body Area Networks (WBAN) which operate close to human body. This PHY layer is primarily responsible for establishing a reliable and physical link to transmit binary data between receiver and transmitter or other equipments with a speed going up to 1.3125Mbits/s and uses a low carrier frequency centered on 21MHz and a large 3 dB bandwidth of 5.25MHz [START_REF]IEEE standard for Local and metropolitan area network -Part 15.6: Wirless Body Area Networks[END_REF].

On the contrary of usual standardized wireless communications such as ZigBee or Bluetooth which use air as a propagation channel [START_REF] Shannon | A Mathematical Theory of Communication[END_REF], the HBC mark a real technological breakthrough since they propose to use the human body as a propagation medium, which potentially offers more favorable propagation conditions, better data protection and more integrated transmitters.

HBC can be associated with two types of Body Coupled Communications (BCC) where antennas are replaced by electrodes as shown in Fig. 1. The first is the galvanic coupling [3][4] where the four required electrodes are signal electrodes fixed on the human body, two on the emitter side and two on the receiver side. The signal is also differentially applied between the two emitter electrodes. The second is the capacitive coupling [START_REF] Callejón | Distributed Circuit Modeling of Galvanic and Capacitive Coupling for Intrabody Communication[END_REF] [START_REF] Seyedi | An empirical comparison of limb joint effects on capacitive and galvanic coupled intra-body communications[END_REF] which equally implies the use of four electrodes. However, two of the four required electrodes are signal electrodes attached to the human body, while the two others are ground electrodes detached from the body. Thus, a current loop, passing through the human body and the close environment such as the ground, is created and allows the signal to go from the emitter to the receiver. In this context, the propagation medium can be considered as a transmission line where the human body is the forward path and the close environment is the return path.

Since HBC were standardized, several studies [START_REF] Wegmueller | Signal Transmission by Galvanic Coupling Through the Human Body[END_REF]- [START_REF] Cai | Impulse radio intrabody communication system for wireless body area networks[END_REF] have been conducted to characterize the human body as a propagation channel. A comparison between measured human body path loss founded in literature is shown in Fig. 2 [START_REF] Maity | Characterization of Human Body Forward Path Loss and Variability Effects in Voltage-Mode HBC[END_REF]. These measurements show that the human body path loss around 21MHz is distributed on a large range which is between 22dB and 62dB. This variability can be explained in part by the different experimental conditions used to carry out the measurements such as the coupling type (capacitive or galvanic), the electrodes type, the electrodes position, and the inter-electrodes distance. However, the different equipments used to perform the measurements (balun, couplers, cables, vector network analyzer, etc.) have a higher impact on the measured path loss, especially in the case of a capacitive coupling where the path loss highly depends of the close environment. Indeed, parasitic couplings can be created by the use of main-supplied equipments via the electrical earth, or just by the use of long wires. Therefore, to make the measurement under realistic conditions, a compact batterypowered transmitter capable of estimating the emitted power and a compact battery-powered receiver capable of estimating the received power, are required to properly characterize the path loss of the human body in the context of HBC.

For this reason, this work proposes a transmitter capable of estimating the emitted power for the HBC frequency band without external measurement devices. It is based on a microcontroller and is supplied by a battery to isolate it from the main. Finally, to allow the estimation of the emitted power, the Goertzel algorithm [START_REF] Goertzel | An Algorithm for the Evaluation of Finite Trigonometric Series[END_REF], which is a light discrete Fourier transform algorithm, has been implemented using particular parameters values.

The paper is organized as follows. In section II, the architecture of the proposed transmitter is presented. In section III, the Goertzel algorithm is described and the choice of its associated parameters is detailed. Finally, measurement results are shown in section IV.

II. DESIGN OF THE PROPOSED TRANSMITTER

A. Architecture of the proposed transmitter

The complete block diagram of the proposed transmitter is shown in the Fig. 3.a. A 21MHz square wave is also generated on a DSPIC microcontroller (MCU) pin and applied to the signal emitting electrode, the MCU being clocked at 84MHz using a 20MHz crystal oscillator. In addition, this signal is connected to the ADC MCU which is triggered by rising edges of a 480kHz signal generated by the DSPIC PWM peripheral (PWMH signal) to under-sample the 21MHz square wave at 480ksps. Next, the samples are used to estimate the power level emitted in dBm with help of Goertzel discrete Fourier transform algorithm. Finally, the power emitted is displayed on an OLED screen.

To overcome the bandwidth limitations of the ADC MCU pin which is equal to 0.5MHz, an external Sample & Hold circuit (S&H) which consists in a switch (74LVC1G3157) have been added as shown in the Fig. 3.b. This switch is controlled by a 480kHz signal generated by the MCU PWM peripheral (PWML signal). During the sampling phase, PWML is equal to 0 and the 21MHz square wave is connected to the storage capacitor C. During the hold phase, PWML is equal to 1 and the voltage stored in C is connected to the ADC input which enables the analog-to-digital conversion.

To ensure a synchronization between the hold phase and the analog-to-digital conversion, the PWMH signal which triggers the ADC is shifted a quarter period from the PWML signal which control the switch as shown in the Fig. 4.

B. Choice of storage capacity

The proposed external sample and hold circuit is controlled by a 480 kHz signal generated by the DSPIC PWML pin with a 50% duty cycle. During the sampling phase (PWML = 0), the 21MHz square wave is also transmitted to the storage capacity C across the switch which can be considered as a resistor Ron. Thus, during this phase, the circuit can be approximated by a simple RC circuit as shown in Fig. 5 [10] [START_REF] De Teyou | Statistical analysis of harmonic distortion in a differential bootstrapped sample and hold circuit[END_REF].

Considering that after 5τ the capacitor is fully charged (where τ = RonC is the time constant of the circuit), the condition to have a sufficiently fast charging time is expressed as follows:

5

(1)

which leads to:

(2)

where, TS is the sampling period. Since the typical value of Ron for the used switch is 6.5Ω, it also appears that C must be less than 32nF. During the hold phase (PWML = 1), the stored voltage is connected to the ADC input and it can be assumed that the capacitor discharges at a constant current Ileak which is equal to the leakage current of the ADC input as shown in Fig. 6 (Ileak=6.5µA from DSPIC datasheet). Thus, the condition to ensure that the capacitor discharge is lower than the ADC quantum q is: [START_REF] Callejón | Distributed Circuit Modeling of Galvanic and Capacitive Coupling for Intrabody Communication[END_REF] where Tc is the time necessary for analog-to-digital conversion (Tc=0.5µs from DSPIC datasheet) and q is the quantum which is equal to Vref /2 N (Vref = 3.3V and N = 10 bits). As result, C must be greater than 93pF.

Finally, it has been found in measurement that the optimal capacitor value is 120pF which fulfills the conditions (2) and (3).

III. GOERTZEL ALGORITHM

Goertzel algorithm [START_REF] Goertzel | An Algorithm for the Evaluation of Finite Trigonometric Series[END_REF] is an algorithm used in signal processing, which performs a discrete Fourier transform of a signal v[n] for a single frequency f. Therefore, it is proposed here to use this algorithm to estimate the power of the 21MHz carrier.

First, a parameter α is defined as follows: [START_REF] Seyedi | An empirical comparison of limb joint effects on capacitive and galvanic coupled intra-body communications[END_REF] where f is the nonzero frequency of interest and FS is the sampling frequency, the both being chosen to ensure that α is not a multiple of π which never happens if Nyquist-Shannon sampling theorem is respected. Then, the sequence g[n] is computed using N samples of the voltage v[n] such that:

1 ! 1 (5) 2 ! 2 # 2 cos 1 (6) ' ! ' # 2 cos ' ( 1 ( ' ( 2 (7)
Finally, magnitude and phase of the signal v[n] associated to the frequency f can be computed as follows:

A. Choice of sampling speed and number of samples

The sampling frequency FS and the number of samples N on which the Goertzel algorithm is applied must be carefully chosen. Indeed, FS can be expressed as a function of the frequency of interest f as follows:

E F G H ( 12
)
where p is an integer greater than or equal to 3 and q is an integer greater than or equal to 1 to ensure the convergence of Ag (resp. φg) to the magnitude (resp. the phase) associated to the signal v[n] at the frequency f in the Fourier's domain. In addition, using ( 12) in ( 4), α can be written as follows:

G I (13)
where p and q must be chosen so that α is not a multiple of π.

In other words, 2q/p must not be an integer to avoid that sinα = 0 which freezes the Goertzel algorithm in real domain (φg = 0). According to (4), this means for under-sampling condition (FS < 2f) that 2f must not be a multiple of FS.

Considering the proposed architecture, the Goertzel algorithm must be implemented in the DSPIC which generates the 21MHz square wave (21MHz being the standardized carrier frequency for HBC communications) and acquires the N samples. Therefore, the DSPIC oscillation frequency (Fosc) has been chosen as a multiple of f as follows:

H = LMN O (14)
which leads to Fosc = 84MHz since f is equal to 21MHz.

By associating ( 12) and ( 14), the following equation is obtained:

LMN O G ( 15
)
A way to guarantee that 2p/q is not an integer, as required by the Goertzel algorithm since α must not be a multiple of π, is to consider an odd number for 4p/q. Thus, a simple solution is to choose p equals 4 and q equals an odd number.

To determine q, it is now necessary to take into account the maximum sampling frequency of the DSPIC which is here 500ksps. Therefore, the odd integer q that enables to obtain the higher finite whole sampling frequency less than 500kHz is 175. As result, the obtained sampling frequency is 480kHz. Since the convergence between the Fourier series and the Goertzel algorithm is p-periodic for a noise-free signal, the algorithm must be applied for a true signal on N samples with:

P = Q R (16)
where r is an integer. To ensure the algorithm is stabilized, r is here taken equal to 10. As result, the number of samples obtained is 40.

B. Test of the Goertzel algorithm using MATLAB

The Goertzel algorithm has been first implemented with the parameters indicated above (f=21MHz and FS=480 kHz) using the MATLAB software and applied on ideal 21MHz square and sine signals between 0V and 3.3V sampled at 480kHz as shown in the Fig. 7.

To verify the accuracy of the Goertzel algorithm with theses under-sampling conditions, the magnitude Ag obtained using the Goertzel algorithm are compared in Tab. I to the magnitude associated to the frequency f in Fourier's domain. The latter can be obtained using the usual Fourier series and is equal to 4A/π for a square wave, and A for a sine wave (with A=1.65V for the both since waves are between 0V and 3.3V). An error of 11% is obtained for the wave due to aliasing of harmonic frequencies, which is acceptable considering the under-sampling condition. However, results perfectly match for the sine wave, but requires an additional band-pass filter to be obtained from a square wave as the one generated by the DSPIC. Finally, φg is not provided here but perfectly matches for the both waves when f is considered. The proposed transmitter has been implemented in a 5cm x 5cm PCB as shown in Fig. 8. It is based on a DSPIC supplied with a 3.3V battery and clocked at 84MHz using a 20MHz crystal oscillator. As indicated before, the DSPIC has been programmed to generate a 21MHz square wave and to under-sample it at 480ksps using an external sample & hold circuit controlled by the DSPIC too. To allow power estimation of the 21MHz carrier, the Goertzel discrete Fourier transform algorithm has been implemented as indicated in section III. To get an image of the emitted power, Ag is converted in dBm as shown in Fig. 8 using: STUVWX = 10 log 0\ :

]+ -= # 30 (17

)
where R is the fictive resistor equal to 1Ω. Finally, the 21MHz square wave is applied to a standard 3cm electrode used for electrocardiogram measurement and the ground electrode necessary for capacitive coupling consists of the PCB ground floor.

Fig. 9 shows a comparison between the values sampled with and without the external sample & hold circuit, which should be ideally 0 or 1023. As expected, it appears that the signal obtained using the external sample & hold circuit is closer to the ideal under-sampled signal than the one obtained without it. However, its Ag value leads to an error of 24% when compared to the one of the ideal under-sampled square wave as shown in Tab. II. This difference can be explained by the bandwidth of the external sample & circuit which is unfortunately not enough (even if better) to obtain an undersampled signal closer to the ideal under-sampled square wave since most of square harmonic frequencies are filtered. Nevertheless, the signal obtained using the external sample & hold looks like the ideal under-sampled sine wave and its Ag value leads to an error of only 6.7% when compared to the one of this sine wave as shown in Tab. III. V. CONCLUSION Human Body Communications (HBC), which has been standardized around 21MHz, potentially offers more favourable propagation conditions than usual wireless technologies such as ZigBee, or Bluetooth. However, due their sensitivity to the close environment, the signal attenuation cannot be accurately determined using lab equipment. Therefore, to make the measurement under realistic conditions, a battery-powered emitter with emitted power estimation capability has been fully described in this paper. The prototype implemented in a 5x5 cm PCB is supplied with a 3.3V battery and has been built around a MCU. The MCU has been programmed to generate a 21MHz square carrier and to under-sample it at 480ksps using the MCU embedded Analog-to-Digital Converter (ADC). To enable the carrier power estimation with this constraint, the Goertzel light discrete Fourier transform algorithm has been implemented using particular parameters values. Moreover, to overcome the bandwidth limitations of the ADC MCU pin, an external sample & hold circuit driven by the MCU have been added and optimized. However, a filter could be added in the designed transmitter to completely filter harmonic frequencies of the square wave and also to improve power estimation using Goertzel Algorithm when an undersampling is considered as in this work. Finally, to properly characterize the path loss of the human body, a compact receiver capable of estimating the received power has now to be designed.

Fig. 1 .

 1 Fig. 1. The two types of body-coupled communications: a) the galvanic coupling, b) the capacitive coupling [4].

Fig. 2 .

 2 Fig. 2. Comparison between the measures of the human body path loss found in [9].

Fig. 3 .

 3 Fig. 3. Block diagram of the proposed Transmitter: without external S&H (a) and with external S&H (b).

Fig. 4 .

 4 Fig. 4. 480kHz PWMH and PWML signals (1V/div and 2µs/div).

Fig. 5 .

 5 Fig. 5. Equivalent circuit of the external sample and hold circuit during the sampling phase.

Fig. 6 .

 6 Fig. 6. Equivalent circuit of the external sample and hold circuit during the hold phase.

Fig. 7 .

 7 Fig. 7. 21MHz square and sine waves between 0V and 3.3V sampled at 480kHz.

Fig. 8 .

 8 Fig. 8. Photographs of the designed transmitter.

Fig. 9 .

 9 Fig. 9. Comparison of the forty 10bit-samples when the 21MHz square wave is under-sampled at 480kHz.

TABLE I :

 I Magnitude comparison obtained with Goertzel algorithm and Fourier series.

				IV. MEASUREMENT RESULTS
		Goertzel Algorithm	Fourier series	Error
		using MATLAB (V)	(V)	(%)
	Under-sampled square wave	2.33	2.10	11.0
	Under-sampled sine wave	1.65	1.65	0

TABLE II :

 II Comparison of the Ag value obtained with the Goertzel algorithm using the designed prototype and an ideal under-sampled square wave.

		Without external S&H	With external S&H	Ideal under-sampled square wave (MATLAB)
	Ag (V)	0.84	1.76	2.33
	Error (%)	64%	24%	0

TABLE III :

 III Comparison of the Ag value obtained with the Goertzel algorithm using the designed prototype and an ideal under-sampled sine wave.

		Without external S&H	With external S&H	Ideal under-sampled sine wave (MATLAB)
	Ag (V)	0.84	1.76	1.65
	Error (%)	49%	6.7%	0