Arnaud Blouin
email: arnaud.blouin@eseo.fr

Olivier Beaudoux
email: olivier.beaudoux@eseo.fr

Malai : un Mod èle Conceptuel d'Interaction pour les Syst èmes Interactifs

Keywords: Interaction, action, instrument, IDM D.2.10 Design : Methodologies ; H.5.2 Information Interfaces and Presentation :Theory and methods Interaction, action, instrument, MDE

RESUME

Malgré l'évolution des modèles et des techniques d'interaction, les plate-formes d'implémentation de systèmes interactifs (SI) se focalisent sur la définition d'interfaces utilisateur (IU), ignorant la notion d'action et d'instrument. Cet article présente Malai, un modèle conceptuel d'interaction dédié au développement de systèmes interactifs. Malai vise à réunir différents principes du modèle d'action de Norman, de l'interaction instrumentale, de la manipulation directe, des interacteurs et du modèle DPI. Il complète nos travaux déjà réalisés sur les techniques de manipulation de données utilisées pour lier des données sources à des IU. A l'aide d'une étude de cas, nous présentons les quatre modèles composant Malai, à savoir : les modèles d'interface, d'action, d'interaction et d'instrument. A partir de ces modèles, le code du SI peut être généré dans une approche IDM (Ingénierie Des Modèles).

INTRODUCTION

La visualisation, la navigation et l'édition de données sont les tâches principales des Systèmes Interactifs (SI). Les plate-formes de développement modernes, tel que WPF [START_REF] Sells | Programming Windows Presentation Foundation[END_REF], formalisent la définition du lien entre les données sources et les interfaces utilisateurs (IU) par le biais du ≪ data binding ≫. Malgré l'évolution des techniques d'interaction, ces plate-formes restent toujours centrées sur le concept de widget ignorant ainsi les notions d'interaction, d'action et d'instrument. A un niveau plus abstrait, les environnements de développement fondés sur les modèles, tel que UsiXML [START_REF] Limbourg | UsiXML : a user interface description language for specifying multimodal user interfaces[END_REF], suivent, pour une majorité d'entre eux, un processus de développement composé de différents niveaux d'abstraction. Cependant, ils n'appliquent pas le principe de l'instrument, tandis que la description des interactions se limitent toujours à la notion d'évènement associé à un objet.

Le but de notre travail est de pouvoir générer une partie ou l'ensemble du code des SI à partir de modèles exhaustifs. Pour cela, nous proposons un modèle conceptuel pour la conception de SI manipulant des données. Celui-ci se compose de cinq modèles regroupés en deux parties. La première partie, appelée Malan, concerne le modèle des données sources que le SI manipule et son lien vers l'IU du SI [START_REF] Blouin | A mapping language for the data manipulation[END_REF]. La seconde partie, présentée dans cet article, consiste en un modèle conceptuel d'interaction nommé Malai, composé d'un modèle d'interface, d'un modèle d'action, d'un modèle d'interaction et d'un modèle d'instrument. Malai réunit des principes du modèle d'action de Norman [START_REF] Norman | User-Centered System Design : New Perspectives on Human-Computer Interaction[END_REF], de l'interaction instrumentale [START_REF] Beaudouin-Lafon | Instrumental interaction : An interaction model for designing post-WIMP interfaces[END_REF], de la manipulation directe [START_REF] Shneiderman | Direct manipulation : a step beyond programming languages[END_REF], du concept d'interacteur [START_REF] Myers | A new model for handling input[END_REF] et du modèle DPI [START_REF] Beaudoux | A toolkit for developing document-centered environments[END_REF]. Malai vise à retenir les avantages de ces modèles tout en apportant les améliorations suivantes :

-définir et fournir une librairie d'interactions prédéfinies pouvant être utilisées tel quel dans différents SI ;

-clarifier la définition du feedback intérimaire [START_REF] Myers | A new model for handling input[END_REF] des actions et des instruments ; -réutiliser une même action pour différentes IU qui utilisent le même modèle de données sources ; -clarifier l'implémentation du processus d'undo/redo des actions.

Le tout dans une approche IDM fondée sur trois des quatre composants du modèle Cameleon : l'interface abstraite (AUI) ne contient aucune information graphique du SI ; l'interface concrète (CUI) spécifie la disposition graphique des éléments de l'interface ; l'interface finale (FUI) correspond au code du SI pour une plate-forme d'application donnée [START_REF] Calvary | A unifying reference framework for multi-target user interfaces[END_REF] La plupart des frameworks actuels sont fondés sur le modèle MVC [START_REF] Krasner | A description of the model-view-controller user interface paradigm in smalltalk80 system[END_REF]. MVC (Modèle-Vue-Contrôleur) est un design pattern séparant les données sources manipulées par le système (le modèle), des vues présentant ces données à l'utilisateur, et des contrôleurs qui gèrent les évènements en provenance des périphériques d'entrée (HID -Human Input Devices). Cependant, MVC ne fournit ni modèle d'interaction ni modèle d'action, comme le résume la figure 1.

Le système Garnet a introduit la notion d'interacteur pour faciliter le développement d'IU, notamment en séparant les interactions des widgets [START_REF] Myers | A new model for handling input[END_REF]. Ce principe de séparation est utilisé dans des boîtes à outils afin de rendre des interacteurs indépendants des interactions et de leur contexte d'utilisation [START_REF] Crease | A toolkit of mechanism and context independent widgets[END_REF]. Proche de l'interacteur, l'interaction instrumentale définit un instrument comme étant la métaphore d'un outil qu'un utilisateur manipule pour réaliser des actions [START_REF] Beaudouin-Lafon | Instrumental interaction : An interaction model for designing post-WIMP interfaces[END_REF]. Un instrument se compose d'une partie physique définissant les interactions et les HID. Sa partie logique correspond à la représentation de l'instrument dans le SI. Fondé sur l'interaction instrumentale, DPI est un modèle conceptuel orienté documents utilisant la notion d'action [START_REF] Norman | User-Centered System Design : New Perspectives on Human-Computer Interaction[END_REF] dans un instrument [START_REF] Beaudoux | A toolkit for developing document-centered environments[END_REF] : la partie logique d'un instrument génère des actions consommées par l'IU.

Le formalisme ICO décrit le fonctionnement de SI à l'aide de réseaux de Petri [START_REF] Navarre | Structuring interactive systems specifications for executability and prototypability[END_REF]. Le fonctionnement de notre modèle d'instrument est assez proche du composant ObCS de ICO, permettant de décrire le comportement d'un objet interactif, dans le sens où des évènements HID sont utilisés en entrée et des services sont appelés en sortie. ICO ne considère cependant pas les actions et les interactions comme des objets de première classe en utilisant la notion d'évènement au lieu de la notion d'interaction et en appelant directement des méthodes du modèle de données pour modifier ces dernières.

Les récentes plate-formes de développement RIA (Rich Internet Applications), tel que WPF [START_REF] Sells | Programming Windows Presentation Foundation[END_REF], sont dédiées à la création d'applications classiques (de bureau) et pour le Web, à l'aide d'un langage de description d'IU (UIDL -User Interface Definition Language) et d'un langage de programmation. Cependant, ces plate-formes sont centrées sur la définition de l'IU et ne fournissent, par conséquent, pas de modèle d'action, d'interaction et d'instrument. Les environnements de développement fondés sur les modèles (MB-UIDE), qui suivent généralement les quatre niveaux de conception proposés par Cameleon, visent à faciliter l'adaptation d'un SI devant s'exécuter sur différentes plate-formes d'exécution. Ces MB-UIDE, tels qu'UsiXML [START_REF] Vanderdonckt | A MDA-compliant environment for developing user interfaces of information systems[END_REF] et UIML [START_REF]User interface markup language (UIML)[END_REF], ne proposent cependant pas de modèle d'instrument et utilisent toujours la notion d'évènement pour décrire les interactions. Bien qu'UsiXML dispose d'un modèle de tâche similaire sur certains points à notre modèle d'action, ce MB-UIDE ne considère pas les actions comme des objets possédant leur propre cycle de vie géré par les instruments et en relation avec des interactions. La notion de feedback pour les tâches n'est également pas abordée.

Au contraire, d'autres boîtes à outils issues de travaux de recherche se concentrent principalement sur l'interaction. ICON (Input CONfiguration) est notamment une boîte à outils permettant la configuration d'interactions physiques (souris, clavier, etc.) et leur connexion à une IU [START_REF] Dragicevic | Input device selection and interaction configuration with ICON[END_REF]. SwingStates est une librairie ajoutant à la boîte à outils Java Swing, la description d'interactions sous la forme de machines à états, remplac ¸ant ainsi les écouteurs habituellement utilisés [START_REF] Appert | SwingStates : adding state machines to Java and the Swing toolkit[END_REF].

VUE D'ENSEMBLE DU MODELE

Notre modèle conceptuel, dont l'organisation est illustrée par la figure 3, se compose de deux parties complémentaires. La première, appelée Malan, permet la définition du lien entre les données sources et les différentes présentations d'un SI, où une présentation affiche des données aux utili-

MODELE D'INTERACTION D éfinition

Une interaction est décrite par une machine déterministe à nombre fini d'états prenant en entrée des évènements HID pour en déduire l'état de sortie. Une interaction peut être implémentée sur une plate-forme d'exécution telle que SwingStates [START_REF] Appert | SwingStates : adding state machines to Java and the Swing toolkit[END_REF]. La modélisation à l'aide de machines à états fournit un moyen simple et efficace de décrire des comportements, en particulier des interactions hommemachine [START_REF] Appert | SwingStates : adding state machines to Java and the Swing toolkit[END_REF], et permet la spécification d'interactions complexes indépendamment des actions et des instruments des SI. Un développeur dispose ainsi d'une librairie d'interactions prédéfinies pouvant être facilement utilisées dans différents SI (p. ex. une interaction glisser-déposer). Il peut également définir de nouvelles interactions agrandissant ainsi cette librairie. La figure 6 présente le cycle de vie d'une interaction. Une interaction démarre lorsqu'un premier évènement conforme survient. Elle continue tant qu'un état final n'est pas atteint et tant qu'aucun évènement avortant n'apparait. Un évènement est dit avortant lorsqu'il stoppe l'interaction en cours (p. ex. l'appui sur la touche 'échap' lors d'un glisser-déposer). Ainsi, ce processus suit la recommandation de la manipulation directe stipulant qu'un uti-lisateur doit pouvoir arrêter toute interaction effectuée sur un système [START_REF] Shneiderman | Direct manipulation : a step beyond programming languages[END_REF].

Cycle de vie

Exemple

La figure 7 représente une machine à états décrivant une interaction glisser-déposer permettant l'utilisation du clavier. Cette interaction est utilisée par des instruments de l'exemple de l'arbre pour effectuer des actions, comme décrit dans les sections suivantes. Elle peut démarrer avec un évènement pression du dispositif de pointage, s'exécute tant que des évènements glissement surviennent, et se termine avec un évènement relâchement. Elle peut être avortée si l'utilisateur appuie sur la touche 'échap', ou si un évènement relâchement apparaît alors que l'état courant est pressé. Les évènements touchePressée et relâchementTouche, des états pressé et glissé, permettent l'utilisation du clavier pendant l'interaction. Le pseudo-code suivant est un extrait de la définition de l'interaction glisser-déposer présentant ses attributs (lignes 2 à 6) modifiés au cours de l'interaction lors des transitions d'états (lignes 9 à 20).

Sp écification d'actions

Une action est définie par une classe UML dans laquelle les attributs décrivent l'état de l'action. En utilisant l'exemple de l'arbre, l'action SélectionnerNoeud définit l'attribut sélection correspondant au noeud sélectionné de la présentation, comme l'illustre la figure 8.

Cycle de vie

La figure 9 correspond au cycle de vie d'une action. Il étend le cycle de vie une action du design pattern Commande dans lequel les actions sont des objets parametrables et dont les effets peuvent être annulée [START_REF] Freeman | Design Patterns[END_REF]. Bien que la définition d'une action soit indépendante des interactions et des instruments, le cycle de vie d'une action se déroule en relation avec celui de l'interaction, grâce à l'instrument qui leur sert de connecteur. Une fois exécutée et si elle est annulable, une action peut être défaite et refaite par un processus "undo/redo" standard.

Avortement et recyclage Il est possible d'avorter une action en cours. Dans ce cas, elle peut être recyclée en une action différente si certaines conditions sont respectées. Cela évite ainsi à l'utilisateur de recommencer le processus de création de cette dernière action. Cela permet également de pouvoir changer d'action au cours de l'interaction. Le recyclage est géré par l'instrument qui sert de lien entre l'interaction (et donc l'utilisateur) et l'action (l'interface et les données sources). Pour informer l'utilisateur de l'action en cours, et par conséquent du passage d'une action à une autre, l'instrument doit fournir un feedback intérimaire. La figure 10 décrit les liaisons pondérées par des conditions, gérées par un instrument, entre l'interaction ≪ glisserdéposer ≫ de la figure 7 et les actions DéplacerNoeud, SupprimerNoeud et DupliquerNoeud. Lorsqu'un utilisateur fait glisser un noeud d'un arbre, une action DéplacerNoeud est créée, étant donné que la condition ≪ pas de touche et cible =bouton Supp. ≫ est respectée. Si l'utilisateur appuie sur la touche ≪ ctrl ≫, l'action en cours DéplacerNoeud est alors avortée et recyclée en une action DupliquerNoeud puisque les paramètres de l'interaction correspondent maintenant à la condition ≪ touche pressée='ctrl' ≫. Enfin, si la cible du ≪ glisser-déposer ≫ est le bouton Supprimer de l'IU (voir la figure 5), l'action correspondante est maintenant du type SupprimerNoeud, recyclée à partir de l'action DupliquerNoeud. Ce processus continue jusqu'à ce que l'interaction se termine ou soit avortée.

Feedback int érimaire de l'action

Le feedback intérimaire consiste à fournir à l'utilisateur une réponse continue aux actions qu'il effectue sur le système [START_REF] Shneiderman | Designing the User Interface : Strategies for Effective Human-Computer Interaction[END_REF]. Cela se résume à présenter à l'utilisateur le résultat que provoquerait l'exécution de l'action courante [START_REF] Myers | A new model for handling input[END_REF]. Le feedback intérimaire d'une action peut être considéré indépendamment des instruments puisque que plusieurs instruments sont capables de créer une même action et que le feedback intérimaire de l'action reste toujours le même.

Le pseudo-code suivant définit le feedback intérimaire de l'action DéplacerNoeud (cf. figure 8). Il consiste à montrer à l'utilisateur le résultat du déplacement en cours en ajoutant un noeud dans l'arbre (la présentation) et à le rendre transparent. L'instrument est un ≪ médiateur entre l'utilisateur et les données. L'utilisateur agit sur un instrument qui transforme l'interaction réalisée par l'utilisateur en actions s'appliquant sur les données sources ou sur l'IU ≫ [2] (cf. figure 11). Les instruments fournissent un feedback intérimaire permettant à l'utilisateur de disposer des informations relatives à l'état de l'instrument et des actions que ce dernier peut permettre de créer. Un instrument se compose d'une partie physique décrivant les interactions utilisées en entrée, et d'une partie logique définissant la présentation de l'instrument dans l'IU. Un instrument est capable de créer plusieurs actions suivant le contexte courant (type de l'interaction, ses attributs, etc.). En fonction de l'état d'une interaction en cours qu'un instrument écoute, ce dernier crée, met à jour, exécute, termine, avorte ou recycle l'action correspondante.

Processus global

Un instrument est lié à un ensemble d'interactions dont il attend des changements d'états. Ces changements sont le démarrage, la mise à jour, la fin et l'avortement d'une interaction. Pour chacun de ces types de changement, un instrument peut :

-Fournir du feedback intérimaire. Le feedback intérimaire d'un instrument donne un aperc ¸u à l'utilisateur de l'évolution de l'interaction et de l'instrument. Par exemple, la figure 12 -Gérer (créer, avorter, mettre à jour, exécuter, recycler et terminer) l'action correspondante. L'état de l'action est déduit à partir d'un ensemble de paramètres, tels que le type ou les paramètres de l'interaction en cours. Nous considérons les boîtes de dialogue comme étant des instruments pouvant contenir d'autres instruments. Par exemple, une palette de couleur est la présentation concrète d'un instrument visant à choisir une couleur. L'affichage de la palette s'effectue en (dés-)activant l'instrument.

Exemple

La figure 13 présente un instrument décrivant en partie l'instrument Main de notre exemple de l'arbre. Le but de cet instrument est de manipuler les noeuds de l'arbre en appliquant le principe de la manipulation directe.

Cet instrument utilise deux interactions différentes : l'interaction glisser-déposer de la figure 7 et l'interaction Un instrument doit se mettre à jour à chaque changement d'état de l'interaction en cours (démarrage, évolution, fin et avortement). Le pseudo-code qui suit décrit les méthodes de l'instrument Main permettant ce processus.

IMPLEMENTATION

Notre implémentation de Malai est réalisée en Java et fait 5000 lignes de code (pour notre prototype actuel, commentaires exclus). En plus des classes nécessaires à la création d'instruments, d'interactions et d'actions spécifiques, notre prototype fournit un ensemble nonexhaustive d'interactions ainsi qu'un ensemble d'instruments et d'actions génériques. A l'aide de cette librairie, nous avons développé une application dédiée au dessin vectoriel (1300 lignes de code), ainsi que l'application présentée tout au long de cet article, permettant l'édition d'un fichier XML par le biais d'un arbre (1300 lignes de code) 1 . Au travers de l'éditeur de dessins, nous avons utilisé des interactions complexes telles que l'interaction bimanuelle pour redimensionner des formes, ainsi qu'une interaction multimodale composée d'un gyroscope (la Wiimote) contrôlé par la voix pour déplacer et pivoter des formes. Ces interactions ont été décrites et utilisées sans problème par des instruments via Malai. Le code de ces applications servira de patron pour la génération (semi-)automatique du code avec des outils IDM.

EVALUATION

Dans cette section, nous comparons tout d'abord Malai à d'autres architectures et évaluons ensuite notre architecture en utilisant certains critères tirés de [START_REF] Olsen | Evaluating user interface systems research[END_REF].

VIGO et M-CIU proposent respectivement, sur la base de l'interaction instrumentale, une architecture dédiée à la création d'IU distribuées [START_REF] Klokmose | VIGO : instrumental interaction in multi-surface environments[END_REF] et un modèle proposant d'unifier les techniques d'interaction du type glisserdéposer et ses dérivés [START_REF] Collomb | Extending dragand-drop to new interactive environments : A multidisplay, multi-instrument and multi-user approach[END_REF]. VIGO et M-CIU ne séparent pas les notions d'action et d'interaction de l'instrument. Un instrument est décrit par une unique interaction et peut être utilisé dans différents systèmes et IU. Dans Malai, c'est l'interaction qui est considérée comme indépendante et réutilisable et non l'instrument (des instruments génériques sont cependant fournis avec notre prototype pour être utilisés dans différents SI). VIGO et M-CIU n'utilisent pas d'action telle que définie dans Malai, perdant ainsi les avantages de notre modèle d'action, tels que le recyclage d'actions et la définition du feedback intérimaire d'une action. Cependant, ces inconvénients peuvent s'avérer avantageux en terme de simplicité d'utilisation comparés à notre modèle plus complexe à mettre en place mais qui permet la réutilisation d'interactions et d'actions. De plus, ils ne fournissent pas de langage dédié au lien entre les données et les présentations tel que notre langage Malan [START_REF] Blouin | A mapping language for the data manipulation[END_REF]. MPDC est une architecture qui étend MVC en sortant le processus de picking, récupérant des éléments d'une IU, du contrôleur, permettant à ce dernier d'être modulable pour différentes vues [START_REF] Conversy | Improving modularity of interactive software with the MDPC architecture[END_REF]. Les instruments et les interactions de notre modèle permettent cette modularité : les HID fournissent des paramètres aux instruments, lesquels les utilisent pour en déduire les éléments de l'IU concernés. De plus, MPDC n'utilise pas de langage dédié, comme notre langage Malan, pour lier les données à la présentation.

Nous évaluons maintenant Malai en utilisant quatre critères d'évaluation de SI définis par Olsen [START_REF] Olsen | Evaluating user interface systems research[END_REF] : la généralité de la solution proposée, son pouvoir d'expression, sa flexibilité et la proximité entre les moyens utilisés pour exprimer des choix de conception et le problème à résoudre. Concernant la généralité de la solution, Malai permet la définition de SI de manière générale mais n'aborde pas les notions d'application distribuée, de réalité mixte ou d'interaction multimodale, ce qui, par conséquent, limite le pouvoir d'expression de Malan. Cependant, l'utilisation de Malai dans le cadre de l'interaction multimodale semble envisageable si l'on considère les gestes, la voix, etc. comme des HID fournissant des évènements à des interactions au même titre que le clavier ou la souris. Le pouvoir d'expression de Malai est également limité par rapport à d'autres formalismes tels que ICO qui permet de décrire de manière formelle certaines parties des SI à l'aide de réseaux de Pétri facilitant ainsi la détection d'erreurs de conception [START_REF] Navarre | Structuring interactive systems specifications for executability and prototypability[END_REF]. Grâce à l'interaction instrumentale, Malai permet l'utilisation d'inter-actions WIMP et post-WIMP [START_REF] Beaudouin-Lafon | Instrumental interaction : An interaction model for designing post-WIMP interfaces[END_REF]. La flexibilité de Malai consiste en la possibilité de réutiliser des actions pour différentes interfaces utilisant des mêmes données (le feedback intérimaire de chaque action doit tout de même être réécrit) et l'indépendance des interactions, lesquelles peuvent être utilisées dans différents SI. Dans notre modèle, nous utilisons les concepts d'action, d'interaction et d'instrument fondés sur la métaphore de l'outil du monde réel : un utilisateur interagit à l'aide d'outils (les instruments) sur des objets afin d'effectuer des actions. Il est ainsi possible de développer une partie d'un SI tout en raisonnant de manière naturelle ce qui peut favoriser la proximité entre le problème et les moyens utilisés.

VERS UNE APPROCHE FONDEE SUR L'INGENIERIE DES MODELES

Elever le niveau d'abstraction lors de la définition de systèmes n'est pas un problème nouveau. Le langage de modélisation UML [START_REF] Rumbaugh | The Unified Modeling Language Reference Manual[END_REF] fournit un ensemble de diagrammes permettant la conception de manière abstraite de systèmes, dont le diagramme de classes utilisé, entre autres, pour générer automatiquement du code Java ou C#. L'ingénierie des modèles (IDM) [START_REF] Bézivin | On the unification power of models[END_REF] est un paradigme dont l'idée centrale est de considérer le cycle de développement d'un logiciel comme une chaîne de transformations de modèles de différents niveaux d'abstraction, où un modèle est une abstraction d'un système étudié. Des travaux en IHM reprennent ces principes en les appliquant aux problèmes de l'ingénierie [START_REF] Silva | User interface declarative models and development environments : A survey[END_REF][START_REF] Paternò | Authoring pervasive multimodal user interfaces[END_REF] des interfaces en se basant sur une architecture à quatre niveaux proposée par Cameleon [START_REF] Calvary | A unifying reference framework for multi-target user interfaces[END_REF] : le modèle de tâches, l'interface abstraite (indépendante de tout détail graphique), l'interface concrète (décrit les informations graphiques) et l'interface finale (le code du SI). Nos modèles Malai et Malan s'accordent avec cette architecture : -l'interface abstraite regroupe le modèle de données du SI, sa présentation abstraite, le lien entre ces deux derniers défini avec Malan, et le modèle d'action sans le feedback intérimaire des actions, ni les actions dépendantes de la présentation concrète ; -l'interface concrète complète l'interface abstraite en y ajoutant la présentation concrète, le lien entre les présentations abstraite et concrète, les instruments, les interactions, les feedbacks intérimaires des actions, et les actions dépendantes de la présentation concrète ; -l'interface finale sera générée grâce à des transformations de modèles prédéfinies utilisant en entrée l'interface concrète et un choix de plate-forme d'application (Java, .NET, etc.).

CONCLUSION ET PERSPECTIVES

Cet article présente Malai, un modèle d'interaction conceptuel dédié au développement de Systèmes Interactifs (SI). Malai se compose de quatre modèles : les modèles d'interface, d'instrument, d'action et d'interaction. Il complète nos travaux précédents sur le modèle de données et son lien avec l'interface [START_REF] Blouin | A mapping language for the data manipulation[END_REF]. Malai vise à mettre au profit les principes majeurs des interacteurs [START_REF] Myers | Past, present, and future of user interface software tools[END_REF], de l'interaction instrumentale [START_REF] Beaudouin-Lafon | Instrumental interaction : An interaction model for designing post-WIMP interfaces[END_REF], de la manipulation directe [START_REF] Shneiderman | Direct manipulation : a step beyond programming languages[END_REF], du modèle d'action de Norman [START_REF] Norman | User-Centered System Design : New Perspectives on Human-Computer Interaction[END_REF] et du modèle DPI [START_REF] Beaudoux | A toolkit for developing document-centered environments[END_REF]

FIGURE 1 :

 1 FIGURE 1 : Classification des travaux connexes.

FIGURE 4 :

 4 FIGURE 4 : Différences entre une interface utilisateur, une présentation et un instrument.

FIGURE 5 :

 5 FIGURE 5 : L'UI de l'exemple de l'arbre.

FIGURE 6 :

 6 FIGURE 6 : Le cycle de vie d'une interaction.

  !='echap' glissement pressionTouche | touche='echap' ou relâchement pressionTouche | touche='echap' pressionTouche | touche !='echap' ou relâchementTouche

FIGURE 7 :

 7 FIGURE 7 : Exemple d'une interaction glisser-déposer représentée par une machine à états.

FIGURE 8 :

 8 FIGURE 8 : Exemple d'actions de l'exemple de l'arbre.

FIGURE 9 :

 9 FIGURE 9 : Le cycle de vie d'une action.

  touche pressée='ctrl' GlisserDéposer p as d e to u ch e p re ss ée et SupprimerNoeud p as d e to u ch e p re ss ée et DupliquerNoeud DéplacerNoeud ci b le = b o u to n S u p p . instrument ci b le = b o u to n S u p p .

FIGURE 10 :

 10 FIGURE 10 : Exemple de recyclage d'actions.

1 2

 2 I n t e r im F e e d b a c k de DeplacerNoeud { DeplacerNoeud e r i m F e e d b a c k ( ) { 7 s i ( a c t i o n . p e u t F a i r e ( ) e t ! a c t i o n . f a i t ( ) ) { 8 E n t i e r p o s 9 NoeudUI c i b l e 10 s i ( noeudTemp== n u l l ) { 11 noeudTemp = nouveau NoeudUI ( a c t i o n . s é l e c t i o n ) 13 noeudTemp . d é f i n i r O p a c i t é ( 0 . 5 ) 14 } 15 s i ( a c t i o n . c i b l e . p e u t C o n t e n i r ( ) ) { 16 c i b l e = # o b t e n i r C i b l e ( NoeudUI , 17 a c t i o n . c i b l e l e = # o b t e n i r C i b l e ( NoeudUI , 21 a c t i o n . c i b l e . p a r e n t ) 22 p o s = c i b l e . p o s i t i o n ( a c t i o n . c i b l e ) 23 } 24 noeudTemp . d e p l a c e r ( c i b l e , p o s i t i o n ) 25 } 26 s i n o n noeudTemp . s u p p r i m e r (

FIGURE 11 :

 11 FIGURE 11 : Le passage d'une interaction à une action dans un instrument.

  présente une partie d'une IU d'un logiciel de dessin développé avec notre prototype. Cette IU montre que l'utilisateur réalise un glisser-déposer de la couleur du bouton Remplir, vers la forme ciblée. Dans cet exemple, le feedback intérimaire d'un instrument consiste à associer la couleur du bouton au curseur pendant le glisser-déposer. Ce processus permet à l'utilisateur d'être informé de l'état courant de l'instrument qu'il manipule et de l'action courante que de ce dernier peut créer.

FIGURE 12 :

 12 FIGURE 12 : Un exemple de feedback intérimaire d'un instrument.

  Un instrument peut être activé et désactivé. Par exemple, lorsqu'un noeud est sélectionné, l'instrument Suppression, qui gère la suppression des noeuds, est activé et vise versa. Un développeur dispose de deux manières de gérer l'activation d'un instrument. Tout d'abord, si aucune action que nécessite les actions que créé un instrument n'existe, alors l'instrument peut être désactivé. Par exemple, l'instrument Suppression ne peut produire que l'action SupprimerNoeud qui nécessite l'action SélectionnerNoeud ; durant l'exécution du SI, si aucun noeud n'est sélectionné, alors l'action Supprimer-Noeud ne peut exister et l'instrument Supprimer est désactivé. Ensuite, des actions génériques Activer-Instrument et DesactiverInstrument peuvent être créées par des instruments pour en (dés-)activer d'autres.

FIGURE 13 :s i n o n 23 c r é e r une a c t i o n DeplacerNoeud 24 }} 25 r e t o u r n e r r e c y c l e 26 }}

 1326 FIGURE 13 : Un exemple d'instrument.

  1 i n t e r a c t i o n D é m a r r e ( I n t e r a c t i o n i ) { 2 s i ( i n s t r u m e n t d é j à a c t i f ) f i n 3 s i ( i e s t P r e s s i o n B o u t o n e t 4 c i b l e de i e s t NoeudUI ou A r b r e U I ) { 5 c r é e r une a c t i o n Sé l e c t i o n n e r N o e u d 6 a c t i v e r l ' i n s t r u m e n t 7 } s i n o n s i ( i e s t G l i s s e r D e p o s e r e t 8 e x i s t e une a c t i o n S e l e c t i o n n e r N o e u d ) { 9 s i ( t o u c h e de i == n u l l e t s o u r c e e s t NoeudUI 10 e t c i b l e == i n s t r S u p p . b o u t S u p p ) 11 c r é e r une a c t i o n DeplacerNoeud 12 a c t i v e r l ' i n s t r u m e n t 13 } 14 } 15 i n t e r a c t i o n E v o l u e ( I n t e r a c t i o n i ) { 16 s i ( i n s t r u m e n t p a s a c t i f ou i ! = i n t e r ) f i n 17 s i ( ! r e c y c l a g e A c t i o n s ( i ) e t a c t i o n == n u l l e t 18 i e s t G l i s s e r D e p o s e r e t t o u c h e de i == ' c t r l ' 19 e t o b j e t de i e s t NoeudUI ) 20 c r é e r une a c t i o n DupliquerNoeud 21 m a j A c t i o n ( ) 22 } 23 i n t e r a c t i o n T e r m i n e ( I n t e r a c t i o n i ) { 24 s i ( i n s t r u m e n t p a s a c t i f ou i ! = i n t e r ) f i n 25 t e r m i n e r l ' a c t i o n en c o u r s 26 d é s a c t i v e r l ' i n s t r u m e n t 27 f e e d b a c k i n t é r i m a i r e 28 } 29 i n t e r a c t i o n A v o r t e ( I n t e r a c t i o n i ) { 30 s i ( i n s t r u m e n t p a s a c t i f ou i ! = i n t e r ) f i n 31 a v o r t e r l ' a c t i o n en c o u r s 32 d é s a c t i v e r l ' i n s t r u m e n t 33 f e e d b a c k i n t é r i m a i r e 34 }}

  . L'autre composant, les concepts et les tâches qui y sont liées, n'est pour l'instant pas abordé dans Malai. Après une présentation des travaux connexes, nous décrivons les modèles d'interface, d'action, d'interaction et d'instrument qui composent Malai. Un même exemple est utilisé tout au long de l'article pour illustrer chacun des modèles.

	TRAVAUX CONNEXES
		Données	Présentation	Action	Interaction	HID
	(p. ex. Swing) MVC	Modèle	Vue		Contrôleur
	DPI	Document	Présentation	Instrument	Instrument	Instrument
	frameworks RIA	Data binding	UIDL	Langage de programmation
	SwingStates	Java Swing	SwingStates	Java
	ICON	Java Swing		ICON
	UIML	Logic	Structure -Style -Content	Behavior

  Une action résulte du besoin pour un utilisateur ou un système de réaliser une tâche ou une partie d'une tâche soit sur des données sources via une IU, soit sur une IU sans modifier les données. Il existe deux types d'actions : les actions utilisateurs et les actions systèmes. Les actions utilisateurs sont produites par des instruments que manipule un utilisateur par le biais d'interactions. Au contraire, les actions systèmes sont lancées par le système lui-même et ne dépendent d'aucun instrument ou interaction (p. ex. la recherche de mises à jour d'un logiciel). La spécification d'une action s'effectue indépendamment des instruments qui peuvent la lancer. Notre modèle d'action diffère des modèles de tâche, tel que celui d'UsiXML, principalement du fait d'une différence de niveau d'abstraction : un modèle de tâche est généralement utilisé pour générer une interface abstraite, or notre modèle d'action opère au niveau de celle-ci. Un modèle d'action Malai enrichit un modèle de tâche en y apportant le feedback intérimaire et le code des méthodes faire, peutFaire, etc. des actions. De plus, une action est considérée comme un objet possédant son propre cycle de vie géré par l'instrument, comme le détaille les sections suivantes. Dans nos travaux futurs, nous envisageons de générer un modèle d'action à partir d'un modèle de tâche.

	MODELE D'ACTION
	D éfinition
	2 P o i n t	p o i n t I n i t i a l
	3 P o i n t p o i n t F i n a l
	4 Objet o b j e t C i b l e
	5 Objet o b j e t
	6 Touche t o u c h e
	7 / / . . .
	8
	9 E t a t p r e s s é{
	10	T r a n s i t i o n g l i s s e H I D -> g l i s s é{
	11	p o i n t F i n a l = g e t P o i n t ( )
	12	o b j e t C i b l e = g e t O b j e t ( p o i n t F i n a l )
	13	n o t i f i e r que l ' i n t e r a c t i o n é v o l u e
	14	}
	15
	16	T r a n s i t i o n p r e s s i o n T o u c h e = ' échap ' -> p r e s s é{
	17	t o u c h e = g e t T o u c h e ( )
	18	n o t i f i e r que l ' i n t e r a c t i o n é v o l u e
	19	}
	20	/ / . . .
	21 }
	22 }

1 I n t e r a c t i o n g l i s s e r -d é p o s e r {

  ce cas, les présentations doivent être partiellement mises à jour en actualisant uniquement leurs éléments concernés par les modifications. Par exemple, l'action SupprimerNoeud supprime un noeud du document source XML. Par conséquent, la présentation concrète (ArbreUI) doit supprimer son noeud correspondant au noeud du docu-

	3	Noeud p a r e n t
	4	E n t i e r p o s i t i o n
	5
	6	f a i r e ( ) {
	7	s i ( p e u t F a i r e ( ) ) {
	8	p a r e n t	= s é l e c t i o n . p a r e n t
	9	p o s i t i o n	= p a r e n t . p o s i t i o n ( noeud )
	10	# s u p p r i m e r ( p a r e n t . noeuds , s é l e c t i o n )
	11	}
	12	}
	13	p e u t F a i r e ( ) { r e t o u r n e r s é l e c t i o n ! = n u l l }
	14	d e f a i r e ( ) {
	15	# a j o u t e r ( p a r e n t . noeuds , s é l e c t i o n ,
	16		p o s i t i o n )
	17	}
	18	r e f a i r e ( ) {
	19	# s u p p r i m e r ( p a r e n t . noeuds , s é l e c t i o n )
	20	}
	21 }

ment XML, comme l'illustre le pseudo-code suivant, définissant l'action SupprimerNoeud. L'exécution de cette action (lignes 6 à 12) sauvegarde le noeud à supprimer, puis l'efface des données sources. Cette action est à même d'être défaite (ligne 14) et refaite (ligne 18), et nécessite une action SélectionnerNoeud pour exister (la variable sélection provient de l'action SélectionnerNoeud). 1 A c t i o n SupprimerNoeud n e c e s s i t e 2 S e l e c t i o n n e r N o e u d e s t Annulable {

  . Malai permet une définition claire du feedback intérimaire des actions et des instruments, et fournit un nouveau modèle d'action visant à faciliter le développement de SI notamment grâce au recyclage d'actions. En rendant les interactions indépendantes des SI, une liste extensible d'interactions peut être ainsi utilisée pour la définition d'instrument dans différents SI.Dans nos travaux futurs, des études seront réalisées pour comparer l'implémentation de Malai à d'autres boîtes à outils afin d'évaluer les bénéfices potentiels de notre modèle. Nous continuerons également l'utilisation de Malai et de Malan dans le contexte de l'ingénierie des modèles dans le but de générer des SI pour différentes plate-formes à partir de modèles abstraits et complets entre lesquels l'instrument jouera le rôle de pivot.

L'implémentation de Malai et les applications développées sont disponibles à l'adresse suivante : http ://gri.eseo.fr/software/malai/


Ali Kara Omar

Daniela Dragomirescu

Anthony Coustou

Robert Plana

A Kara-Omar

Low Power Ultra Wideband Up

I. INTRODUCTION

This paper reports the design of a low power UWB upconversion mixer, operating in the ISM UWB frequencies bands from 6 up to 8.5 GHz [START_REF] Razavi | A UWB CMOS transceiver[END_REF]. This work contributes at the design of an UWB-OFDM transceiver for wireless sensor network (WSN) for instrumentation measurements. This wireless sensor network intend to measure pressure and temperature in real time for aero-space applications [START_REF] Henaut | FPGA Based High Date Rate Radio Interfaces for Aerospace Wireless Sensor Systems[END_REF]. There are two main applications: satellite structure test (Figure 1.a) and in flight aircraft testing (Figure 1.b).

The first application is the ground test of the satellite structure before launching by using strain gauges or thermal sensors. For the second application, the in flight aircraft testing, the most critical test is the measurement of pressure around the wings to validate the designed computer model before industrialization, This wireless sensor network has to replace the wired system used in theses applications. The wired sensor measurement system presents strong drawbacks in term of weight and cost especially for aircraft due to aircraft immobilization for cable routing.

The main features of the new wireless sensor network are:

 high number of measurements points (more than 400 points); so an significantly increase in the number of used sensors  High data rate

 Low error rate

The most challenge for this system is the high data rate, thus the nodes are self-powered and cannot introduce sleeping mode to minimize power consumption. To respond to this challenge we propose an UWB transmitter using MB-OFDM [START_REF] Henaut | FPGA Based High Date Rate Radio Interfaces for Aerospace Wireless Sensor Systems[END_REF] and an RF front end between 6 GHz and 8.5 GHz. The architecture of the RF front-end transceiver is described in Figure 2. Transmitter (Tx) and receiver (Rx) channels are timely multiplexed by using an RF switch. Tx and Rx channels are driven by one quadrate local frequency synthesizer. The down-conversion mixers and VGA (variable gain amplifier) will provide I and Q base-band signals on Rx. Up-conversion mixers will provide RF signal on Tx channel, without the using of a power amplifier. Due to the usage of OFDM modulation technique, base-band signals spectrums of this transceiver, will be spread from 10 to 510 MHz.

The design of transceiver, with such a value of bandwidth, requires novel topologies of RF circuits in order to reach a high transmission quality (low distortion), which is a critical point of high speed telecommunications. In such an OFDM transmitter the up-conversion mixers, must be able to shift a 10 MHz up to 510 MHz signal around the 6 GHz to 8.5 GHz microwaves frequencies, with a low distortion. Due to components parasitic, and thus impedance mismatching effect, this condition is not obvious over a large frequency. This paper reports the design of such an up-conversion mixer, which can handle OFDM signals without these distortion effects by using a simple feedback loop device.

II. CIRCUIT DESIGN

The base-band signal, to transmit over the RF channel, will be provided by DAC device (see Figure 2). Due to parasitic behavior, capacitive effects appear on input impedance of transistor, which are responsible of impedance mismatching at high frequencies. Such phenomenon will be encountered on the design of up-conversion mixers requiring a very high bandwidth. This will result in amplitude and phase distortion of base-band signal. The amplitude distortion phenomenon will decrease the signal to noise ratio, on high frequencies of base-band signal's spectrum. This phenomenon can not be removed by digital filtering. This will result in an increase of mixer noise factor and thus a degradation of radio budget link. For these reasons, it's important to reach good impedance matching on IF (intermediate frequency) input of an up-conversion mixer, for an efficient OFDM transmission.

Due to the important value of bandwidth required on IF mixer's input (10-510 MHz), the usage of impedance matching devices based on lumped elements, is not possible. Our investigations on mixer design have shown that this problem can be overcome by using a feedback device in order to obtain good impedance matching over a large frequency band.

The operating principle and the quantitative analysis of this feedback device are discussed in the following paragraphs. The Figure 3 shows a simplified schematic and its electrical small signal model of the transconductance stage of the mixer with the feedback device. From Figure 3, we can extract the gain formula of the transconductance stage. This formula is reported below:

) ( .  j H K Av 
With K, which is a constant factor against frequencies variations done by the following equation:

    01 . 02 02 . 02 . . 01 . 1 1 1 . gm R Rg gm gm R gm R R gm K L F F L      
And H, which is the frequency transfer function :

2 3 2 2 2 1 02 . . 02 1 ) 1 ( 1 ) ( c c c F j j j gm R j j H                      
 gm 01 and gm 02 , are respectively Q 1 and Q 2 MOS transconductance. K is the static gain and ω c1 , ω c2 , ω c3 are the cut-off frequencies. The input impedance formula is given by: ) ( .

1 , 1  j H K Zin Zin Rg Ie Eg Zin     (1) with:                      01 01 . 02 . 02 1 ) ( and 1 1 . 1    j j H gm R gm R gm K L F
From equation (1), we can extract an equivalent circuit of mixer's input stage and its input impedance. The input impedance of mixer can be modeled by a resistor R 0 in series with an inductance L 0 . 01 0 0 01 .

.

02

0 1 1 . 1  R L and gm R gm R gm R L F            
From these formulas, real and imaginary parts of input impedance can be simplified to the following expressions: ). This will improve linearity and conversion gain of the mixer at the same time.

The simulations of electrical behavior of this mixer, performed with Cadence Spectre software, are depicted on the next section of this paper.

III. SIMULATIONS RESULTS

The simulations conditions of this investigation are the following: the IF signal power is set to -20 dBm and LO power to -3 dBm.

The Figure 6 depicts the input return loss on IF input of mixer. Figure 7 reports the conversion gain versus power level of local oscillator, at a 6 GHz frequency. The conversion gain reaches a maximum of 0 dB for a local oscillator power between -4 and -3 dBm. We have chosen to work at a -3 dBm level, concerning the LO signal, for DC power consumption reasons, which are very important in our application. Reducing the required power level of the local signal to apply on a mixer is equivalent to reduce the required DC power by the local oscillator (a VCO), which will drive the mixer. This point is important for the development of a low DC power consumption OFDM transceiver.

We have simulated the conversion gain versus the RF frequency. This result is reported on Figure 8. During this simulation, the power level of the local oscillator was maintained to -3 dBm and the IF signal frequency was fixed to 500 MHz. In order to compare our results to the state of the art, we have reported several previously published data in the field of CMOS up-conversion mixers. These data are reported on table 1 as well as data of our design.

Because the available designs, don't have common characteristics (conversion gain, bandwidth, operating frequencies), we have used a Figure of merit in order to be able to compare each design with the others. Such a factor has been already defined in [START_REF] Liao | Lossy Lc Ladder Matching Network For Ultra-Wideband Cmos Gilbert Cell Mixer Design[END_REF] In Figure 10, we present the Figure of merit, versus the DC power consumption; for our work and all of the mixers listed on table 1. We can note that our mixer exhibits a good trade-off between DC power consumption and conversion gain. Our design has a DC power consumption of 4.2 mW and a conversion gain in the range of 0 dB. this work [START_REF] Huang | A Miniature, Folded-Switching, Up-conversion Mixer for UWB Applications Using 0.18-um CMOS Process[END_REF] [ Concerning the power handling capabilities, our mixer is matched to the condition of our targeted architecture (not use of power amplifier in our transmitter architecture). Thus, our mixer must be able to handle the required power to transmit information over the RF channel.

The FCC allowed a maximum power spectral density radiation of -41.3dBm/MHz [10], at the antenna level. For a 500 MHz bandwidth, the maximum transmit-power is: P EMAX =-41.3+10 log (500) = -14.3 dBm Our circuit exhibits an IIP1 in the range of -11.6 dBm, which satisfies the FCC condition, if we plan to connect directly antenna filter and multiplexing switch on our mixer's output.

IV. LAYOUT DESIGN

The layout of the mixer without the ground plane is presented in Figure 11. DRC test and LVS are used to check our circuit and we performed the back-annotation in order to include the extra interconnect parasitic elements.

The mixer is designed using UMC 0.13 µm RF CMOS process provided by United Microelectronics Corporation (UMC) The die size including RF-pads is 0.64 mm 2 . In conclusion, we have presented a low DC power, ultra wide band, up-conversion mixer. This mixer consumes only 2.34 mA current under a DC voltage of 1.8 V, which is the best result in the state of the art. A new topology of mixer, based on the Gilbert structure, was proposed here to obtain a good input impedance matching condition over a large frequency range. This condition is reached at a very low cost: the proposed feedback device requires a very low silicon surface and exhibits a very low DC power consumption. This matching impedance condition is important in order to remove any amplitude distortion effects, which induces a degradation of the signal to noise ratio.

Figure 1 :

 1 Figure 1: WSN applications for in flight aircrafts testing and ground satellite testing

Figure 2 :

 2 Figure 2: Transceiver Architecture

Figure 3 :

 3 Figure 3: (a) Simplified schematic of the transconductance stage, (b) Small signal model of the transconductance stage

RgmFigure 4 :Figure 5 :

 45 Figure 4: Schematic of the mixer A transconductance stage (Q 1 -Q 2 ) drives the switching stage (Q 3 -Q 8 ) by using an IF differential input signal. A feedback loop (Q 9 -Q 10 ) is added to the transconductance stage in order to match input impedance over a 10-510 MHz frequency range.Transistors Q 7 and Q 8 are used as current injection source to increase conversion gain[START_REF] Maceachern | A charge-injection method for Gilbert cell biasing[END_REF]. The principle is described in Figure5. As shown, a current injection source (I CR ) is added to the basic core of the mixer, at the drain of Q 1 and Q 2 .

Figure 6 :

 6 Figure 6: The input return loss

Figure 7 :

 7 Figure 7: Conversion gain versus LO power at 6.5 GHz frequency with IF power of -20 dBm.

Figure 8 :

 8 Figure 8: Conversion gain versus RF frequency with IF frequency of 500 MHz and IF power of -20 dBm, the LO power is -3 dBm.

Figure 9

 9 Figure 9 report the simulation results of the noise factor of our mixer. The noise factor level is around of 7 dB.

Figure 9 :

 9 Figure 9: Noise factor in dB



  In this factor of merit, the definition area is limited to positive values of conversion gain and the required power level of local signal, for efficient mixing, is not taken into account. For these reasons, we propose a modified definition of this Figure of merit. This new definition is given below: is the conversion gain in dB, BW is the bandwidth of the RF output.

Figure 10 :

 10 Figure 10: Figure of merit versus power consumption for our work and the state of the art samples

Figure 11 :

 11 Figure 11: Layout view of the mixer without ground plane V. CONCLUSIONS

Table 1 :

 1 State of art of up-conversion mixers

	6]	[7]	[8]	[9]


A Lecointre

D Dragomirescu

R Plana

A Largely Reconfigurable Impulse Radio UWB Transceiver

This letter presents a new and efficient digital baseband architecture for impulse radio ultra wideband (IR-UWB) transceivers with reconfigurable capacities for wireless sensor network (WSN) applications. An implementation is done on FPGA for validation by measurements. It achieves the highest data rate and the largest number of operating modes for an IR-UWB reconfigurable transceiver, to the best of the author's knowledge. Measurements demonstrate reconfigurability in data rate (from 4 to 125 MBits/s), chip duration (from 8 to 32 ns), pulse duration (from 1 to 4 ns), bit duration (from 8 to 256 ns), processing gain (from 1 to 8 pulses/bit), occupied bandwidth, radio range, BER performance, synchronization accuracy (from 0.5 to 4 ns), pulse amplitude, transmitted power density, duty cycle (from 3 to 50 %), power consumption, spectrum occupation, and maximum supported UWB channel delay spread (from 4 to 31 ns).

Introduction:

IR-UWB is well known to be a radio technology which answers WSN requirements thanks to the use of low duty cycle transmissions [START_REF] Yang | Ultra-wideband communications, an idea whose time has come[END_REF]. IR-UWB advantages' are low size, low complexity and low cost, as well as the possibility of low power implementation [START_REF] Yang | Ultra-wideband communications, an idea whose time has come[END_REF]. This letter presents a dynamically reconfigurable IR-UWB digital baseband transceiver, for fulfilling efficiently the large needs of WSN applications, at the best cost at any time even with some changes in the operating conditions, the local regulation, or the radio channel behavior.

Implemented IR-UWB transceivers reconfigurable in multiple properties are rare in literature. Reference [START_REF] Anderson | A software defined ultra wideband communication system testbed[END_REF] demonstrates a pulse modulation and data rate reconfigurable FPGA based IR-UWB transceiver. Reference [START_REF] O'donnel | A flexible, low power, DC-1GHz impulse-UWB transceiver front-end[END_REF] presents an IR-UWB transceiver reconfigurable in data rate (up to 30 Mbits/s), pulse duration, occupied bandwidth and modulation. The most complete IR-UWB reconfigurable transceiver is introduced in [START_REF] Verhelst | A low power, reconfigurable IR-UWB system[END_REF]. Its reconfigurability is in data rate (up to 50 Mbits/s), processing gain, and synchronization algorithm. Thus it has the ability to adjust its radio range, bit error rate (BER) performance, power consumption and spectrum occupation.

The proposed IR-UWB transceiver:

The proposed IR-UWB reconfigurable digital baseband is implemented on FPGA using VHDL. Analog-to-digital and digital-to-analog converters (ADC/DAC) are used for interfacing the digital baseband and the radio frequency (RF) front-end. The proposed transceiver uses BPSK modulation and a direct spread spectrum technique. Thanks to its time domain signal processing, IR-UWB is well suited for a digital implementation, and thus lends itself to the implementation of reconfigurability. Each property (the bit duration T b , the chip duration T c , the pulse duration T p , the synchronization accuracy S a ) of the transceiver can be defined by reconfigurable input parameters M* x (with x = T b , T c , T p , S a ) and with the time resolution of the transceiver r. The time resolution is defined by the ADC/DAC performance. For example, the reconfigurability in chip duration T c is expressed as

T c = M * Tc x r. (1)
By adjusting the reconfigurable input parameter values M* x , the transceiver can dynamically change its properties. This scheme is applied for the pulse duration, the chip duration, the bit duration and the synchronization accuracy.

These four basic reconfigurability parameters imply also reconfigurability in data rate, processing gain, BER performance, synchronization performance, duty cycle, power consumption, spectrum shape, supported channel delay spread, and radio range. In addition, by being able to change the output pulse amplitude, reconfigurability in emitted power and in compliance with local regulations is added to the IR-UWB transceiver.

The digital baseband transmitter is designed for driving a high speed time interleaved DAC. The transmitter comprises a data generator block which generates a frame with a start frame delimiter (SFD) and a payload data field.

A BPSK pulse generator and a spreading block are also implemented. The

Figure captions:

Fig. 1 The proposed reconfigurable BPSK IR-UWB receiver digital baseband implementation.

Fig. 2 Pulse duration, pulse amplitude, occupied bandwidth and power spectral density reconfigurability measurements.

  spreading block is dedicated to map the data with the spread spectrum sequence. The BPSK pulse generator drives the DAC with a signal of length T p (pulse duration) at either the positive full scale range or the negative full scale range.The digital baseband receiver implementation is presented in the fig.1. The proposed reconfigurable BPSK IR-UWB receiver is a coherent Rake-like receiver with parallel search frame level synchronization on N bins. These N bins are used for covering the synchronization search space. The length of a bin is equal to the synchronization accuracy S a . For each bin, the receiver dedicates two cross correlators, an energy comparator, a de-spreading block and a SFD code correlator to carry out a first bit demodulation and the frame level synchronization. In function of the N SFD code correlators outputs, N sync outputs are considered as frame level synchronized outputs and can be seen as fingers of the Rake. From the N bins, the receiver selects N sync fingers on a frame level synchronization criteria. A vote procedure combines the N sync correlators outputs for improving the signal to noise ratio and the BER performance. The output bit value is equal to the value of the majority of the N sync correlators outputs. It's a high level (bit level after frame synchronization) combining technique.To allow reconfigurability based on a time domain signal processing, a chip duration retiming block, a clock generator block are necessary as well as a block for dealing with the reconfigurability parameters M* x , at both the transmitter and the receiver side.Reconfigurability measurements results:Due to the higher complexity of the receiver with respect to the transmitter, the overall performance is limited by the receiver implementation. The synchronization accuracy, the chip duration and the pulse duration of the transceiver directly impact the size of the circuit. The number and the range of features are limited by the hardware, i.e. the size of the FPGA, and the sampling frequency of the DAC and ADC.Two Xilinx Virtex4lx15 FPGAs are used for baseband implementation, one for the transmitter and one for the receiver. A 1 GSPS DAC, a 2 GSPS ADC and RF blocks are commercial off-the-shelf (COTS) components. The RF front-end contains LNA, PA, filters, antennas and up-and down-conversion stages centred at 3.9 GHz. Measurements are made in an industrial environment.The transmitter-receiver distance is 4 meters. The SNR is changed thanks to digital step attenuator in the receiver RF front-end. The measurements in fig.2 to 4 show the vast reconfigurability (48 distinct operating modes) of the implemented IR-UWB transceiver. The implemented transceiver achieves reconfigurability in chip duration (8 to 32 ns), bit duration (8 to 256 ns), data rate (4 to 125 Mbits/s), processing gain (1 to 8 pulses/bit), pulse duration (1 to 4 ns) (fig.2), pulse amplitude (25 to 200 mV) (fig.2), occupied bandwidth (250MHz to 1 GHz) (fig.2), transmitted power density (fig.2), radio range (fig.2), BER performances (fig.3), synchronization accuracy (0.5 to 4 ns) (fig.3), synchronization performances (fig.3), duty cycle (3 to 50 %), power consumption (fig.4), and maximum supported UWB channel delay spread (4 to 31 ns). Fig.3presents the reconfigurability in synchronization accuracy, demodulation performance (BER) and synchronization performance, versus the E b /N 0 measured at the FPGA receiver input. By reconfiguring its synchronization accuracy itself from 4 to 0.5 ns, the proposed transceiver achieves a gain of 6 dB in E b /N 0 for the synchronization, and a gain of 5 dB in E b /N 0 for a 6.10 -5 BER level. The radio range is thus also reconfigurable. Link budget calculations show distinct ways for using the available dB obtained thanks to reconfigurability. For example, as illustrated by the fig.4, the 4 ns synchronization accuracy allows a 72 mW decrease of the FPGA receiver dynamic power consumption in comparison with the 0.5 ns synchronization accuracy. These measurements demonstrate the validity of the proposed reconfigurable IR-UWB digital baseband transceiver architecture. The transceiver offers a larger range of reconfigurable properties and a higher data rate than the transceivers presented in [2]-[3]-[4]. Conclusion: A new and efficient IR-UWB reconfigurable transceiver architecture for digital baseband is proposed. Validation implementation is done on FPGA. Measurements show that the implemented reconfigurable IR-UWB transceiver achieves the largest number of reconfigurable properties and the highest data rate, i.e. 125 Mbits/s, for a reconfigurable UWB transceiver to the best author's knowledge. It allows to the radio to use its best operating parameters for answering any kind of change in the environment (channel response, communication range, energy available, local regulation, and applications needs). The proposed transceiver architecture and implementation are well suited for wireless sensor networks applications with time-variant operating performance requirements.

Fig. 3

 3 Fig. 3 Synchronization and BER performances reconfigurability measurements in function of the synchronization accuracy and the number of channels N over a multipath indoor channel.

Fig. 4

 4 Fig. 4 Dynamic power consumption reconfigurability measurements of the digital baseband for both the transmitter and the receiver side.

  Figure 1 2 GSPS TI-ADC

  Figure 4


Mariano Ercoli
email: mercoli@laas.fr

Michael Kraemer
email: mkraemer@laas.fr

Daniela Dragomirescu
email: daniela@laas.fr

Robert Plana
email: plana@laas.fr

An Ultra Small Passive Balun for 60 GHz Applications in CMOS 65nm Technology

Keywords: Marchand Balun, Coupled Line, 60GHz Band, CMOS

In this paper a novel method to design a miniaturized Marchand Balun is proposed. The main issue in the balun design is to limit its physical dimension. This work shows a device that achieves good performance, in terms of phase and amplitude balance, and reduced size at the same time. To allow such a result, an accurate analysis of the electromagnetics phenomena was done. The principal equations that characterize the structure are resumed in the paper. The device has been fabricated in 65 nm CMOS technology. The balun has been designed for application in a wide band low power 60 GHz transceiver.

I. INTRODUCTION

In the last decade the development of wireless applications has had a rapid growth. This trend has interested especially the 60 GHz unlicensed band that offers a very large applications scenario. The need for cheap, small and at the same time very efficient transceivers has pushed the researcher to find new solution or to improve the actual resources. Moreover the miniaturization of the circuits, due to the scaling of the technology nodes of the transistors, promoted the employment of differential circuits to realize the fundamentals blocks on the transceivers. The balun is a fundamental component for this type of architecture. This device finds wide application in antennas, push-pull amplifiers, balanced mixers, etc. If necessary such a device can be designed to achieve impedance matching the other blocks [START_REF] Kidwai | Fully integrated 23dBm transmit chain with on-chip power amplifier and balun for 802.11a application in standard[END_REF] [START_REF] Van Vorst | Low-power 1V 5.8 GHz bulk-driven mixer with on-chip balun in 0.18m CMOS[END_REF]. The employment of differential architectures on RF front-end exhibits many attractive characteristics. For the double balanced mixer, the differential circuit configuration increases the RF-to-IF and LO-to-IF isolation [START_REF] Komoni | Modeling, simulation and implementation of a passive mixer in 130nm CMOS technology and scaling issues for future technologies[END_REF] and at the same time it maximizes the common mode rejection. On the other side, in the push-pull amplifier configuration its use increases the linearity and power efficiency of the amplifier. The advantages shown are directly correlated to the performances of the balun. If its behaviour is far from ideal, the performance of the entire system fails. The principal features employed to estimate the quality of the balun are the amplitude balance and the phase balance. The balun can be realized in two different manners: active and passive configurations. For our application, we will focus our attention on passive baluns to obtain a low power circuits.

II. MARCHAND BALUN THEORY

Regarding the Radio Frequency Integrated Circuit (RFIC) design, the principal issue about the use of this component is its very large occupied area compared with the active circuit parts. In literature, many architectural solutions for the balun design are available, and a large part of those have as objective the reduction of the physical dimension. Table 1 summarizes the state of the art of these devices. The various typologies of baluns taken into account are realized on different technologies (Si, SiGe and GaAs). The interest of this work is focused on the geometrical realization. The different substrate typologies influence the total dimension through the relative permittivity constant (ε r ), but by an efficient architectural solution it is possible to achieve the minimum area in all types of substrate. The cost of the chip fabrication increases proportionally with the area, thus the miniaturization of this component is extremely important. The different topologies of passive baluns can be classified in two different categories: Classical Marchand Balun [START_REF] Ng | Analysis and design of a high-performance planar Marchand balun[END_REF][START_REF] Liu | A 60-GHz Millimeter-wave CMOS Marchand Balun[END_REF], with its derivations, [START_REF] Ang | A compact MMIC balun using spiral transformers[END_REF][START_REF] Zhang | A new planar Marchand balun[END_REF][START_REF] Sun | Design and Realization of Spiral Marchand Balun Using CPW Multilayer GaAs Technology[END_REF][START_REF] Ma | A Miniaturized Multilayered Marchand Balun Using Coupled Artificial Transmission Lines[END_REF][START_REF] Lai | An Integrated 20-26 GHz CMOS Up-Conversion Mixer with Low Power Consumption[END_REF] and Conventional transformer [START_REF] Felic | An Integrated Transformer Balun for 60 GHz Silicon RF IC Design, Signals, Systems and Electronics[END_REF]. On silicon bulk CMOS technology the effective permittivity constant ε ef f is about 4.2 as a consequence the wavelength (λ) at 60 GHz is 2.6 mm, so λ/4 = 650 µm. To avoid this large waste of area, literature proposes different solutions: spiral approach [START_REF] Ang | A compact MMIC balun using spiral transformers[END_REF] or meandered displacement [START_REF] Liu | A 60-GHz Millimeter-wave CMOS Marchand Balun[END_REF][START_REF] Lien | Analysis and Design of Reduced-Size Marchand Rat-Race Hybrid for Millimeter-Wave Compact Balanced Mixers in 130-nm CMOS Process, Microwave Theory and Techniques[END_REF]. However the problem is not completely solved since the dimension of the structure is still too big. In this work the different solutions shown in Table 1 have been analised to understand which possible architecture can be employed to further improve its response. The operating principle for the proposed balun is based on the classical Marchand balun design [START_REF] Marchand | Transmission-line conversion transformer[END_REF]. The difference between the classical and the proposed structure is the dimension of the coupled lines that form the balun. The Marchand balun architecture is realized by a pair of coupled lines with a λ/4 length. In this new solution the lines's length have been reduced. To achieve a response comparable with the one of the Marchand balun it has been necessary to understand the physics phenomena of the structure. The analysis started by a pair of general equation that model the Marchand Balun [START_REF] Ang | A compact MMIC balun using spiral transformers[END_REF]:

S 21,bal = -T C + T 3 C 1 + C 2 (1)
S 31,bal = T C - T 3 C 1 + C 2 = -S 21 (2)
Where T and C represent respectively the transmitted and the coupled power into the balun (bal). To understand the [START_REF] Liu | A 60-GHz Millimeter-wave CMOS Marchand Balun[END_REF] 25-65 <10 <10* 1.5 < ±10 0.55 [START_REF] Sun | Design and Realization of Spiral Marchand Balun Using CPW Multilayer GaAs Technology[END_REF] 14-28 <1.5 <15 1 < ±10 0.26 [START_REF] Lai | An Integrated 20-26 GHz CMOS Up-Conversion Mixer with Low Power Consumption[END_REF] 20-26 <2 N.A. 0.5 Max 5 0.05 [START_REF] Felic | An Integrated Transformer Balun for 60 GHz Silicon RF IC Design, Signals, Systems and Electronics[END_REF] 50-65 3 N.A. <3 Max 5 0.05 [START_REF] Van Vorst | Low-power 1V 5.8 GHz bulk-driven mixer with on-chip balun in 0.18m CMOS[END_REF] 20-70 1 N.A. 1 5 0.06 [START_REF] Hamed | A monolithic double-balanced direct conversion mixer with an integrated wideband passive balun, Solid-State Circuits[END_REF] 15-45 1.5 <13 1 5.5 0.40 [START_REF] Yu | K-band balun with slot pattern ground for wide operation using 0.18 m CMOS technology[END_REF] 18-32 0.8 <12 1 5 0.07 [START_REF] Chiou | Low-Loss and Broadband Asymmetric Broadside-Coupled Balun for Mixer Design in 0.18-µm CMOS Technology, Microwave Theory and Techniques[END_REF] 27-54 1.1 <15 <1 5 0.06 functioning of the balun is necessary to know the distribution of principal power flows of the structure. Equations ( 1) and ( 2) summarize the different contributions that came out at the balanced ports but do not show which is the importance of each coefficient. This can be explained using Figure 1, which shows the propagation of power into the structure at different time steps. The values of the coupling and transmission factor into the two couplers must be chosen carefully. For example, a too high value for the C1 factor is counter-productive because the power reflected by the short R is reflected back to the input. The same problem occurs for a too low value of C1: in this condition the power transmitted on Port 2 is too low. For the project described in this paper the Fig. 1. Balun with its principal power flows coupled lines theory [START_REF] Sorrentino | Ingegneria delle microonde e radiofrequenze[END_REF] is employed to model the couplers that constitute the balun. The model of the coupled lines is based on the field mode analysis. It is known that two different modes exist on the coupled structure: the even and odd modes. The superposition of those two groups of modes influences the coupler behaviour and therefore the balun response. Equation (3) describes the influence that the two modes have on the output imbalance [START_REF] Liu | A 60-GHz Millimeter-wave CMOS Marchand Balun[END_REF].

S 21,bal S 31,bal = Z 0e -Z 0o Z 0e + Z 0o (3)
The coefficients Z 0e,o are impedances of the even and odd modes respectively.

Z 0e = L + M C (4)
Z 0o = L -M C + 2C m (5)
For the single coupled line, equation ( 3) can be used to characterize the coupling factor:

C = Z 0e -Z 0o Z 0e + Z 0o (6)
In the silicon bulk technology limits exist for the maximum value for the even mode impedance. Its value is limited by the parasitic capacitance between the lines and the ground [START_REF] Hamed | A monolithic double-balanced direct conversion mixer with an integrated wideband passive balun, Solid-State Circuits[END_REF]. A reasonable value for this parameter is around 100 to 150 Ohm [START_REF] Lien | Analysis and Design of Reduced-Size Marchand Rat-Race Hybrid for Millimeter-Wave Compact Balanced Mixers in 130-nm CMOS Process, Microwave Theory and Techniques[END_REF]. Such a limit for the maximum even impedance moves the attention to the optimization of the other parameters: the mutual capacitance C m and the mutual inductance M between the coupled lines become fundamental design variables [START_REF] Liu | A 60-GHz Millimeter-wave CMOS Marchand Balun[END_REF].

Moreover the even and odd mode impedances are important in terms of matching impedance among the balun and the other devices. For the coupled line model, a relation holds [START_REF] Sorrentino | Ingegneria delle microonde e radiofrequenze[END_REF] for the even and odd mode impedances to achieve a low return loss:

Z 0e * Z 0o = Z 2 o ( 7
)
where Z 2 o is the characteristics line impedance. To take into account the dimension of the coupler's line a set of equations are well explained in [START_REF] Sorrentino | Ingegneria delle microonde e radiofrequenze[END_REF]. The principal equations, employed to show the power propagation are:

S 12 = √ 1 -C 2 √ 1 -C 2 cos θ + j sin θ (8)
S 13 = jC sin θ √ 1 -C 2 cos θ + j sin θ (9)
In those equations the S parameters are referred to a single coupler device. Therefore they can be employed to model the parameters shown in the (1,2):

S 21,bal = - jC1 sin θ T 1 2 cos θ + j sin θ T 1 T 1 cos θ + j sin θ + ... (10)
S 13,bal = T 1 T 1 cos θ + j sin θ jC sin θ T 2 cos θ + j sin θ -... [START_REF] Felic | An Integrated Transformer Balun for 60 GHz Silicon RF IC Design, Signals, Systems and Electronics[END_REF] where T = √ 1 -C 2 . The extended formula was stopped at fist step of approximation to show the complexity of the calculus. Moreover, in the case of which the electrical length of the two couplers are different there will be two different values of θ , which add further complexity to the problem.

III. LAYOUT REALIZATION

Taking into account the consideration exposed in the Section II the design of balun is started. To minimize the occupied area the branches of the coupled lines have been rolled up like an inductor. Moreover the lines that constitute each coupler have been placed on two different metal layers, more precisely on metal 7 for the input lines (green in the Figure 2) and metal 6 for the coupled lines (cyan). This solution guarantees the increase of the mutual magnetic coupling and the mutual capacitance is achieved by stacking the lines. Employing two identical couplers to build the structure, the simulations show a power imbalance of 1.5 dB and around 6 degrees of phase imbalance. This result is expected because the boundary condition for the two coupler are more different (see Figure 1). To reduce the imbalance and perform the input reflection loss, a session of optimization has been set up. The variables chosen for optimization are the physical dimension of the lines that constitute the device. Figure 2 shows the group of optimized variables. The black arrows represent the dimension of secondary coupled lines, the red ones instead are referred to the input lines. W is the line width, D is the vertical and horizontal diameter of the coils, P is the pitch between the lines, C is the coils distance. The variables A and G model the free ground area around the structure and the ground contact respectively. The ground layer has a strong influence for the balun response. There are two reasons: the first, concerns the confining of the magnetic field around the coils and the second one, concerns the role of ground contact for the output coupled coils. The sizing of the ground contact is a trade off between the parasitic coupling with the lines (that cause a drop off of impedance for the even mode) and the parasitics inductance generated on the metal contact. This inductance value, estimated around at 10-15 pH for a worst case, has a very strong effect on the balun in terms of power imbalance. Simulations show a maximum imbalance of 0.4 dB for a wrong design of the contact. To reduce the parasitics inductance, A 1 must be high. In this condition the inductance effect is attenuated and the imbalance removed. The only constraint to assure the same impedance on the balanced output is on the values of the width of the secondary coils:

W 1,2 = W 2,2 .
For the other parameters the only limit imposed is the respect of the design kit dimension rules. After the optimization run the dimensions of the two coils structures are very different. Figure 3 shows a 3D representation of the structure.

IV. SIMULATION RESULTS

The optimization run and the simulation were performed using Sonnet Release 12, version 12.52. This approach, for the design of miniaturized passive structures, has been widely employed and confirmed by measurements in a previous work [START_REF] Kraemer | Accurate electromagnetic simulation and measurement of millimeter-wave inductors in bulk CMOS technology[END_REF]. The balun response demonstrates the effectiveness of the design flow. Figure 4 shows the transmission power in terms of S 21 and S 31 and the reflection loss S 11 The power dissipation for the balun after the 3 dB subtraction due to the power split is 1.12 dB at 60 GHz. Regarding the peak of the reflection coefficient, it is possible to observe that it is not placed in the middle of the band of interest. This is due to the trade off among the minimization of power imbalance, (see equation ( 3)), and the input matching condition, (see equation [START_REF] Zhang | A new planar Marchand balun[END_REF]). However, the value of reflection coefficient is lower than 13 dB within the whole band of interest. On Figure 5 is shown the power and phase difference of the balanced output. The power imbalance for the structure is very limited with a maximum gap of 0.1 dB. Regarding the phase imbalance there is a topology constraint that imposes a limit to the minimum phase imbalance. The constraint is represented by the tapered The imbalance for the power and the phase increases for the higher frequencies but it remains limited to 0.4 dB and 8 degree at 90 GHz. At frequencies below 40 GHz the balun response drops down. This behaviour can be explained considering equations [START_REF] Lai | An Integrated 20-26 GHz CMOS Up-Conversion Mixer with Low Power Consumption[END_REF] and [START_REF] Felic | An Integrated Transformer Balun for 60 GHz Silicon RF IC Design, Signals, Systems and Electronics[END_REF]. The coupling to the balanced output depends on the electrical length of the line. For the frequency below 40 GHz θ is too short to guarantee a sufficient coupling. The power remains on the primary lines until the open circuit point, where the open line reflects the signal to unbalanced port (Port 1). The real dimension for the lines length employed to build the balun are about λ/12. This very low value for the length becomes too short for the low frequencies. The behaviour of the S 11 parameter for low frequency confirms this explanation.

V. CONCLUSION

This paper shows a new method to synthesize a balun for the RFIC design. The size, including of the ground free area is 127×79 µm, which is, to our knowledge, the smallest structure of this kind demonstrated so far. The balun simulations show a very flat response, for the whole operating band (57 -63 GHz) and the device can also be employed to higher frequencies without severe performance degradation. The balun decreases its performance for very high frequencies but the S11 remains however lower than -10 dB until 90 GHz. The low input return loss assures a good matching for the 50 Ohm source. The balun will be employed in a 60 GHz up-converter mixer. The fabrication run is scheduled for March 2010, using the CMP facilities (http://cmp.imag.fr/).

Fig. 2 .

 2 Fig. 2. Layout of Balun.

Fig. 3 .

 3 Fig. 3. Balun layout after the cycle of optimization.

Fig. 4 .

 4 Fig. 4. Output balun response and input return loss

Fig. 5 .

 5 Fig. 5. Power and phase imbalance between the differential output ports

Fig. 6 .

 6 Fig. 6. Wide band response of the balun

TABLE I STATE

 I OF THE ART FOR MILLIMETER-WAVE BALUNS (* SIMULATED VALUES )

		Bandwidth [GHz]	Insertion Loss	Input Return Loss	Amplitude	Phase	Occupied Area
			[dB]	[dB]	Imbalance [dB]	Imbalance[Deg]	[mm 2 ]
	This Work	50-70	1.1*	<14*	<0.1*	Max 5	0.01


Michael Kraemer
email: mkraemer@laas.fr

Mariano Ercoli
email: mercoli@laas.fr

Daniela Dragomirescu
email: daniela@laas.fr

Robert Plana
email: plana@laas.fr

A Wideband Single-balanced Down-mixer for the 60 GHz Band in 65 nm CMOS

This paper presents a single-balanced direct down conversion mixer for the unlicensed 60 GHz band. It is based on a differential pair employing the current bleeding technique. An integrated 60 GHz wideband passive balun allows the use of a single ended local oscillator (LO). The circuit is fabricated using the 65 nm bulk CMOS technology of ST Microelectronics. The mixer's baseband reaches from DC to 2 GHz. The measured radio frequency (RF) bandwidth exceeds 11 GHz, ranging from 54 GHz to at least 65 GHz. The measured RF, IF (intermediate frequency) and LO port return losses lie below -20 dB, -15 dB and -10 dB, respectively, within this entire band. A maximum conversion gain of 9.1 dB at 100 MHz from the carrier is achieved, while typical values measured at 1 GHz IF lie around 6 dB throughout the entire band. The mixer delivers these optimum results for LO powers as low as -5 dBm. The output referred 1 dB compression point reaches -5 dBm for -1 dBm of LO power. The simulated single sideband (SSB) noise figure is of 9 dB. Including the differential IF buffer that can drive two singleended 50 Ω loads, the circuit draws 16.8 mA from a 1 V supply. The pad-limited die size is only 0.52 x 0.49 mm 2 .

I. INTRODUCTION

A major research topic in the domain of microwave engineering today is the design of low-power low-cost silicon transceiver front-ends for the unlicensed frequency band around 60 GHz. While the first CMOS circuits for this purpose were designed in a 130 nm technology [START_REF] Emami | A 60-GHz down-converting CMOS singlegate mixer[END_REF], the use of lower scale CMOS technologies (like 65 nm for the circuit in this paper) with higher unity gain frequencies (f T , f max ) is one way to improve performance. However, the potential advantages are accompanied by drawbacks such as lower supply voltages (and thus decreased linearity), more stringent density rules and scaling of the metal back-end. This has to be taken into account during design. Furthermore, many early 60 GHz radio frequency integrated circuits (RFICs) employ transmission lines for matching purposes. This often results in large circuit sizes and thus increased cost. The better solution is the use of spiral inductors, as they do not only reduce circuit size, but also improve performance [START_REF] Kraemer | Accurate electromagnetic simulation and measurement of milimeter-wave inductors in bulk CMOS technology[END_REF], [START_REF] Dickson | 30-100-GHz inductors and transformers for millimeter-wave (Bi)CMOS integrated circuits[END_REF]. This paper presents the design and implementation of one of the basic building blocks of a 60 GHz direct conversion receiver: the down-mixer. Different mixer circuits downconverting signals from the 60 GHz band are described in literature, employing a multitude of topologies (cf. to [START_REF] Emami | A 60-GHz down-converting CMOS singlegate mixer[END_REF], [START_REF] Razavi | A 60-GHz CMOS receiver front-end[END_REF], [START_REF] Tsai | A 25-75 GHz broadband gilbertcell mixer using 90-nm CMOS technology[END_REF], [START_REF] Zhang | A 60-GHz double-balanced gilbert cell down-conversion mixer on 130-nm CMOS[END_REF], [START_REF] Kantanen | 60 GHz frequency conversion 90 nm CMOS circuits[END_REF], [START_REF] Kim | A single-balanced 60-GHz down-conversion mixer in 0.13 um CMOS technology for WPAN applications[END_REF], [START_REF] Sakian | A 60-GHz double-balanced homodyne down-converter in 65-nm CMOS process[END_REF], [START_REF] Ercoli | A high performance integrated balun for 60 GHz application in 65nm CMOS technology[END_REF]). Compared to these circuits, the solution discussed in this paper presents a better compromise for integrated low-cost low-power receivers operating in the 60 GHz band by achieving excellent performance while maintaining small size, low LO power requirements and a low power consumption (cf. section IV). The characterization of the circuit is facilitated by an integrated passive balun and a differential IF output buffer.

II. MIXER DESIGN

Fig. 1 illustrates the block diagram of the designed mixer. The RF input port is single-ended, because the preceding circuit block is going to be the single-ended low noise amplifier presented in [START_REF] Kraemer | A low-power high-gain LNA for the 60 GHz band in a 65 nm CMOS technology[END_REF]. The differential intermediate frequency (IF) output of the mixer is connected to a buffer amplifier based on a differential pair. The IF buffer is designed to drive a differential 100 Ω load (or alternatively two grounded 50 Ω terminations). The total circuit's power consumption is dominated by this buffer, which dissipates around 14 mW. In an integrated receiver circuit, this power-wasting buffer is not necessary because higher load impedances can be employed. The differential local oscillator (LO) port of the mixer core is connected to a passive on-chip balun. The balun is composed of two symmetrical spiral transformers connected as indicated in Fig. 1. Its layout can be identified in Fig. 3, where the balun is connected to the LO pad by an inductor that resonates

1 R 1 C 1 C 1 L D L B R B V DD V DD LO LO IF IF RF M1 M3 M2 V DD C B L 1 C 2 L 1 C 2 L 3 L 4 L 2
Fig. 2. Simplyfied schematic of the mixer core without biasing details the pad capacitance. The employed balun topology allows a much better amplitude and phase balance compared to a single transformer with center tap. For further details on the balun design refer to [START_REF] Ercoli | A high performance integrated balun for 60 GHz application in 65nm CMOS technology[END_REF]. Fig. 2 illustrates the schematic of the mixer core. A current bleeding [START_REF] Lee | Current-reuse bleeding mixer[END_REF] approach is employed to permit ideal biasing of both transconductance (M1) and switching (M2, M3) transistors. The current which by-passes the switching pair is determined by the resistor R B . It passes by the inductance L B , which at the same time resonates the parasitic capacitances present at the drain of M1 [START_REF] Razavi | A 60-GHz CMOS receiver front-end[END_REF]. The transistor M1 is biased at the minimum noise current density of 0.15 mA/μm. Its width is 14x1 μm. M1 is degenerated by the inductor L D . This improves linearity and allows a simultaneous noise and power match. Inductors L 2 , L 3 and L 4 realize this match (taking into account also the pad capacitance) over a wide bandwidth. Transistors M2 and M3, which are 35x1 μm wide, are biased at very low current densities to allow fast switching. The resistive loads R 1 are short-circuited by C 1 for RF and LO frequencies to increase conversion gain G C and improve isolation. The inductor L 1 is used to match the gates of the switching pair to the balun's impedance. The capacitor C 2 is inserted to cut the DC path and permit a bias voltage at the gates of M1 and M2.

A. Transistors

The transistors used in the mixer circuit are general purpose (GP) devices. GP devices exhibit higher f max as low power (LP) devices at a given bias current, thus mandating their use for low power RFIC design. A consequence of the use of GP transistors is the lower breakdown voltage of 1 V, which limits the circuit's supply voltage. While the intrinsic performance of the transistors is well represented by the BSIM4 model, it does not take into account

B. Passive Devices

As inductive matching elements, spiral inductors are used rather than transmission lines. Their performance with respect to size, realizable values and quality factor are superior even when using CMOS technologies with low resistivity substrates [START_REF] Dickson | 30-100-GHz inductors and transformers for millimeter-wave (Bi)CMOS integrated circuits[END_REF]. The design of the spiral inductors is lined out in [START_REF] Kraemer | Accurate electromagnetic simulation and measurement of milimeter-wave inductors in bulk CMOS technology[END_REF]. The focus lies on minimizing the parasitics rather than maximizing the quality factor. Electromagnetic simulations using Sonnet are employed to do quantitative optimizations [START_REF] Kraemer | Accurate electromagnetic simulation and measurement of milimeter-wave inductors in bulk CMOS technology[END_REF]. The Sparameters obtained by simulation of the final geometry are used to parametrize a 2-π model, which is employed in the circuit simulations to precisely predict the inductor's behavior. A similar approach is employed for the balun [START_REF] Ercoli | A high performance integrated balun for 60 GHz application in 65nm CMOS technology[END_REF].

C. Fabricated Circuit

The presented circuit was fabricated in the 65 nm CMOS technology of STMicroelectronics with two thick copper metal layers. Fig. 3 shows a photo of the fabricated circuit. A very small, pad limited die size of only 0.49mm x 0.52mm = 0.255mm 2 is obtained, which is further reduced when integrating the mixer into the receiver.

III. RESULTS

The mixer circuit of Fig. 3 is measured on-wafer up to 65 GHz. The return loss is determined using an Anritsu ME7808A vector network analyzer (VNA). The LO signal is generated by an Agilent E8257D source that provides up to 14 dBm output power at 60 GHz. Power conversion gain and power sweep measurements are done using the VNA as RF signal source and connecting While measuring, the mixer is biased at the current densities determined during design. The bias of the switching pair depends on the LO power and is set to 0.7 V for the LO power sweep. For the other measurements, P LO equals -1 dBm and the switching pair is biased at 0.56 V. The circuit including buffers is drawing 16.8 mA from a 1 V supply, from which only 2.8 mA are attributed to the mixer core. Fig. 4 illustrates the measured return loss (RL) at the RF and LO ports. An excellent broadband match is assured, contributing to a very flat, wideband response of the mixer. The measured IF RL stays below -15 dB up to 5 GHz. The LO power is swept for f LO = 60 GHz and f RF = 61 GHz to find the value that achieves maximum conversion gain G C . According to Fig. 5, this is the case at the low LO power value of -5 dBm. Fig. 6 plots the conversion gain G C in the lower (LSB) and upper sideband (USB). A very flat, wideband response can be observed for IF frequencies of 1 GHz and 2 GHz, while both sidebands are very symmetric. The conversion gain is around 6 dB at 1 GHz from the carrier over the whole band. The measurement frequency range was bounded to below 65 GHZ due to the used equipment. Fig. 7 shows the conversion gain for five fixed LO frequencies Isolation is less important for this mixer, due to the distance from baseband to LO/RF which allows inexpensive low pass filtering on-chip. LO to IF isolation is observed to be around 30 dB, while RF to IF isolation is around 28 dB. The former value is expected to increase if an integrated differential oscillator with even better phase balance provides the LO signal. The simulation of the Noise Figure (NF) using SPECTRE RF shows a double sideband (DSB) value of around 9 dB, which gives a theoretical SSB value that lies at 12 dB. Measurement results are expected to be very close due to the use of BSIM4 models and the fact that all parasitics are taken into account during simulation.

G C in dB f RF in GHz f LO =56GHz f LO =58GHz f LO =60GHz f LO =62GHz f LO =64 GHz

IV. COMPARISON TO THE STATE OF THE ART

Table I compares the presented work to 60 GHz CMOS mixers found in literature. The proposed mixer has the smallest chip size, one of the highest gain values and a very wide IF and RF bandwidth that makes it one of the best candidates for an integrated receiver covering the worldwide available 60 GHz band. The fact that its DC power consumption is relatively high originates from the IF buffers that drive two 50 Ω loads. Yet, the proposed mixer requires the lowest LO power.

V. CONCLUSION

The single balanced down-mixer presented in this paper is the perfect choice for low-cost low-power direct conversion receivers due to its conversion gain of up to 9.2 dB, its low LO power requirement of -5 dBm, its moderate DSB noise figure of 12 dB and its high OP -1dB of -5 dBm. Its ultra-wide bandwidth respects the channel requirement of all standards that exist for the unlicensed 60 GHz band. The very small (pad-limited) size of only 0.49x0.52 mm 2 allows low cost integration.

Fig. 1 .

 1 Fig. 1. Block diagram of the downconversion mixer with integrated balun and IF buffer

R

Fig. 3 .

 3 Fig. 3. Die photo of the fabricated down-mixer

Fig. 4 .PFig. 5 .

 45 Fig. 4. Measured return loss at LO and RF port

Fig. 6 .

 6 Fig. 6. Conversion gain G C versus LO frequency for two different intermediate frequencies, both in upper (USB) and lower sideband (LSB).

Fig. 7 .Fig. 8 .

 78 Fig. 7. Conversion gain G C versus f RF for different LO frequencies. The IF is varying from 100 MHz to 5 GHz.0 6

TABLE I COMPARISON

 I OF PUBLISHED 60 GHZ DOWN-CONVERSION MIXERS IN CMOS

	10
	8
	6
	4
	2
	0
	-2
	52 -4	54	56	58	60	62	64
	Reference			Topology		Technology (nm)	IF (GHz)	RF (GHz)	Gc (dB)	NF (SSB) (dB)	P LO (dBm)	OP -1dB (dBm)	P diss (mW)	Die Size (mm 2 )
	Emami et al., 2005 [1]		single gate		130	2	55-61	-2	11.5	0	-5.5	2.4	1.6x1.7
	Razavi, 2006 [4]		half Gilbert cell	130	N.A.	60	12	18*	N.A.	N.A.	0.9	N.A.
	Tsai et al., 2007 [5]		Gilbert cell		90	0.01	25-75	3 ± 2	N.A.	6	-4	93	0.55x0.55
	Zhang et al., 2007 [6]		Gilbert cell		130	1	55-63	3	N.A.	0	-12	N.A.	0.9x0.9
	Kantanen et al, 2008 [7]		resistive		90	0.4	60	9.8	N.A.	5	-2	14	1.1x0.7
	Kim et al., 2009 [8]		half Gilbert cell	130	5	60	4.9	N.A.	N.A.	-15	8.2	N.A.
	Sakian et al., 2009 [9]		Gilbert cell		65	0 -1.3	60	4	12	N.A.	-16**	6	N.A.
	Ercoli et al., 2010 [13]		passive		65	N.A.	50-70*	-6.9*	11.4*	-5*	N.A.	15*	N.A.
	this work		half Gilbert cell	65	0-2	54 -65	9.1	12*	-5	-5	16.8	0.49x0.52
				*simulated; **obtained from OIP3-10 dB # voltage conversion gain;

ACKNOWLEDGEMENT

Thanks go to Prof. Sorin Voinigescu of University of Toronto and his students for valuable hints on mm-wave circuit design and STMicroelectronics for technology access.