Mariano Ercoli
email: mercoli@laas.fr

Michael Kraemer
email: mkraemer@laas.fr

Daniela Dragomirescu
email: daniela@laas.fr

Robert Plana
email: plana@laas.fr

An High Performance Integrated Balun for 60 GHz Application in 65nm CMOS Technology

Keywords: Balun, differential balanced transformer, millimetre-wave devices

This paper shows a new design approach that allows to integrate an efficient balun into mm-wave radio frequency integrated circuits (RFIC). The proposed device is an evolution of the simple transformer. Thanks to the modification on the device's shape, the performance increases considerably, maintaining limited dimensions. The proposed balun shows a very good power division with only 0.5 dB of maximum amplitude imbalance within the whole band of interest and less than 10°phase imbalance around 60 GHz. The measurement campaign confirms the accuracy of the electromagnetic simulations and the behaviour of the device. The balun is implemented in ST Microelectronic's 65nm bulk CMOS technology and is employed in a double balanced 60 GHz down-conversion mixer to provide the balanced the LO and RF signal.

I. INTRODUCTION

In mm-wave circuit design, the use of balanced architectures in the transceiver components is strongly beneficial. The benefits, like improvement of noise immunity [START_REF] Dickson | 30-100-GHz inductors and transformers for millimeterwave (Bi)CMOS integrated circuits[END_REF], resulting from these architectures, often justify the increase of the number of active components and the higher power consumption. The most common applications of differential architectures are mixers, voltage-controlled oscillators (VCO) and single-stage or multi-stage input and output amplifiers. Often input and output signals are only available in single ended form, so a structure that converts them into differential signal is needed. The behaviour of the single-ended-to-balanced transformation becomes crucial when determining the performance of the entire transceiver. There are several examples for the importance of the conversion: 1) In a mixer: an unbalanced power division of the LO signal causes asymmetry of the control signal that drives the switching block. The conversion gain of the mixer drops and, increasing non-linearity, degrades the performance. Even a phase imbalance influences the behaviour of the mixer: the phase error leads to degradation of the signal isolation of the structure. 2) In a single or multi stage amplifier: a non-optimal power splitting degrades the performance of a differential amplifier. An excessive imbalance can generate a saturation of one of the two sides while the other one is still far from the compression. The phase imbalance decreases the combined signal's power at the output. To avoid these issues, different architectural solutions exist when designing a balun. The active balun solution is good, thanks to its very limited occupied area, but in the 60GHz band the problem of common mode rejection is very difficult to solve. A passive structure instead, is not effected from this problem.

In literature tree different classes of passive balun are described: the Marchand Balun, the Rat-race Hybrid and transformers. The size of the classical Marchand Balun and the Rat-race configuration are proportional to the wave length of the signal. In 60 GHz RFIC's the chip surface dedicated to build these types of device is excessive. Moreover, wide passive circuits in silicon bulk technology show high losses and consequently thermal noise. To reduce the occupied area while maintaining good performance for the balun, the transformer architecture is often employed [START_REF] Dickson | 30-100-GHz inductors and transformers for millimeterwave (Bi)CMOS integrated circuits[END_REF]. This solution can perform two different actions, imbalanced to balanced signals conversion and impedance matching between different blocks. [START_REF] Dickson | 30-100-GHz inductors and transformers for millimeterwave (Bi)CMOS integrated circuits[END_REF] that it is possible to build transformer structures in the 60 GHz frequency band that achieve a very high coupling factor for extremely limited dimensions. This is an important property that can be further improved by the use of a stacked coil configuration. The superposition of the primary and secondary coils into different metal layers reduces the covered silicon surface of the transformer and increases the density and the regularity of the magnetic field in the centre of the component. For a planar transformer, with primary and secondary coils in the same layer, [START_REF] Dickson | 30-100-GHz inductors and transformers for millimeterwave (Bi)CMOS integrated circuits[END_REF] shows a coupling factor k of around 0.5. Instead, for stacked transformers, (k) can be close to 0.75 and even higher. In [START_REF] Dickson | 30-100-GHz inductors and transformers for millimeterwave (Bi)CMOS integrated circuits[END_REF] the occupied area for the structure, including the free space around the coils, is a square with a lateral dimension of less than 60µm, while the external coil diameter is only 30 µm.

II. DESIGN

A. Transformer Features

Dickson et al. show in

B. Balun Structure Design

The small size of the component and its high value of k make of the integrated transformer [START_REF] Dickson | 30-100-GHz inductors and transformers for millimeterwave (Bi)CMOS integrated circuits[END_REF] an ideal instrument to develop a balun with reduced dimensions and high performance. Transformer showed in Figure 1 (a) has the secondary coil connected directly to a component with differential input (ports 2 and 3). This configuration generates a virtual ground in correspondence of the middle of the output coil (b). The virtual ground guaranties an equal power splitting and a 180°of phase shift for each output signal. Simulations show that the real behaviour of the transformer balun is very far from the ideal one. This is due to the presence of a parasitic phenomena that modify the ground constraint. The main problem, when using transformer as a balun, is the control of these effects. The amplitude and phase balance depends on the proper behaviour of the ground at the secondary coils. Fig. 2(a) shows a first example of modified transformer employed as a balun: while the single ended signal is applied to the primary coil (1), the secondary coil is divided (2,3) by a ground connection in the middle. The contact, placed on metal 3, forces the ground condition by a real ground contact. The response of the structure improves but it still does not assure equal power distribution and phase balance. Another solution that helps to reduce the imbalance is the use of a planar balun instead of a stacked. In this case, the imbalance is lower, but also the power passing from primary to secondary coils. This is due to the decrease of k, involving a worsening of the performance of the entire block.

Theoretical analysis shows that the imbalance at the output of the balun is principally due to the strong capacitive couplings in the structure. This is confirmed by the behaviour of the planar balun; in the latter, the absence of stacked metal reduces the capacitive coupling effect. In the stacked version, the coupling is much stronger and, in fact, the increase of k can be justified by the increase of mutual magnetic field between the coils and by the capacitive effects between the superposed structures. The use of a stacked transformer is discouraged by these parasitic interferences but the good performance presented by the stacked structure can be exploited in a different configuration: to build a balun that employs the transformer shape, it is necessary to create a new type of pattern.

The balun architecture described in this paper is inspired to the Marchand balun proposed in [START_REF] Chao-Shiun | A 0.13m CMOS fully differential receiver with on-chip baluns for 60GHz broadband wireless communications[END_REF]. The fundamental difference between the structures is the termination of the primary coils. In [START_REF] Chao-Shiun | A 0.13m CMOS fully differential receiver with on-chip baluns for 60GHz broadband wireless communications[END_REF] the Marchand architecture imposes an open circuit instead, for the transformer the primary coil is shorted to ground. The technique adopted to create the balun is based on the splitting of the transformer and the creation of two different small devices wherein the primary coil is a single line with a particular shape that joins the two structures, see Figure 1(b). Thanks to this configuration the effects that are considered parasitic in a classical transformer enhance the coupling in the structure shown in Figure 1(b). A little imbalance between the differential output persists because different boundary conditions exist between the two different coils. Moreover, the propagating modes inside the structure influence the coupling of electromagnetic fields in the two coils. The contribution of even and odd modes are different for each output [START_REF] Ercoli | An ultra small passive balun for 60 GHz applications in CMOS 65nm technology[END_REF]. To compensate the imbalance, the dimensions of the coils are modified. The optimization is focused in particular on the coil diameter and the width of the lines. Simulations show that a different diameter for the coils and a different width of the lines can affect the response of each single transformer in the balun's architecture. The phase difference (w.r.t 180°) and amplitude imbalance after a cycle of optimization are strongly reduced. At port 1, a return loss coefficient of around -8 dB is achieved within the whole band of interest. To improve this S 11 value a little series inductor can be employed.

III. SIMULATIONS AND MEASUREMENTS A. Simulation Environment and Optimization

The optimization of the balun structure is focused on the transmission parameters: S 12 and S 13 are analysed to minimize the amplitude and phase differences. Figure 3 shows the response in terms of transmission parameter of the two structures described and shown in Figure 2. The differences, between configuration (a) and (b), are obvious: in the classical shape transformer balun, the amplitude difference between the output increase with the frequency. The parasitics effect on the stacked metals became strong and an high part of the signal is directly transferred to the port 2. The S 12 coefficient simulated for the classical configuration agrees with the measurement showed in [START_REF] Dickson | 30-100-GHz inductors and transformers for millimeterwave (Bi)CMOS integrated circuits[END_REF](S 12 parameter in Figure 10). For the phase imbalance (not shown) the difference on the behaviour between the two architectures is even worse.

In the developed structure (1b) the change of the shape and the optimization on the coil result in a very similar response for both transmission coefficients. Figure 3 shows the simulated response. Figures 5 and6 show the amplitude and phase difference between the differential output. It is evident that the amplitude imbalance is lower than 0.5dB and the phase difference is under 10°within the whole band of interest. This response confirms the efficacy of the optimization procedure and assures a very large operative bandwidth for the component. The simulations are done using Sonnet Software, release 12.52. The method to assure a high precision of simulation was already investigated and confirmed by previous studies [START_REF] Kraemer | Accurate Electromagnetic simulation and measurement of millimeter-wave inductors in bulk CMOS technology[END_REF].

B. Measurements Setup and Results

The measurements are done using the balun test structure shown in Figure 4. The balun is connected by three lines with 102µm length to a coplanar ground-signal-ground (G-S-G) single ended probe for the input and a differential G-S-G-S-G coplanar probe for the output. Not all ground pads are shown to focus the attention on the structure. In order to obtain the real value of the balun parameters, (without pads and connections) firstly a short-open-load-thru (SOLT) calibration was done using an impedance standard substrate (ISS). With this operation, an error correction is applied on the measured equipment. The second step was a de-embedding procedure to remove the influence of the connection lines and pads [START_REF] Kraemer | Accurate Electromagnetic simulation and measurement of millimeter-wave inductors in bulk CMOS technology[END_REF][START_REF] Kraemer | On the De-embedding of small value millimeter -wave CMOS inductor measurement[END_REF]. The de-embedded results are showed in Figures 5,6,7. Figure 5 shows the comparison between simulation and measured results. The simulations well agrees with the measurements up to 35GHz. Above this frequency the behaviour degrades. This is due to the absence of the influence of the dummy metals during the simulation phase of the balun. The compliance with minimum metal density rules is mandatory for the production of the device. Thus the dummies must to be inserted. The effect of dummies in the RF lines is demonstrated in [START_REF] Seo Munkyo | A 150 GHz Amplifier With 8 dB Gain and + 6 dBm Psat in Digital 65 nm CMOS Using Dummy-Prefilled Microstrip Lines[END_REF]. The difference in the electromagnetic fields, due to the presence of dummies, modifies the response and increases the losses of the balun. Regarding the relative difference between the differential outputs, the simulations confirm the measurements. In particular, concerning amplitude imbalance, the mean value of the measured curve matches with the simulated curve. Regarding the phase, an increase of the difference is shown. The value of the phase shift, however, remains limited to below under the 10°. To confirm the supposed effect of the dummies presence a simulation was done. It is necessary to underline that a complete simulation of the balun with the dummies is not possible. This type of analysis in fact requires an extremely high memory resources. The cause of this condition is the combination between the simulation set-up described in [START_REF] Kraemer | On the De-embedding of small value millimeter -wave CMOS inductor measurement[END_REF] and the number of the dummies. To extract an estimation of the dummies influence a trade off can be done. The first step is the evaluation of the structures behaviour using the process simulation proposed in [START_REF] Kraemer | On the De-embedding of small value millimeter -wave CMOS inductor measurement[END_REF] and after, a simplified simulation is made to evaluate the variation of the electromagnetic field in the structure with dummies. The principal problem in fact is the number of mesh levels used to simulate a single metal layer of the technology. To obtain an high precision at least one metal layer for each 100nm of metal height is needed. This, together at the increase of the metal bricks, due to the presence of dummies, causes an explosion on the number of mesh cells and consequently on the memory needed. In these condition the simulation becomes impossible. However, with this simplified approach it is possible to ascertain a little variation on the component's behaviour.

Fig. 1 .

 1 Fig. 1. Ideal behaviour for a transformer balun

Fig. 2 .

 2 Fig. 2. Balun configurations simulated; classical transformer balun (a) and modified balun (b): Violet Alucap layer, Green metal 7 layer, Cyan(coils) metal 6, Pink metal 5, Cyan(external) metal 1

Fig. 3 .

 3 Fig. 3. Tranmission coefficient in terms of S12 and S13 for the two explored solution.

Fig. 4 .

 4 Fig. 4. Balun test structures used to confirm the simulations)

Fig. 5 .

 5 Fig. 5. Transmission parameter for the balun

converts the RF and the LO signal in the differential inputs of the mixer. Figure 9 shows the down-converted response of the mixer. The measure is performed by a LeCroy SDA 813ZI oscilloscope with an averaging of 1000 samples to minimize the noise floor. The extracted phase difference between the two channel is 10 and the amplitude difference is only 0.21 dB. Figure 10 shows the behaviour of the output signal with a persistence of 20 seconds to demonstrate the stability of the measurements.

V. CONCLUSION This paper shows the design and the performance of a balun for 60GHz applications. The measurement confirms the excellent behaviour predicted by simulations. The balun shows very limited loss better than the state of the art for 60GHz applications [START_REF] Ercoli | An ultra small passive balun for 60 GHz applications in CMOS 65nm technology[END_REF] (Table 1). Concerning balun dimensions it is the smallest device found in literature with only 0.011 mm 2 , 80×139µm comprehensive of free space around the coils. The good performance of the balun in an active device have been demonstrated by the use in a fully balanced down conversion mixer.


Keywords: RF-MEMS, Capacitive, Shunt Switches, Phase Shifters, Modeling

This paper presents an approach to RF MEMS capacitive shunt switch design from K-band up to W-band, based on the scalability of the RF MEMS switch with frequency. The parameters of the switch's equivalent circuit model also follow scaling rules. The measurement results of the fabricated switches show an excellent agreement with simulations which allow to validate the MEMS model in the entire band from 20 GHz up to 94 GHz. This model is going to be used in phase shifter circuit design for antenna array applications. First 60-GHz phase shifter results are also reported here.

I. INTRODUCTION

ADIO Frequency Micro-Electro-Mechanical Systems (RF MEMS) are a new kind of passive circuit components, used in broad-band communication systems. They have multiple applications, such as varactors, switches and resonators.

RF MEMS micro-switches have demonstrated good performances in terms of isolation, loss, and power handling far better than those of conventional diodes. So, the purpose is to replace the currently used active circuits in systems by passive circuits (based on RF MEMS), and one of the challenges is to obtain RF circuits operating at very high frequencies (up to 94 GHz for radar systems).

Table I [START_REF] Hung | Distributed 2-and 3-bit Wband MEMS phase shifters on glass substrates[END_REF][START_REF] Hong-Teuk | V-band 2-b and 4-b low-loss and lowvoltage distributed MEMS digital phase shifter using metal-air-metal capacitors[END_REF][START_REF] Byung-Wook | Ka-Band BiCMOS 4-Bit Phase Shifter with Integrated LNA for Phased Array T/R Modules[END_REF][START_REF] Kwang-Jin | 0.13-µm CMOS Phase Shifters for X-, Ku-, and K-Band Phased Arrays[END_REF][START_REF] Yu | A 60GHz digitally controlled phase shifter in CMOS[END_REF] shows the state-of-the-art of integrated phase shifters. Note that RF MEMS phase shifters exhibit low insertion loss with very low phase error. The challenge is to design 60-GHz phase shifters on a silicon circuit compatible substrate for future antenna array systems. Therefore, at first, a shunt capacitive switch model scalable up to millimeterwave frequencies has been developed from previous 20-GHz MEMS measurements.

Manuscript received January 31, 2009. This work was supported by the French National Research Agency (Radio Soc Project, JC05-60832) and the Aeronautic and Space Research Foundation (LIMA Project).

Authors are with CNRS ; LAAS ; 7 avenue du colonel Roche, F-31077 Toulouse, France and Université de Toulouse ; UPS, INSA, INP, ISAE ; LAAS ; F-31077 Toulouse, France (corresponding author to provide phone: +33561336816; fax: +33561336969; e-mail: vpuyal@ laas.fr). Then this model has been extrapolated to build different RF MEMS models up to 94 GHz. The electrical model of RF MEMS is based on a RLC shunt circuit [START_REF] Qian | A parametric model of low-loss RF MEMS capacitive switches[END_REF][START_REF] Muldavin | High-isolation CPW MEMS shunt switches. 1. Modeling[END_REF][START_REF] Bartolucci | Circuital Modelling of Shunt Capacitive RF MEMS Switches[END_REF]. In Ref. [START_REF] Qian | A parametric model of low-loss RF MEMS capacitive switches[END_REF], a parametric capacitive switch model based on the bridge width is shown. Compared to this last model, the new aspect is to introduce frequency scalability on the switch model to improve and simplify RF MEMS integrated circuit design at millimeter wave frequencies. Other approaches have been also published including the dynamical [START_REF] Halder | Compact RF Model for Transient Characteristics of MEMS Capacitive Switches[END_REF] or mechanical [START_REF] Marcelli | Mechanical modelling of capacitive RF MEMS shunt switches[END_REF] aspects of RF MEMS.

Based on different graphs plotted in this paper, straightforward creation of an RF MEMS model is possible. Finally, to illustrate the results obtained when employing the electrical switch model, it is used to design 60-GHz phase shifters. Preliminary results of this study were presented in [START_REF] Puyal | RF MEMS Design for Phase Shifters from K-band up to W-band[END_REF].

II. TECHNOLOGY FOR MILLIMETER WAVE MEMS

A. Fabrication Methodology

The fabrication process flow of RF-MEMS capacitance microswitches is depicted in figure 1

B. Description of the switch

The switch structure (Fig. 2) is a RF MEMS capacitive shunt switch [START_REF] Bordas | High quality medium power RF-MEMS based impedance tuner for smart microsystem integration[END_REF]. It consists of a 50 Ω Grounded CoPlanar Waveguide (GCPW) over which a metal membrane is suspended at a height of 3µm. The membrane consists of two main elements:

-the bridge having two states, corresponding to two capacitance values (up state or down state),

-two actuation electrodes located on each side of the bridge. A layer of silicon nitride of 0.25 µm thickness is placed between the bridge and the signal line of the coplanar waveguide line. The thickness of the dielectric partially defines the down state capacitance value. Actuation is provided by a voltage (around 30 V) between the bridge and the different under-layer structures (electrodes and signal line). The bridge capacitance depends on the bridge area (w x W µm²) as shown in the switch microphotograph in figure 2.

III. RF MEMS MODELING

A. Electrical Part

To design RF MEMS phase shifter circuits, an accurate wideband RF MEMS model is needed. The model (Fig. 3) is based on a classic RLC equivalent circuit [START_REF] Rebeiz | RF MEMS: Theory, Design, and Technology[END_REF] with two short series sections of transmission line (t-line). To simplify the model, a scaling parameter K s is defined according to [START_REF] Puyal | A New Approach to RF MEMS Shunt Switch modeling from K-band up to W-band[END_REF]. To build this model, previous simulation and measurement results of 20-GHz fabricated RF-MEMS shunt switches are used. The bridge capacitance is 80 fF in up state, and 2300 fF in down state for a 20-GHz MEMS (K s =1). The series resistance is equal to 0.1 Ω. The S parameters of the reference switch show that it has a very low insertion loss of 0.27 dB (including access lines) and excellent isolation of 48 dB (Fig. 4) at 20 GHz. These positive results validate the technology and show their potential concerning integration into circuits. The first model at 20 GHz has been extrapolated up to 94 GHz. The scaling parameter K s is defined as: .

1 1 , x x x s W W w w K   (1a)
The letter x denotes the considered switch (x = 1 for M1 switch, 2 for M2 switch, 3 for M3 switch, etc.)

Note that in all sections where x is not explicity given, it remains unchanged (K s = K s,x ; w= w x ; W= W x ; etc.).

When K s =1, the MEMS dimensions correspond to the (previously measured) reference switch (20-GHz switch). Hence, the reference bridge dimensions are 200 µm × 160 µm (w 1 × W 1 ).

The scaling parameter is applied to all dimensions (Fig. 2) of the switch structure except for the vertical one: the metal thickness and the bridge height are constant. Consequently, beam dimensions (1b) and actuation electrode dimensions (1c) are also reduced according to

    , ( 1 , , 1 , , 1 , ,
) x eb eb x ea ea x s l l l l K , 1 , , 1 , ,   . (1c) 1b

Capacitance

The capacitance is represented by the bridge area. The down-state capacitance is

d r down t A C      0 (2)
where ε 0 is the electric constant, A the area of the bridge, t d the dielectric thickness (0.3 μm) and ε r the relative permittivity (6.5).

The up-state capacitance [START_REF] Puyal | A New Approach to RF MEMS Shunt Switch modeling from K-band up to W-band[END_REF] is

r d up t g A C      0 ( 3
)
where g is the air gap (3 μm) between the dielectric and the bridge.

The capacitances of the device scale according to

2 1 s K C C  (4)
with C 1 the capacitance of the 20-GHz MEMS.

Inductance

The series inductance is mainly formed by the beams of the RF MEMS switch. The equation for the inductance [START_REF] Wadell | Modeling Circuit Parasitics: Part 2[END_REF] of a rectangular conductor is

                                c c c c c c c l t w t w l l pH L 224 . 0 5 . 0 2 ln 10 2 ) ( 1 (5)
where l c is the length (in µm), w c the width and t c the thickness of the conductor.

After some calculations, we obtain an effective parameter K' that equals to L/L 1 for each inductance. This factor depends on the inductance size. Reference [START_REF] Puyal | A New Approach to RF MEMS Shunt Switch modeling from K-band up to W-band[END_REF] shows the different K' parameters obtained versus K s . We can conclude that

s K L L 1  with 4  s K (6)
where L 1 is the inductance of the 20-GHz MEMS.

Resistance

The series resistance is principally due to the conductor loss inside the beams.

The series loss R of a conductor is calculated from the material's resistivity ρ :

.

c c c t w l R   (7)
where l c is the length, w c the width and t c the thickness of the conductor.

It's straightforward to calculate the DC value but, at high frequencies, the effective area w c t c_eff decreases because the

C = C 1 /K s 2 L = L 1 /K s R TLIN TLIN 1 2
skin effect becomes relevant. So, when frequency increases (decrease of K s ), the area w c t c and the length l c decrease. That's why, it is difficult (for a first, simple model) to determine specifically the part of each effect. Indeed, Si 3 N 4 dielectric loss is added and the model becomes quite complex.

Transmission Line

The two t-line sections are of length l TLIN given by: . 2

1 s TLIN K w l  (8)
Their characteristic impedances are equal to 50 Ω. Other parameters (as dielectric loss and conductor loss) are included within the model of the microwave circuit simulator (ADS) and depend on frequency.

The resonance frequency f 0 [START_REF] Rebeiz | RF MEMS: Theory, Design, and Technology[END_REF] (maximum isolation) is inversely proportional [START_REF] Halder | Compact RF Model for Transient Characteristics of MEMS Capacitive Switches[END_REF] to the square root of the LC factor

LC f  2 1 0  . ( 9
)
When dimensions decrease, the series inductance and the down-state capacitance drop; so, the resonance frequency increases.

With ( 4) and ( 6), equation ( 9) becomes

2 3 01 0 s K f f  , (10)
where

1 1 01 2 1 C L f   . (11)
So, the scaling parameter is given by

3 2 01 0          f f K s . ( 12
)
For example, if a 60-GHz MEMS is to be designed, f 0 = 60 GHz, f 01 = 20 GHz, so K s = 2.08 from [START_REF] Bordas | High quality medium power RF-MEMS based impedance tuner for smart microsystem integration[END_REF]. Thus, all dimensions of the reference switch have to be divided by 2.08. Table II summarizes the different simulated values using this model. The first model has been extrapolated up to 94 GHz. This model shows that using a scaling parameter (K s ) simplifies the search for the optimum frequency response when the resonance frequency is specified.

B. Mechanical Part

The reference switch (20-GHz MEMS) has a low residual stress of 5-7 MPa and a spring constant of 22-27 N/m. The air gap is around 2.5 µm (and thus differs from the predicted 3 µm) for the first fabricated MEMS. Now, it is interesting to investigate the actuation voltage versus the scaling parameter K s . The "pull-in" (or "pulldown") voltage is defined as the applied voltage for collapse the bridge on the signal line.

The "pull-in" voltage [START_REF] Rebeiz | RF MEMS: Theory, Design, and Technology[END_REF], for a central single actuation, when the effect of the dielectric layer is neglected, is given by

3 0 0 , 27 8 g w W k V x x x x in pull    (13)
where k x is the effective spring constant of the RF MEMS and g 0 the bridge height in up-state. From (13):

1 1 1 2 1 , , k w W w W k V V x x x in pull x in pull             , ( 14
) so x s in pull x x in pull K V k k V , 1 , 1 ,      (15)
In the equation ( 15), the "pull-in" voltage V pull-in increases according the scaling parameter K s and the square root of the ratio of effective spring constants k x /k 1 .

IV. MODEL RESULTS

A. Capacitance Measurements

The experimental results of the first measurement step of the RF MEMS switches described previously are presented here. In figure 5, the results of capacitance measurement of a 60-GHz MEMS (K s = 2.08) are shown (C up = 20 fF, C down = 550 fF, C on /C off = 28).

At the beginning, the bias voltage is 0 V: the switch is in up-state (small capacitance close to 0-pF line). Then, the bias voltage is increased linearly up to 44V. Around 15 V, the central bridge begins to actuate and the capacitance changes to 0.25 pF. When the voltage equals 30 V, the lateral electrodes finish to descend and the actuation is complete (down-state).

Here, the capacitance is 0.55 pF. Finally, the bias voltage is decreased linearly down to -44V and finally increased again to 0V. A hysteresis appears on the graph; Thus, the voltages necessary to change from down-state to up-state and viceversa are different (V pull-in = 30 V, V pull-out = 16V). For all switches, a comparison between the theoretical and the experimental down-state capacitance (Tab. III) is done. For each kind of RF MEMS, an average of about 20 measurements is shown.

Some very small differences can appear between the capacitance values in the

B. S-parameter Measurements

The RF performance of each MEMS from M1 up to M5 are presented in this section.

The RF characterization of the switches requires adequate measurement equipment as well as special care in the measurement setup. The switch characteristics were measured using on-wafer probing. The measurement setup is composed of:

-a 65-GHz Anritsu 37397C VNA (Vector Network Analyser), -a personal computer that allows to display S-parameter waveforms and download S-parameter files (for electrical simulations), -65-GHz probes and 1m V-type coaxial cables. Above 65 GHz, 110-GHz probes, an Anritsu mixer and source modules have to be used to measure from 65 GHz to 110 GHz (for switches M4 and M5).

A 100-V bias tuning voltage module allows to easily adjust the actuation voltage between 0 and 45V.

For all S-parameter presented measurements, the CPW access lines are de-embedded.

The microphotographs of all switches from M1 (20 GHz-MEMS) up to M5 (94 GHz-MEMS) are presented in figure 6. Insertion loss and isolation of all switches versus frequency are shown in figure 7.

The Tab. IV resumes all characteristics of the measured RF MEMS. The dimensions of the switches are comprised between 800 µm × 1000 µm (for the largest one) and 350 µm × 340 µm (for the smallest one). Insertion loss and isolation of the switches are defined at the modeled resonance frequency (For example, for M1, at 20 GHz and not at 19.5 GHz). The capacitive shunt switches present good insertion loss (average value: 0.6 dB) and excellent isolation (average value: 30 dB). The operating band (specified by a 20-dB isolation level) of the switch is about 15 GHz. The resonance frequency is close to the predicted one for each switch (± 1 GHz). The actuation voltage lies between 20 V and 35 V.

Note that the many holes present on the top metal layer (bridge and electrodes) is used to facilitate the removing of the sacrificial layer during the switch fabrication. For the MEMS M1, their impact on the effective bridge area is negligible, whereas for smaller MEMS, they decrease it slightly.

C. Scalability

In this section, the dependency of most of the switch parameters on K s is described and compared to the theoretical model. All model parameters have been extracted from previously fitted graphs (C down , L, R from the isolation graph and C up from the insertion loss graph).

Figure 8 shows the up-state capacitance C up versus the scaling parameter K s . Good agreement with the model equation ( 4) is observable. In this case, from the fitted equation (y = 151x -2 ), the parameter C 1 is found to be 151 fF (Theory: 80 fF). The difference between the experimental value and the theory value of the capacitance can be explained by the difference of sacrificial layer thickness (i.e. bridge height) used in simulation (3 µm) and obtained after fabrication (2-2.5 µm). Figure 8 also shows the down-state capacitance C down versus the scaling parameter K s . An excellent agreement with the model equation ( 4) is observable as well. In this case, from the fitted equation (y = 2239x -2 ), the parameter C 1 is equal to 2239 fF (Theory: 2300 fF).

So, the capacitances of the switch in up-state and in down-state as a function of K s are:

2 151 ) ( s up K fF C  , ( 16
)
and 2 2239 ) ( s down K fF C  . ( 17
)
As seen previously, the capacitance of the switch changes between the up-state and the down-state. Now, analysis of the two last static parameters, the series inductance and the series resistance, is done. The evolution of the series inductance (cf. figure 9) confirms the model predictions. The inductance decreases according to 1/K s , which corresponds to model equation [START_REF] Qian | A parametric model of low-loss RF MEMS capacitive switches[END_REF]. From the fitted graph (y = 30x -1 ), the obtained parameter L 1 is 30 pH (Theory: 32 pH) and the inductance is Figure 10 shows that the series resistance of the shunt switches increase linearly. That differs from the first resistance equation ( 7) that assumes a constant value (same shape for all MEMS). Indeed, in this case, the skin effect is very pronounced when the frequency approach 100 GHz.

s K pH L 30 ) (  . ( 18
The fitted equation (y = 0.33x) allows to write that:

s K R    33 . 0 ) ( . (19)
In the next figure (Fig. 11), the evolution of the actuation voltage is shown. It is interesting to note that V pull-in increases linearly with K s (y = 7.8x) as indicated in (20a):

x s x in pull K V V , , 8 . 7 ) (    (20a) and x s in pull x in pull K V V , 1 , ,     . ( 20b
)
Combining (20b) and ( 15) yields the scaling law that applies to the effective spring constant k x as

1 1  k k x , (21a)
and thus For a triple actuation, the effective spring constant seems to be constant for all switches and the "pull-in" voltage is very low (7-22 V).

1 k k x  . (21b)
Note that, in Fig. 11, the "pull-in" voltage for a single central actuation (20-35 V) is also reported for each switch.

Figure 12 shows that the resonance frequency (y = 20x 1.5 ) fits well with the electrical model [START_REF] Marcelli | Mechanical modelling of capacitive RF MEMS shunt switches[END_REF]. This result depends directly on the result for C down .

The relationship To resume, from ( 16), ( 17), ( 18), ( 19) and ( 23) the parameters of the experimental model can be expressed as functions of frequency by:

2 3 0 20 s K f  (22) is obtained from the fit. So 3 2 0 20        f K s . (23)
3 4 0 20 151 ) (          f fF C up , ( 24
) 3 4 0 20 2239 ) (          f fF C down , ( 25
) 3 2 0 20 30 ) (          f pH L , ( 26
) 3 2 0 20 33 . 0 ) (          f R and (27)
3 2 0 20 100 ) (          f m l TLIN  . ( 28
)
with f 0 (GHz) being the desired resonance frequency of the MEMS.

In the previous equations, it is interesting to note that all model parameters depend only on f 0 so the model is truly frequency scalable.

The insertion loss (fig. 13) of the switches decreases according to the scaling parameter K s . Compared to the 20-GHz MEMS, the insertion loss of the 77-GHz MEMS is very small (about one quarter). The isolation (fig. 13) decreases, but the value remains below 27 dB at resonance frequency. The graphs presented in this section simplify considerably the task of parameterizing the model for a future RF MEMS switch working at another frequency.

Consider the case of the design of an RF MEMS based impedance tuner at 50 GHz. The first step is to design a 50-GHz MEMS. With the model presented, it is not necessary to fabricate a test structure of the MEMS. It is sufficient to get the scaling parameter K s from fig. 12. In this particular case, in, the K s obtained is 1.8. Then, the value can be applied to the other graphs to find all model parameters (C up = 30 fF; C down = 500 fF; L = 18 pH; R = 0.6 Ω).

Note that, while the graphs presented depend in part on the technology, the methodology can be easily transferred.

V. APPLICATIONS

One future use of the proposed design methodology is to realize phase shifters based on RF MEMS. This approach allows to model and predict the behavior of the entire phase shifter. This circuit is needed for smart antennas for wireless sensor networks.

A 1-bit V-band phase shifter loaded line type has been developed from the shown RF MEMS model.

For this first design, the circuit has been designed to provide a phase shift of 22.5°. It uses capacitive shunt switches and 50 Ω GCPW access lines. It has been optimized for operation at 60 GHz. The designed loaded-line phase shifter (Fig. 14) operates in class III [START_REF] Harry | Circuit design of the loaded-line phase shifter[END_REF]. The circuit uses two RF MEMS switches. The main line is loaded with two stubs whose lengths depend on the state of the switches (l 1 +l 2 in up-state, l 1 in down-state). This load difference provides the two output phase states. The stub lengths are calculated based on:

  90  , ( 29
)   s Y B 1 1 1 tan    , ( 30
)   1 2 1 2 tan      s Y B , (31)
 

2 tan 0 2 1     Y B B , (32)
where θ is the electrical length of the line l, θ 1 the electrical length of the line l 1 , θ 2 the electrical length of the line l 2 , B 1 and B 2 are the complex-conjugate loading susceptances, Y s the characteristic admittance of the stub line, and ∆Φ the desired phase shift.

The Class III architecture was chosen because it has a constant loss per bit and relatively constant phase response.

The Fig. 15 shows the good performance of the 60-GHz phase shifters designed for future antenna array systems. The measured phase shifter presents a low insertion loss (1.1 dB in l 1 +l 2 state and 1.9 dB in l 1 state) and a good return loss (24 dB in l 1 +l 2 state and 11 dB in l 1 state) at 60 GHz. Compared to the phase shifters of Tab. I, the circuit presents state-of-the-art performance.

The measured phase shift is 22.2°, quite close to the 22.5° modeled. 18 phase shifters have been measured: 13 (72%) are functional. The phase shifts in worst case are 17.1° and 32.7°, respectively. These large variations are due to the fact that the used RF MEMS technology is still under development.

VI. CONCLUSION

This paper presents an approach to millimeter-wave circuit design based on MEMS models. These models are completely frequency scalable over the entire millimeter-wave band, by using the methodology based on the scaling parameter K s . Indeed, the measured switches exhibit good characteristics very close to the ones obtained by the developed MEMS capacitive shunt switch model. This model was used to design high performance circuits based on RF MEMS shunt switches, for example the presented 60-GHz loaded-line phase shifter.

  and the steps are described below.(a) Each process begins with a cleaning step to remove the chemical oxide. (b) On the clean silicon substrate, a 20 µm thick BenzoCycloButene (BCB) layer is deposited.

Fig. 1 .

 1 Fig. 1. RF-MEMS micro-switch fabrication process.

Fig. 2 .

 2 Fig. 2. Microphotograph of the fabricated reference switch.

Fig. 3 .

 3 Fig. 3. Electrical model of the RF MEMS capacitive shunt switch.

Fig. 4 .

 4 Fig. 4. Insertion loss and isolation of the reference MEMS (K s =1).

Fig. 5 .

 5 Measured capacitance versus voltage for 60-GHz MEMS.

Fig. 6 .Fig. 7 .

 67 Fig.6. RF MEMS microphotographs (using the same objective).

Fig. 8 .

 8 Fig. 8. Capacitance in up-state and down-state versus K s .

Fig. 9 .

 9 Fig. 9. Series inductance versus K s .

Fig. 10 .

 10 Fig. 10. Series resistance versus Ks.

Fig. 11 .

 11 Fig. 11. Actuation voltage versus Ks.

Fig. 12 .

 12 Fig. 12. Resonance frequency versus K s .

Fig. 13 .

 13 Fig. 13. Insertion loss and isolation versus K s .

Fig. 14 .

 14 Fig. 14. Fabricated 60-GHz loaded line phase shifter. Electrical schematic (a) and microphotograph (b).

  . S-parameter measurements of the 60-GHz loaded line phase shifter. Insertion loss (a), return loss (b) and insertion phase (c).

l 1 l 2 l

 2 At millimeter wave frequencies, a similar approach is also possible for DC-contact switches. Further extensions towards rapid prototyping of RF MEMS based circuits like tuners and phase shifters can be prospected.Patrick Pons received his "Doctorat" in electronics from Paul Sabatier University, Toulouse, France, in 1990. Since 1991 he is a researcher in the CNRS-LAAS (National Scientific Research Center -Laboratory of Analysis and Architecture of Systems) in Toulouse.His research interests are focused in microtechnology and microsensors. In 1995, he started in the lab the study on microtechnology for microwave applications. Now he manages the development of this technology for high frequency Microsystems and develops also pressure sensors for specific applications.RobertPlana (M'98-SM'08) was born on March 1964 In Toulouse. He obtained its PhD in 1993 at LAAS-CNRS and Paul Sabatier University on the Noise modeling and characterization of Advanced Microwave devices (HEMT, PHEMT and HBT) that includes the reliability. In 1993, as associate professor at LAAS-CNRS, he has started a new research area concerning the investigation of millimeterwave capabilities of Silicon based technologies. More precisely, he has focussed on the microwave and millimeterwave properties of SiGe devices and their capabilities for low noise circuits. In 1995, he has started a new project concerning the improvement of the passives on silicon through the use of MEMS technologies. In 1999, he has been involved with SiGe Semi-conductor in Ottawa where he was working on the low power and low noise integrated circuits for RF applications. In the same year, he has received a special award from CNRS for his works on Silicon based technologies for millimeterwave communications. In 2000, he has been professor at Paul Sabatier University and Institut Universitaire de France and he has started a research team at LAAS-CNRS in the field of Micro and Nanosystem for RF and millimeterwave communications. Its main interests are on the technology, design, modelling, test, characterization and reliability of RF MEMS for low noise and high power millimeterwave applications and the development of the MEMS IC concept for smart microsystem. He has built a network of excellence in Europe in this field "AMICOM" regrouping 25 research groups. He has authored and co-authored more than 200 international journals and conferences. In 2004, he has been appointed as Deputy Director of the Information and Communication Department at the CNRS Headquarter. From January 2005 to January 2006, he has been appointed director of the Information and Communication Department at CNRS. He is now heading a research group at LAAS-CNRS in the field of Micro and Nanosystem for wireless communications.

TABLE I

 I

		STATE-OF-THE-ART OF DIFFERENT PHASE SHIFTERS
	Frequency	Type/bits Phase	Insertion	Bulk	Area	Ref.
	(GHz)		error	loss		(mm²)
				(dB)
	75-110	MEMS/3	3°	-2.7	Glass	5x1.9	[1]
	50-70	MEMS/2	5°	-4.6	Quartz	2.1x1.5	[2]
	30-38	0.12u	7°	+1	Si	0.9x0.4	[3]
		BiCMOS/4
	18-26	0.13u	9.7°	-3.8	Si	0.8x0.6	[4]
		CMOS/4
	55-65	65nm	9.2°	-9.4	Si	/	[5]
		CMOS/4

  Table III and the capacitance values extracted from S 21 graphs (cf. sections III.B. and III.C) because the wafer measured isn't the same in both cases.

ACKNOWLEDGMENT

The authors would like to thank A. Rumeau and J. Lattes for measurement and T. Beluch for Matlab assistance.


Mohamed Mehdi Jatlaoui

Daniela Dragomirescu

Samuel Charlot

Patrick Pons

Hervé Aubert

Robert Plana

3D Heterogeneous Integration of Wireless Communicating Nano-Sensors on Flexible Substrate

Keywords: 3D Heterogeneous Integration, Nano-Objects, Flexible Substrate, Flex Assembly, Wireless sensor networks

des établissements d'enseignement et de recherche français ou étrangers, des laboratoires publics ou privés.

INTRODUCTION

The "Internet of thing" domain arouses many research efforts in Europe and in the world [START_REF] Dragomirescu | Wireless Communicating Nano-Objects for Structure Health Monitoring as Enabler for Safer, Greener Aircrafts[END_REF]. This domain aims at spreading ultra miniaturized sensor networks presenting the most low consumptions and offering the advantage of being localized in areas with access difficulty (in particular in non planar area) or even in inaccessible places, with conventional technologies. Designers have exploited the inherent flexibility to design flex circuits to be flexurally deformed during assembly, adjustment, and repair procedures, to have complex three-dimensional final geometries, and to sustain multiple flexural cycles during functional life [START_REF] Fjelstad | Flexible circuit Technology[END_REF]. Taking into account the integration in difficult access zones, it is recommended to exploit the potentialities of nano-objects for wireless nano-components realization. Nanotechnologies made it possible the development of ultra-sensitive nano-sensors based on nanoparticles deposition [START_REF] Reda | [END_REF][4]. Transceivers become more and more miniaturized and one can investigate assembly techniques to postpone them on flexible substrate. One can also take benefits from nano-ink with silver nanoparticles or with nanotube of carbon to develop antennas on flexible substrate. These antennas can be directly integrated on the same flexible substrate as the developed nano-sensors and miniaturized transceivers, which is the very innovative approach of this work. How, then, does one choose an assembly process and method for flex circuits? To begin appropriately, it is necessary to consider a number of important parameters: What is the flex circuit base material? What types of components will be used? How many assemblies will be built? These and other important questions must be addressed before one can adequately make the proper decisions regarding assembly. In the following sections, answers to these questions are proposed.

DESCRIPTION OF THE COMMUNICATING NANO-OBJECT

For the development of reliable and reconfigurable communication architectures, the aeronautical domain imposes severe specifications dealing with the reliability, the electromagnetic compatibility, the interferences immunity, the consumed power and the determinism of protocols. From the system integration point of view, the research work presented here aims at assembling various elements constituting the communicating nano-object (as shown in Figure1): nano-sensors, transceivers and E/R antenna. For that purpose, an innovative solution will be developed. This solution consists in the integration on flexible substrate of these various components realized in different technologies. The nano-sensors are based on nano-particles deposition. The sensor can be done directly on the flexible substrate or on silicon and then reported on the flexible substrate. Different kind of sensors can be integrated on this substrate, as constraint gauge [5], crack wire or chemical / humidity detection sensor [START_REF] Boyer | Modelling of urea-EnFETs for haemodialysis applications[END_REF]. The transceivers are composed of mixers, VCOs and LNAs realized in 65nm technology. These components will be reported on the substrate by flip-chip. Finally, the antenna made of CNTs ink deposition and designed at 60 GHz is directly printed on the substrate. The 60 GHz frequency is chosen for the short connections and the very short distances to limit multi-path effect and to minimize the energy losses induced by the dispersive radiation pattern. Here, the challenge deals with the choice of the flexible substrate, the pads materials and the interconnection architecture to minimize the RF losses and parasitic effects. This paper brings the first response elements and presents a proof of concept for such heterogenous integration.

FLEX CIRCUIT BASE MATERIAL AND CIRCUIT DESIGN

As mentioned in the previous section, the communicating nano-object is composed of various components realized in different technologies. Each component has specific constraints: Indeed, the nanosensors impose constraints on the roughness, mechanical properties and surface condition. As for Transceiver and antennas, they require good thermal properties (for the postponement by FlipChip) and good RF properties (to minimize the parasitic effect, for the RF interconnections). Based on these criteria, two solutions can be envisaged. The first is to achieve a postponement on a same substrate but with the need for an intensive study to improve the surface condition. The second solution is to make the sensor on a flexible substrate first and then the transceiver and antenna are added on a second substrate. Fig. 2 Selected solutions for the assembly In order to demonstrate the feasibility of such integration, the second solution was chosen for the remainder of this paper. For the nanosensors, the selected substrate is PET. This substrate has a good surface state. However, a 130µm Kapton substrate was chosen for the transceiver and antennas because of its good RF and thermal properties. For physical, thermal properties and effect of humidity please refer to [7]. After achieving, the components follow a variety of tests. In a first step, non-destructive flaw detection means will be used to check if all the bumps are connected properly and that there is no breaking problem between the chip and substrate. RF tests and tests to characterize the bumps resistivity are also planned (Daisy chain, four probe method). To perform these tests, circuits on Kapton were designed to receive the chip after postponement by FlipChip. Given the cost of chips made with 65nm technology, VCOs and LNAs dummies must also be manufactured. Some of the designed circuits (on kapton and on Silicon) are shown in figure 3.

ASSEMBLY PROCESS

For the work reported here, a polyimide flexible substrate made of Kapton is chosen. Packaging of the transceivers and the Nano-sensors on the Kapton film is schematically illustrated in Figure 4. First, a polymer spin coating is performed for the adhesion of the polyimide on the holding wafer. Second, polyimide was patterned on the substrates by lamination. The polyimide is a Kapton film with a 130µm thick. Third, resin lift spin coating is realized. Fourth, a photolithography is done for the Cu/Ni/Au UBM metallization placement. The UBM metallization is realized by lift-off. Finally, after a passivation layer deposition, soldering screen printing and reflow are performed before the substrate peeling. The different elements interconnections are performed using Cu lift off process. Fig. 4 Processing methods for copper circuits (a) polymer spin coating for the adhesion; b) polyimide lamination; c) resin lift spin coating; d) photolithography; e) Cu/Ni/Au UBM metallization resin stripping by lift off; f) passivation layer; g) soldering screen printing and reflow; h) substrate peeling)

REALIZED TEST STRUCTURES

Adopting the process shown in the previous section, test structures were performed. In figure 5, realized circuits on kapton are presented. One can see the location for the postponement of the dummy chip (VCO or LNA) with the passivation layer. In a first step, the antennas on kapton have been performed separately following the same assembly process. One can see from the figure below (Figure6), different patch antennas made on Kapton. One can also distinguish resonant circuits that are capable of measuring the real permittivity of Kapton. This value will be used for permittivity retro-simulations and the future design of interconnections between the RF Transceiver and antennas.

Fig. 6 Realized antennas and resonators on Kapton Dummy chips were also fabricated. These chips have the same access that real components, but with connections to allow RF tests. On Figure 7, one can see the diced dummy chips on which studbumps were deposited.

CONCLUSION

Flexible circuit technology opens doors to new opportunities for engineers and system designers to make a complete transition to the 3D interconnection. Flexible circuits are obviously unique among electronic packaging technologies to offer a wide variety of solutions where compactness and heterogeneous integration, of different components on the same substrate, are needed. In this paper the very first and unique results presenting the potentialities of flexible substrate for the heterogeneous integration of communicating Nano-Objects are presented. The choice of the technological process (selection of materials and choice of process technology) has been discussed. The design and fabrication of test structures for the verification of the RF interconnections quality have been made. The design and implementation of antennas on Kapton substrate was also performed. Future works will address the characterization of the realized structures under real working conditions. Real case applications (on aeronautical composite components) are being investigated by the authors to bring a proof-of-exploitation of these new wireless Nanosensor nodes.

Fig. 1

 1 Fig. 1 Concept of the communicating nano-object on flexible film

Fig. 3

 3 Fig. 3 Designed circuits on Kapton and Silicon dummies (VCOs, LNAs and Daisy chain chip)

Fig. 5

 5 Fig. 5 Realized circuits on Kapton (location for the postponement of the dummy chip)

Fig. 7

 7 Fig. 7 Realized Silicon dummies (VCOs and LNAs)

Acknowledgments

This work has been supported by the French National Research Agency (ANR) in the frame of its program" Recherche Technologique Nano-INNOV/RT" (NANOCOMM project n°ANR-09-NIRT-00 4).


Daniela Dragomirescu

Michael Kraemer

Mohamed Mehdi Jatlaoui

Patrick Pons

Hervé Aubert

Andrew Thain

Robert Plana

60GHz Wireless Nano-Sensors Network for Structure Health Monitoring as Enabler for Safer, Greener Aircrafts

Keywords: Structure Health Monitoring, Wireless Sensor Networks, Safer, smarter, green aircrafts, 60GHz communication, ultra low power 65nm CMOS circuits, Flexible substrate integration

des établissements d'enseignement et de recherche français ou étrangers, des laboratoires publics ou privés.

INTRODUCTION

The major driver for the aircraft industry development [START_REF] Rosner | Smart structures contribution to Airbus aircraft efficiency[END_REF] nowadays is the eco-efficiency. That means the need of the development of future greener aircrafts with lowest carbon emission which pass by less weight in the aircraft while keeping the passenger comfort. The new generation of aircraft has to be also high performance while staying cost efficiency. That will be a global system challenge to develop the new generation aircrafts which can be solved only by a global system solution: rethink all the system architecture of the aircraft.

In this context, structure health monitoring (SHM) becomes a very important issue. The deploying of a whole sensor network on the structure of the aircraft is necessary. Nowadays, it is estimated that there are around 20000 aircrafts in activity and this number has to double by the end of 2025 and more than 1000 sensors are necessary for each one. The facility of sensor deployment becomes also a main issue.

The introduction of sensor network for structure health monitoring will improve the maintenance system and will lead to safer aircrafts, which is another major topic together with eco-efficiency and also to lower airplane ticket cost for passengers. It is estimated that a decrease of 3% to 12% in the prices of airline tickets can be expected. Today the deployment of the sensors is limited by wired connections which lead to excess weight and carburant consumption (the major driver is eco-efficiency, greener aircrafts). The wired sensor put also a huge problem of installation. The wireless solution, wireless sensor network, is imposed by these major constraints.

In the next paragraphs, we will first present the needs of structure health monitoring system, then the proposed network architecture, and we will focus on the ultra low power nano-technology circuits for 60GHz communications and the flexible substrate integration.

STRUCTURE HEALTH MONITORING

The aeronautics is a very exigent industrial field with very high requirements in term of reliability. Compared with the ground transportation, the aircrafts accidents are one thousand times lower. It is estimated that two thirds of aircrafts failure could be eliminated or minimized by a high performance maintenance process and in the future a real time control of the flying aircraft. That shows the importance of the developing a structure health monitoring system. Structure health monitoring will reduce maintenance effort, increase aircrafts availability and can also reduce weight in the future. In the Airbus vision [START_REF] Rosner | Smart structures contribution to Airbus aircraft efficiency[END_REF], it will be five generations, five main steps in the development of structure health monitoring systems. The "Generation 0" structure health monitoring systems, available since 2003 are dealing with structure testing application with benefits for structure analysis and testing. The "Generation 1" structure health monitoring system recently available it consist on a off-board equipment with benefit for maintenance. The "Generation 2" structure health monitoring system will be on-board equipment and the benefit will be the maintenance, the diagnosis and weight saving component level. This system is expected by 2015 -2018. The "Generation 3" structure health monitoring system will be a fully integrated system with benefit for the weight saving aircraft level, the maintenance and the availability and it will include predictive diagnosis. This system is expected by 2018 -2025. The "Generation 4" structure health monitoring system will be implemented on the aircrafts designed using smart materials, self-healing composite materials [START_REF] Trosk | [END_REF] [3]. The SHM system will diagnose the problem and the smart materials of the structure of the plane will be able to heal itself [2] [3]. This system will be expected in the very far future.

In this paper we focus on the development of the 2 nd and later the 3 rd generation of structure health monitoring system. The main problem which has to be solved by the structure health monitoring system is the hard landing (Figure 1). The goals are to reduce the number of false reports, aiding the maintenance process and so reduce the aircraft schedule interrupts. The current maintenance process is initiated by the pilot by "feel" of the hard landing and then the maintenance team will verify the aircraft by visual inspection and recently using off-board equipment (Figure 2). The number of false report of hard landing is around 85% today [START_REF] Young | Structural Health Management, an Evolutionary Journey[END_REF] which lead to an important immobilization of the aircrafts. After deploying SHM system the maintenance process will change. The pilot by "feel" of the hard landing will still initiate the inspection. Flight parameters and structure health monitoring sensor information will be used to predict load information in critical structure areas and recommend maintenance action. Theses informations will also aid maintenance actions. Using this SHM system will decrease the false report down to 5% and lead to an important increase in the availability of the aircraft.

Figure 2. Aircraft inspection after hard landing and maintenance process

In perspective, the SHM system will also allow to keep a component history report and lead to predictive diagnosis.

STRUCTURE HEALTH MONITORING SYSTEM REQUIREMENTS

It is clear now that we need to develop sensor networks for the SHM system. Wired sensor network can't be used because of two main problems: difficulty of deployment and the increase of the weight of the aircraft. So, the wireless solution imposed itself. It has to be a modular communicating object which accepts different kind of sensors. The communication architecture has to support a very high number of communicating nodes. In such a small area like the aircraft cabin this will lead to carefully choice of the frequency to decrease the interferences. Moreover, no interference between this wireless sensor network and the passenger equipment are required. The data rate demand for this application is low or medium. Some measurements of the sensors has to be done in the same moment for all the sensors, so a synchronization measurement protocol is also needed. One important requirement is low power consumption as the wireless communicating objects has to work for years. The easy replacement of the autonomous power source, i.e. the battery, for this communicating nodes or the easy replacement of the all communicating node is also a strong requirement. The last major requirement is a complete independent instrumentation with respect to the aircraft flying instrumentation. Nevertheless, this wireless sensor network has to be able to connect to aircraft network All this requirements can be satisfied by COTS which are usually designed for medium number of nodes, without measurement synchronization protocol and communicating in the same bands as the passenger equipment which will lead at high interferences and can lead at huge security problem. Moreover, the COTS solutions are not so small and can't be placed on curve surfaces and not so low power either. For all these raisons, we propose here a new architecture and technology for wireless communicating objects.

ROBUST COMMUNICATION ARCHITECTURE

We propose to use the 60GHz band to short distance instrumentation and a flexible substrate integration which allow us to instrument event the difficult access areas on the aircraft. We have chosen the 60GHz band for many reasons. First of all, the transmission range at that frequency is short and the directivity of the wave is higher which will lead to lower interferences between the communicating nodes, a major requirement for our system: high number of communicating nodes in a small area. Second, the 60GHz is a free band. And the last, but not the least, the evolution of the CMOS technology in the nanotechnology area allow us today to design circuits in CMOS for 60GHz communication enabling low cost communicating nodes and a large scale production. Figure 3 represents the communicating object as we proposed. On the flexible substrate we will integrate the sensor element, the sensor signal processing, the communication circuit (digital baseband, RF front-end) and the antenna. The communicating node has to be seen as a modular platform which can include different kind of sensors. The sensor can be done directly on the flexible substrate or on silicon and then reported on the flexible substrate. Different kind of sensors can be integrated on this substrate, as constraint gauge [START_REF] Ressier | How to control AFM nanoxerography for the template monolayered self-assembly of 2 nm colloidal gold nanparticules[END_REF], crack wire or chemical / humidity detection sensor [START_REF] Boyer | Modelling of urea-EnFETs for haemodialysis applications[END_REF]. To assure very low power transceivers and low interferences we use Ultra Wide Band Impulse Radio (UWB-IR) communications which will be describe in the next section. The low power millimeter wave front end circuits design enabled by the use of CMOS nano-technology will be presented in section 5. The flexible substrate chooses here for the integration is a polyimide called Kapton. The detailed technological process set up and the technological breakthroughs are presented in [START_REF] Jatlaoui | Heterogeneous integration of Communicating Nano-Objects on Flexible Substrate[END_REF].

From communication architecture point of view the next question to solve is how we can interrogate and retry data from the wireless communicating nodes. We propose a 3D heterogeneous integrated router scanner which will be placed in an area with more available energy (maybe wired powered). This router will use the scanner principle to interrogate the communicating sensor nodes and realize the measurements as represented in Figure 4. The router will contain the same digital base band circuits and RF front end circuits developed in 65nm CMOS technology use a special concern for low power as the communicating node. On Figure 4, one can see that the router will scan the different communicating nodes. To realize the scanning function we will use our own MEMS RF phase shifters [START_REF] Puyal | Frequency Scalable Model for MEMS Capacitive Shunt Switches at Millimeter Wave Frequencies[END_REF] (see Figure 5), coupled with array antenna on silicon. The smart antenna (phase shifter and antenna elements) will be integrated in 3D packaging on the top of the CMOS transceiver.

ULTRA WIDE BAND IMPULSE RADIO TRANSMISSION

The aeronautic application impose severe constrains for the reliability, the electromagnetic compatibility, the immunity of the interferences and the power consumption. All theses constraints lead us to choose to develop short distance wireless communications using ultra wide band impulse radio technique (UWB-IR) followed by a RF front-end at 60GHz. Figure 6 presents the principle of UWB-IR communications versus narrowband communications.

Figure 6. Ultra wide band impulse radio communications versus narrowband communications

We demonstrate the very robust functioning of the UWB-IR in the context of our application by implementing a prototype of the full transceiver (emitter and receiver) completely reconfigurable on the FPGA with very good results [START_REF] Lecointre | Largely reconfigurable impulse radio UWB transceiver[END_REF]. A data rate of 125 Mbps was achieved [START_REF] Lecointre | Largely reconfigurable impulse radio UWB transceiver[END_REF], overcoming the state of the art.

The UWB-IR emitter developed is completely digital and able to generate the UWB-IR pulse, pulse modulation, spread spectrum and regulate the emitted power. The one bit digital emitter was implemented in ASIC using CMOS 65nm technology from ST Microelectronics. The fabricated UWB-IR emitter is presented in Figure 6 and the measurements show a maximum achievable data rate of 375Mbps for a band pass of 2.25GHz and a consumption of only 1.4pJ/bit. We notice here the achievement of this design: very low power consumption allowed by the nano-metric technology used and by a clever, simple design.

60GHz LOW POWER, LOW COST DIRECT CONVERSION ARCHITECTURE

As presented in section 3, the 60GHz communications presents many major advantages for our wireless sensor network for SHM application. The complete System on Chip (SoC) architecture of the full transceiver, including UWB-IR base band and millimeter wave front end is presented in Figure 7.

Figure 7. System on chip architecture of the full transceiver to be implemented in nano-metric CMOS Different architectures for the millimeter wave front-end were studied [START_REF] Kraemer | Architecture Considerations for 60 GHz Pulse Transceiver Front-ends[END_REF], and for our application the best choice is the direct conversion architecture presented in Figure 8. The direct conversion architecture will lead to a small size transceiver, a very low power one and a low cost transceiver if implemented in CMOS. These three criteria are major criteria in Wireless Sensor Network for SHM (see section 2.1) Moreover, is an interesting challenge to design 60GHz completely CMOS circuits. Thanks to the nanotechnologies, the reduction of the gate size of the CMOS transistor in deep nanometric circuits was possible. This leads to the increase of cutoff frequency in the silicon device, achieving the CMOS employment in this slice of market. After the 130nm generation, the CMOS device has prevailed the III-V group materials in particular for commercial applications. The Monolithic Microwave Integrated Circuits (MMICs) have been left to military and spatial applications. The Radio Frequency Integrated Circuits (RFICs) had become the new low cost commercial way.

We designed the main blocs of our transceiver in CMOS 65 nm from St Microelectronics. The major optimization criterion was the low power consumption of each bloc. The complete millimeter wave transceiver has to consume less than 100 mW. The first step was the design of the inductances (see Figure . 9) with very good performances. [START_REF] Kraemer | Accurate Electromagnetic Simulation and Measurement of Milimeter-wave Inductors in Bulk CMOS Technology[END_REF], [START_REF] Kraemer | On the De-embedding of Small Value Millimeter-wave CMOS Inductor Measurements[END_REF] s=2µm w=3µm 100 µm Figure 9. 60GHz Inductances designed in 65nm CMOS We focus our design on the key blocs of the front-end. The Voltage Controlled Oscillator (VCO) at 60GHz was designed using a differential Colpitts architecture. The obtained VCO present very high power efficiency, the rapport between the output power and the DC power is 3.65, which is more than 3 times the value presented in the state of the art. A new Low Noise Amplifier (LNA) was designed using two cascade stages and spiral inductances to minimize the size and the cost. The gain is 18.7dB and the power consumption is 8.5mW for 1V bias [START_REF] Kraemer | A low-power high-gain LNA for the 60 GHz band in a 65 nm CMOS technology[END_REF].

Figure 11. 65nm CMOS low power LNA A dual-gate 60GHz direct up-conversion mixer with active IF balun in 65nm CMOS was also designed (Figure 12). A good conversion gain G c = -4.1dB was measured for a DC power of 23mW [START_REF] Kraemer | A Dual-Gate 60GHz Direct up-Conversion Mixer with Active IF Balun in 65nm CMOS[END_REF].

Figure 12. Up-conversion mixer in 65nm CMOS

The use of the 65nm CMOS technology allow us to design low-power, low cost blocs at 60GHz.The next step is the complete integration of the very low power millimeter wave front-end which is under fabrication. To obtain the communicating node of the Wireless Sensor Network (Figure 3), the CMOS transceiver will be added by Flip Chip on the flexible substrate (Figure 13). The 60GHz antenna will be designed directly on the flexible substrate. The choice of the flexible substrate has to be done very carefully to enable good radiation diagram for the antenna at 60GHz. 60GHz transmission lines between the transceiver and antenna has to be done. The Flip Chip of the 60GHz transceiver on the flexible substrate is a new way to package circuits at 60GHz. The small dimension of the pads, the 60 GHz connections and the low temperature process impose new technological methods described in [START_REF] Jatlaoui | Heterogeneous integration of Communicating Nano-Objects on Flexible Substrate[END_REF]. .

CONCLUSION

Structure Health Monitoring is a main concern nowadays to build safer, greener aircrafts. The main requirements of the wireless communicating nodes are low interferences with passenger equipment and between communicating nodes, high reliability, low power and low cost. The capacity of placing these nodes on the all aircraft surfaces, including non planar one is a major request. We presented in this paper new wireless sensor network architecture, circuits and packaging able to answer the SHM requirements. The 60GHz communication will allow high number of nodes in such a small area like the aircraft cabin. The design of low power and low cost 60GHz transceiver is enabled by the use of nano-metric CMOS technology. The complete integration of the sensor communicating node was done on flexible substrate. The achievement of the wireless communicating nodes for SHM is enabled by the use of the nanotechnologies in the design of the transceivers, antenna and integration on flexible substrate.

Fig. 1

 1 Fig. 1 Hard landing problem

Figure 3 .

 3 Figure 3. Wireless Communicating Node on Flexible Substrate

Figure 4 .

 4 Figure 4. Architecture of router-scanner at 60GHz

Figure 5 .

 5 Figure 5. MEMS RF phase shifters designed and fabricated in the clean room at LAAS-CNRS

Figure 8 .

 8 Figure 8. Direct conversion architecture of the 60Ghz front-end

PFigure 10 .

 10 Figure 10. High power efficiency 60GHz VCO in 65nm CMOS

Figure 13 .

 13 Figure 13. Kapton-100HN flexible substrate ready for the Flip Chip report of the 60GHz blocs

Acknowledgments

This work has been supported by the French National Research Agency (ANR) in the frame of its program" Recherche Technologique Nano-INNOV/RT" (NANOCOMM project n°ANR-09-NIRT-004).