Carlos Carvalho

Nuno Paulino
email: nunop@uninova.pt

A Voltage Limiter Circuit for Indoor Light Energy Harvesting Applications

Keywords: CMOS integrated circuits, Energy harvesting, Voltage limiters, Voltage reference circuits

A voltage limiter circuit for indoor light energy harvesting applications is presented.This circuit is a part of a bigger system, whose function is to harvest indoor light energy, process it and store it, so that it can be used at a later time. This processing consists on maximum power point tracking (MPPT) and stepping-up, of the voltage from the photovoltaic (PV) harvester cell. The circuit here described, ensures that even under strong illumination, the generated voltage will not exceed the limit allowed by the technology, avoiding the degradation, or destruction, of the integrated die. A prototype of the limiter circuit was designed in a 130 nm CMOS technology. The layout of the circuit has a total area of 23414 µm 2 . Simulation results, using Spectre, are presented.

Introduction

To achieve indefinite operation in any location, sensor nodes must obtain their power directly from the environment. In a wireless sensor network, it may be difficult to power each node, either using batteries or the power grid. Sensor networks supplied by grid connections are limited to a small range of applications, as they can never be too far from the power outlet. Although the use of batteries allows for more freedom of sensor distribution, its replacement can become burdensome and costly, if a large number of sensors are deployed. As such, energy harvesting systems tend to take over the powering paradigm for pervasive operation [START_REF] Paradiso | Energy scavenging for mobile and wireless electronics[END_REF], beingalso ecological and thrifty. The sources that can be harvested are diverse [START_REF] Chalasani | A survey of energy harvesting sources for embedded systems[END_REF] and, amongst those, light is the one with the highest energy density per unit of volume [START_REF] Chalasani | A survey of energy harvesting sources for embedded systems[END_REF].Energy can also be obtained from indoor light. In this ambient, powering an electronic application is an increased challenge, as the levels of available light power inside buildings are much lower than those obtained outside. At the most, the Sun can provide about 1 kW/m 2 , but in indoor environments, the Sun and the artificial lighting provide a much lower energy density.

Relationship to Internet of Things

The circuit presented in this paper is an essential part of an indoor light energy harvesting system. This energy harvesting will allow a variety of electronic systems to operate indoors without needing batteries or a connection to the power grid, adding an enormous degree of freedom, regarding their location. These electronic systems can communicate with other devices and, ultimately, connect to the Internet. The light energy source, when compared to other sources, has the highest energy density, making the harvester discreet and small. Light energy harvesters are simple and cheap to build. As long as light is available, any device supplied by energy harvested from ambient light can be deployed to anywhere, making it a part of the Internet of Things.

Motivation and Background

The unpredictable nature of light energy, results in a variable amount of available power. Since the PV cells must be sized in order for the energy harvesting system to be able to receive enough energy even when the light intensity is weak, this can result in large power, available from the same PV cells, when the light intensity is strong. The circuit described in this paper, fits in a system designed to deal with very low levels of light energy, typically found in indoor environments. However, when this system is illuminated by direct sunlight or generally, by high levels of light intensity, the energy harvested by the PV cells is much higher than the value that was expected under the normal indoor illumination conditions. In this situation, the energy harvesting system must be able to manage the excess of energy that is obtaining.

The architecture of the system is depicted in Fig. 1,being constituted by PV cells followed by a voltage step-up converter with MPPT capability. The output voltage of the converter (v dd ) is the one that will be limited by the voltage limiter. The v dd voltage also supplies the inner circuits of the step-up converter. The system can work with one or two PV cells in series and step-up the input voltage (v in ) by two or three, depending on the type of PV cells and on the expected light intensities. The power from the PV cell is dependent on its area and on the illumination level, as shown in Fig. 2.According to a set of light power intensity measurements, the lowest illumination obtained indoors had an ambient irradiance of 0.1 W/m 2 and the typical irradiance using artificial lighting, was about 0.7 W/m showing the power function that corresponds to the lowest irradiance level.

The system must be designed for a worst case scenario (low illumination), meaning that when the light is stronger, the system will produce a large voltage at its output.

The system was designed in a 130 nm CMOS technology, having a maximum voltage of about 1.4V.If this value is exceeded, the devices in the die are stressed, reducing their operating lifetime. With a larger voltage, they can even be destroyed.

The v dd voltage can be used to power any application, for example, a sensor node. Since the available power from the PV cells is low, this application typically will work using an ON-OFF regime with a low duty-cycle. While the application is OFF, the harvested energy is stored in a supercapacitor. When the application is ON, it works using this stored energy. In the case of a high light intensity, after the supercapacitor is charged to a desired voltage value, the step-up converter continues to supply a current to the output node (v dd ). This results in the output voltage to increase to a value that can be dangerous, being necessary to add a circuit to limit it, by absorbing the excess current supplied by the step-up converter.

In literature, some work can be found concerning voltage limiters. In [START_REF] Silva-Martinez | Adjustable CMOS voltage limiters for low-voltage applications[END_REF], the voltage limitation is achieved by opening a switch to a capacitor, whenever its voltage exceeds a certain limit, on a sample-and-hold basis. There are other systems that also use voltage limiter circuits, like in [START_REF] Kim | A design of transceiver for 13.56 MHz RFID reader using the peak detector with automatic reference voltage generator and voltage limiter[END_REF] and [START_REF] Fernandez | Low power voltage limiter design for a full passive UHF RFID sensor[END_REF], where the voltage limitation problem appears under the context of RFID applications. The amount of available power to a RFID tag is dependent upon the distance to the reader device, resulting in a similar problem to the one in a light energy harvesting system. In [START_REF] Kim | A design of transceiver for 13.56 MHz RFID reader using the peak detector with automatic reference voltage generator and voltage limiter[END_REF], the voltage limitation is done by performing a comparison to a desired limit. In [START_REF] Fernandez | Low power voltage limiter design for a full passive UHF RFID sensor[END_REF], a band gap reference circuit is used to generate a reference voltage to which the desired voltage is compared to. This approach is similar to the one proposed in this paper.

Voltage Limiter Circuit Architecture

The architecture of the voltage limiter is shown in Fig. 3. The Thévenin equivalent, at the right hand side of the dashed line, represents the voltage step-up converter. The aim of the circuit is to limit the v dd voltage, to be lower than 1.4 V -5% = 1.33 V. The voltage limiter operates by drawing current through M 1 . This current causes a voltage drop across R test , such that v dd remains constant. The value, at which v dd is to be limited, is controlled by a Voltage Reference Circuit (VRC), providing a stabilized and temperature compensated voltage reference.

In normal operation, the voltage divider formed by R 1 and R 2 , provides at v vdd , a voltage close to v REF . The amplifier A 1 amplifies the error between v vdd and v REF , providing, at its output, a voltage that directly controls the v gs voltage of M 1 , and thus the current drawn from the v dd node, through R D .

Voltage Reference Circuit

The VRC was adapted from [START_REF] Magnelli | [END_REF], and its schematic is shown in Fig. 4 M 2 and M 12 are high voltage transistors (3.3 V), while the others are low voltage ones (1.2 V). The main difference between the original circuit, and the one that was built, is the fact that the one in [START_REF] Magnelli | [END_REF] was built in a 180 nm CMOS technology, whereas the present one uses a 130 nm technology. The transistors were sized based on the sizes presented in [START_REF] Magnelli | [END_REF], making only adjustments due to the different technologies. The layout of the VRC circuit occupied 15647 µm 2 . In normal operation, considering that v dd = 1.2 V and a temperature of 27 ºC, the nominal output voltage of the VRC is 230.9 mV. The supply current, in this case, is 139.5 nA. These data are depictable in Fig. 4 b). In Fig. 5, the supply current and the reference voltage values, as a function of temperature, for different supplying voltages, are shown.

Differential Voltage Amplifier

The topology of the amplifier is shown in Fig. 6.This amplifier has a power down feature (pd) to disable it, in order to turn off the voltage limiter circuit. The layout of the amplifier occupied 7767 µm 2 .The input stage of this amplifier uses PMOS devices, as the typical values of the input voltage are around 200 mV to 300 mV.

Stability Analysis

Since this is a closed loop system (Fig. 3), it is important to analyze its stability. The

v dd = 1.8 V v dd = 1.5 V v dd = 1.2 V v dd = 0.9 V v dd = 0.6 V v dd = 0.3 V ) ( ) ( 2 1 2 1 2 1 2 p c c dd vdd C C R sR R R R R sC R v v s + + + + = = β . ( 1
)
The impedance,Z 22 , of the feedback network from v dd to ground, is

) 1 )( 1 ( ) ( ) ( 2 1 2 1 2 1 22 p c p c C sR C sR C C R sR R R s Z + + + + + = . ( 2
)
The small-signal model of the amplifier, including the output transistor M 1 , is depicted in Fig. 7.As the output variable, i out , is the current drawn from the v dd node, and the input variable is the input differential voltage of the amplifier (v x ), there will be a transconductance function G M (s) given by ) )( ( ) ( The impedance seen from v dd to ground, defining

R o = (g dsM1 ) -1 + R D , is . // 1 // ) ( ) ( 22 o st out R sC s Z s Z       = (4)
The loop of the system in Fig. 3was open before the "+" terminal of A 1 . By injecting signal in the "-" terminal, and computing the ratio to the signal obtained at the output of the feedback network, the loop gain is

). ( ) ( ) ( ) ( s s Z s G s G out M L β = (5)
This function has four poles and one zero. Their expressions are as follows, considering that C st , connected to the output node, is much higher than C c and that C p :

( ) . 2 1 ; ) )( // ( 2 1 ; ) //( 2 1 ; 2 ; 2 1 2 1 4 2 1 3 2 2 2 1 1 1 c z p c p st o p out ds p out ds p C R f C C R R f C R R R f C g f C g f π π π π π = + = + = = = (6)
The values of f p4 and f z can be approximately equal, by selecting an appropriate value for C c , thus canceling out each other, making the system equivalent to having only the other three poles. This is useful in improving the phase margin of the system.

Assuming that R 1 + R 2 >> R o , which is the case, the DC loop gain is given by .

) ( ) 0 ( 2 1 2 1 2 3 2 1 R R g g R R g g g G ds ds o m m m L + = (7)
The magnitude and phase of the open loop gain are determined through an AC sweep analysis and shown next in Fig. 8a) and b), respectively. C c is swept from 0.8 pF to 2.0 pF, to check what would be the consequence on the phase margin of the system. In these simulations, the value of the storage capacitor, C st , was 1.8 µF. It can be seen, in the close up inset in Fig. 8 a), that regardless of the value of C c , the zero crossing of the loop gain function occurs at about 32 kHz. As seen by the close up inset in Fig. 8 b), at the previous frequency, the spreading of phase values according to the value of C c , results in a minimum phase margin value of 57º. In order to have an acceptable phase margin value of 60º, the value of C c was selected to be 1.4 pF. It is important to note that, if the value of C st is increased, the phase margin will also increase. The value of 1.8 µF is still a relatively small value for a storage capacitor, as it can get to units of Farads, or more, thus corresponding to a worst case.

Simulation Results

To check the behaviour of the circuit, it was simulated in Spectre. In Fig. 9 a) and b), there are depicted the results of a DC sweep and a transient response, respectively, to show the correct operation in different ways. The function in Fig. 9 a) was obtained by running a DC sweep, using V test , on the right hand side of the dashed line in Fig. 3.The function in Fig. 9b) was obtained by using a square wave voltage generator for V test . As seen in both Fig. 9a) and b), the voltage limiter comes into action whenever the voltage in the v dd node tends to be higher than the limit. The voltage at which the limitation in v dd is achieved, is about 1.31 V, which is inside the desired limit. In these tests, the supply current of the amplifier was, typically, about 900 nA.

Conclusions

A voltage limiter for indoor light energy harvesting applications has been presented.This circuit is a part of a bigger system, whose function is to harvest indoor light energy, process it and store it so that it can be used at a later time. This system can be used as the supplying module for a node of the Internet of Things.

The circuit that was described, ensured that even under a strong illumination, the generated voltage would not exceed the maximum value allowed by the technology, to avoid stressing, or destroying the devices in the die. Simulation results, using Spectre, were presented and showed that the proposed circuit can successfully limit the output voltage (v dd ) to the desired ceiling.

Fig. 1 .

 1 Fig.1. An energy harvesting architecture that hosts the voltage limiter described in this paper.

Fig. 2 .

 2 Fig.2. a) Available power from the PV cell with 1 cm 2 , for different irradiance levels. b) Detailshowing the power function that corresponds to the lowest irradiance level.

2 Fig. 3 .

 23 Fig.3. Architecture of the voltage limiter described in this paper.

Fig. 4 .

 4 Fig.4. a) CMOS Voltage Reference Circuit. b) Output reference voltage (v REF ), and supply current, as a function of v dd .

Fig. 5 .

 5 Fig.5. Temperature dependence of the supply current, and generated voltage reference, for different supplying voltages, for the VRC that is used.

Fig. 6 .

 6 Fig.6. Differential amplifier circuit.

  feedback network is constituted by R 1 , R 2 , C c and C p (parasitic capacitance at the amplifier input). The feedback factor, β, is given by

Fig. 7 .

 7 Fig.7.Small-signal model of the amplifier and the output M 1 transistor.

Fig. 8 .

 8 Fig.8. Magnitude and phase functions of the feedback loop circuit ofFig.3.

  a) DC sweep response. b) Dynamic transient response, with a period of 15 ms.


Vitor Lopes
email: vlopes@ualg.pt

Diogo Bento

Maria C R Medeiros
email: cmedeiro@ualg.pt

Paula Laurêncio

Optical Strategies for Generating and Transmitting Signals in OFDM-RoF Systems

Keywords: Radio over fiber, Orthogonal Frequency Division Multiplexing, 60 GHz, fiber chromatic dispersion, Optical Single Side Band, optical upconversion

systems which act as backbone networks are very promising means of reducing overall costs of 60 GHz WPANs, as well leading to networks with limited intercellular interference. The main objective of this paper is to compare the performance of two systems that generate Optical Single Side Band (OSSB) OFDM signals at the mm-wave band with and without optical up-conversion. Both systems use a DD-MZM: one is a standard model, whereas the other is a DD-MZM cascaded with a phase modulator (PM). The comparative study presented here also includes practical impairments such as the finite extinction ratio of DD-MZMs.

Introduction

The 60 GHz (mm-wave) band has been allocated worldwide for short range wireless communications due to its inherent high propagation losses. The abundant unlicensed spectrum around the 60 GHz has the potential to support consumer demands for indoor wireless applications that require high bandwidth such as real-time streaming content download for high-definition TV, wireless gigabit Ethernet, etc. Due to the small coverage area, 60 GHz networks require a large number of base stations (BSs) to cover a service area. This requirement has led to the development of system architectures where functions such as routing, signal processing, handover and frequency allocation are performed at the central office (CO). The best solution for connecting the CO with BSs in such radio network is via an optical fiber network, now known as radio over fiber (RoF) [START_REF] O'reilly | Broadband wireless systems and networks: an enabling role for radio-over-fibre[END_REF].

Due to the high data rates of broadband wireless systems, we can have severe time-dispersion effects associated to the multipath propagation. Multicarrier (MC) modulation schemes combined with frequency-domain receiver implementations, especially the ones belonging to the orthogonal frequency division multiplexing (OFDM) class are widely used in several broadband wireless communication systems which have to deal with strong frequency-selective fading channels; as such, they are considered a good contender for this application. However, the optical distribution of high bit rate mm-wave signals is susceptible to fiber chromatic dispersion, which severely limits the transmission distance. This shortcoming can be mitigated using optical single-sideband (OSSB) modulation [START_REF] Laurêncio | Dynamic range of optical links employing optical singleside-band modulation[END_REF].Several techniques have been proposed to generate mm-wave optical signals. Among them, optical frequency upconversion has the potential to provide a simple and cost efficient solution [START_REF] Lin | Ultra-high data-rate 60 GHz radioover-fiber systems employing optical frequency multiplication and OFDM formats[END_REF]. The principle of optical frequency up-conversion is to generate higher-order optical harmonics using an external nonlinear modulator, typically a dual-drive Mach-Zehnder modulator (DD-MZM) driven by a low frequency electrical signal. The efficiency of this technique relies not only on its capacity to maximize the required high-order harmonic, but also on its robustness against fiber dispersion. Therefore, the objective of this paper is to compare two systems that generate OSSB signals at the mm-wave band with and without optical up-conversion. Both systems use a DD-MZM: one is a standard model and other is a DD-MZM cascaded with a phase modulator (PM) [START_REF] Thakur | Optical frequency tripling with improved suppression and sideband selection[END_REF].

The remainder of this paper is organized as follows. In section 2 we discuss the relationship to the Internet of things. In section 3, we present the network architecture. In section 4, we present the simulation results and finally, in section 5, we conclude the paper.

Relationship to Internet of Things

Nowadays, high bandwidth services are supported by wired networks based on fiber optic technologies. However, consumer habits are increasingly changing; as such, users want to be able to use their mobile terminals and enjoy the same user experience as they do when connected to their fixed networks. The abundant unlicensed frequency spectrum around 60 GHz offers the potential for multi-gigabit indoor Wireless Personal Area Networks, removing the bandwidth limitation of the present wireless networks.

The current generation of Wi-Fi systems (IEEE 802.11n) can achieve theoretically 600 Mb/s, but new radio standards allowing higher data rates have emerged or are under development. All these new standards address the unlicensed mm-wave band from 57 to 66 GHz, in this band, divided into four 2.16 GHz-bandwidth channels, it is possible to achieve bit rates up to 7 Gb/s per channel [START_REF] Hansen | Wigig: Multi-Gigabit Wireless Communications in the 60GHz Band[END_REF]. IEEE 802.15.3c [START_REF]WPAN Task Group 3c[END_REF] and WirelessHD standards were finalized in 2009 and are devoted to wireless personal area network communications.More recently, the IEEE 802.11ad group has been created by WiGig consortium: it addresses networking with wireless local area network systems and targets Wi-Fi Alliance certification. 60 GHz technologies are now mature: WirelessHD devices are already available in the market for wireless highdefinition multimedia interface (HDMI) applications and IEEE 802.11ad products.

Due to the high-propagation attenuation and the fact that signals in the 60 GHz band cannot cross walls,the premises should be connected employing optical fiber technology. Consequently, 60 GHz networks require an extensive high capacity feeder network; thisdemands an optical fiber network thatprovides the required performance, while being cost-effective.

Currently, wired and wireless services are separately provided by two independent physical networks. However, with the fast evolution of wireless technology, ubiquitous and always on wireless systems in buildings are expected to emerge in the near future. Hybrid broadband access networks combining wireless and wireline Wavelength Division Multiplexed Passive Optical Networks (WDM-PONs) networks can support the requirement of these future broadband ubiquitous networks. The target of a converged hybrid architecture is to connect, via a common optical WDM-PON infrastructure, both wireline and wireless users. Hybrid wireless WDM-PON architecture integrating mm-wave wireless services within WDM-PONs will be the future access networks.

System Description

The architecture of the 60 GHz OFDM-RoF numerical simulation platform using the VPI Transmission Maker Simulator® is shown in figure 1 and2. The difference between the two figures is the optical external modulator; the first one is a DD-MZM biased in quadrature, whilst the second one is a DD-MZM biased in quadrature cascaded with a phase modulator (PM), both of them generating an OSSB signal. The objective of the second configuration is to demonstrate that the effect of the finite extinction ratio of the DD-MZMs on OSSB modulation can be greatly reduced. The light source considered in the simulation is a laser with 1 MHz of line width and 5 mW of emission power with -150dB/Hz relative intensity noise (RIN). A 2.5 Gbit/s NRZ data is divided in 512 blocks. Each blockis presented as 512parallel data paths to the OFDM transmitter. These 512paths aremodulated onto 512equally-spaced subcarriers using 4 Quadrature-Amplitude Modulation(4QAM). The cyclic prefix (CP) is 12.5%. Each QAM data channel is presented to an input of an IFFT that produces a complex-valued time domainwaveform containing a superposition of all of the subcarriers. This waveform therefore is modulated onto an electrical carrier, f c =60GHz, using an I-Qmodulator, producing a real-valued signal. This carrier gives a band of optical OFDM sub-carriers with a bandwidth of 750 MHz centered in the optical carrier at 193.1 THz plus 60 GHz after external modulation by a DD-MZM. Both DD-MZMs were assumed to have a 6 dB insertion loss and a 30dB extinction ratio (R ext ). The fiber attenuation was set to 0.2 dB/km and the dispersion coefficient to 16 ps/(nm.km). At the receiver side, the signal is directly detected by an ideal highspeed PIN photodetector, amplified and each QAM channel is demodulated to produce 512 parallel data channels. These can be converted into a single data channel by parallel-to-serial conversion and down-converted to recover the 2.5 Gbit/s data. The optical field ( ) MZ E t at the output of the DD-MZM can be expressed as:

( ) ( ) ( ) ( ) ( ) 1 2 2 A A A V t V t V t j j j VA VA VA MZ i E t E t e e e π π π π π π η ξ         = + +           (1)
Where E i is the optical field at the input of the DD-MZM and average power Po, VA π is the switching voltage of the DD-MZM; η and ξ are given respectively by 1 2 r r and ( )( )

1 2 1 2 1 1 r r r r - - -
, where and r 1 and r 2 are the power splitting ratios of the input and output of the Y-junctions of the DD-MZM [START_REF] Lin | Impact of Nonlinear Transfer Function and Imperfect Splitting Ratio of MZM on Optical Up-Conversion Employing Double Sideband With Carrier Suppression Modulation[END_REF] respectively. and a dc term, given by:

( ) ( ) ( ) ( ) 1 2 / 4 ˆ/ 4 A OFDM A OFDM V t s t VA V t s t VA π π = - = + (2)
Where ( ) ˆOFDM s t is the Hilbert transform of the OFDM signal. In the case of DD-MZM cascaded with an optical phase modulator (PM) as illustrated in Fig. 3 the optical field E o is given by

( ) ( ) ( ) ( ) ( ) ( ) 1 2 2 A A A B V t V t V t V t j j j j VA VA VA VB o i E t E t e e e e π π π π π π π π η ξ         = + +          
(3) While the voltage applied to the PM modulator is assumed to be ( ) B V t ; that is the modulated OFDM signal after being applied a adjustable phase shift β+π/2. The objective of this phase shift is to eliminate the residual sideband due to DD-MZM finite extinction rate (ER).

Optical Strategies for Generating and Transmitting Signals in OFDM-RoF Systems 503

Fig. 3. Schematic of the ER compensation scheme.

The principle of optical frequency up-conversion is to generate higher-order optical harmonics using an external nonlinear modulator, typically a Mach-Zehnder modulator (MZM) driven by a low frequency electrical signal, this reduces the bandwidth requirements for the optical modulator, the electrical drive signal source and the drive circuit itself. For example, for OSSB, frequency tripling is obtained by beating, upon square law detection by the photodetector, the ω c harmonic with ω c +3ω m , and all harmonics separated by 3ω m , where ω c is the optical carrier frequency, and ω m is the subcarrier frequency. For dual-arm MZM, the effect of the imbalance in the power splitting ratio generates odd low sideband harmonics as it is shown in Fig. 4a. For this situation, the imbalance leads to an optical spectrum that approaches an Optical Double Side Band (ODSB) signal. In the case of the DD-MZM cascaded with an optical phase modulator the optical spectrum approaches the ideal modulator spectrum (Fig 4b).

Simulation Results

Optical modulation without up-conversion

In order to compare the performance of both modulator schemes when digital data is transmitted, the DD-MZM and the DD-MZM cascaded with a phase modulator OSSB systems illustrated in Fig. 1 and 2 were simulated using the VPI TransmissionMaker TM simulation platform. The simulation parameters were presented in section 3. This includes the noise system, like RIN, shot noise and circuit noise. We named it the real circuit. We also performed an additional test to calculate the beta parameter in the phase modulator, in order to determine the angle at which the optical carrier in the lower sideband is completely suppressed. Intermodulation distortion that generates inter-subcarrier interference at fiber output arises mainly from the nonlinear characteristics of the optical modulator, the linear fiber dispersion and from the square law detection process. Figure 5.a illustrates the Error Vector Magnitude (EVM) versus modulation depth (x c ) for 25km of fiber length. We can conclude the optimal modulation depth is similar for all tested configurations. For lower modulation depths, the system noise is dominant, whereas for higher modulation depths the intermodulation distortion between subcarriers is dominant; this leads to an optimum modulation depth x c =0.155, in the circuit with a DD-MZM, and x c =0.111, in the circuit with a DD-MZM, plus a PM. From Fig 5.b, we can conclude the EVM does not change linearly with the fiber length. There are also periodic oscillations that result from the dispersion, making the EVM increase with the fiber length. This occurs due to the interplay between the laser RIN and the dispersion. When we don't have noise (only the dispersion) the EVM is periodic. Finally, in Fig. 5.c, when we eliminate the attenuation we can see the EVM decreases slightly because the interplay between the RIN and the dispersion is dominant. This result can be confirmed in Fig. 5.d where the received constellation is presented (without attenuation) for different fiber lengths. From these results we concluded that OSSB based systems are robust against imperfect power splitting ratios and the use of an extra PM to eliminate the first low sideband harmonic is not necessary.

Optical modulation with Up-Conversion

In this section we present the results obtained for the two modulation schemes studied, but using optical up-conversion with frequency tripling. The conditions are the same from last section and we use an electrical 20GHz to obtain 60GHz in frequency tripling. Comparing the performance of the systems using up-conversion with the ones from previous section, we can conclude that the results are significantly worse. This is explained on Fig. 6.b. and 6.c. In these figures we can see the dispersion is the main responsible for the performance degradation obtained. This occurs because up-conversion systems have different harmonics that suffer different dispersion ef-fects when the fiber length increases.Furthermore, the beating during the photodetection process results in additional system degradation. This effect combined with the laser RIN can be seen in Fig. 6.d, where the receiver constellation is shown for different fiber lengths. We can conclude that this system needs equalization for distances beyond 15km. The use of the cascaded PM does not solve the problem.

Conclusions and further work

We concluded that OSSB based systems are robust against imperfect power splitting ratios and the use of an extra PM to eliminate the first low sideband harmonic is not necessary when 60GHz native frequency is used. When frequency tripling is used, this system needs equalization for distances beyond 15km. We also concluded that the dispersion is the mainly responsible for the performance degradation of the system when compared with the 60GHz natives. The use of the cascaded PM does not solve the problem.

In further work analytical methods will be developed to evaluate the degradation of performance of those systems caused by laser intensity noise at fiber output, intermodulation distortion and fiber dispersion. In case of optical up-conversion equalization techniques will be applied.

Fig. 1 .Fig. 2 .

 12 Fig.1. System Architecture of a 60 GHz OFDM-RoF with DD-MZM

Fig. 4 .

 4 Fig.4. Optical power spectrum at a) Dual-Arm MZM (Rext=30dB)b) Dual-Arm with PM (Rext=30dB)

Fig. 5 .

 5 Fig.5. EVM for DD-MZM and DD-MZM cascaded with PM versus a) modulation depth for 25km of fiber length b) fiber length real conditions versus only dispersion c) fiber length real conditions versus without attenuation d) QAM constellation diagram.

Fig. 6 .

 6 Fig.6. EVM for DD-MZM and DD-MZM cascaded with PM with up-conversion versus a) modulation depth for 25km of fiber length b) fiber length real conditions versus only dispersion c) fiber length real conditions versus without attenuation d) QAM constellation diagram.


Manuela A Vieira

Vitor Silva

Paula Louro

Manuel Barata

Optoelectronic Logic Functions Based on Reconfigurable SiC Multilayer Devices

Keywords: Optoelectronic logic functions, Reconfigurable devices, Numerical simulation, Transfer characteristics, Coder-encoder devices, Encoder truth tables

WDM multilayered SiC/Si devices based on a-Si:H and a-SiC:H filter design are approached from a reconfigurable point of view. Results show that the devices, under appropriated optical bias, act as reconfigurable active filters that allow optical switching and optoelectronic logic functions development. Under front violet irradiation the magnitude of the red and green channels are amplified and the blue and violet reduced. Violet back irradiation cuts the red channel, slightly influences the magnitude of the green and blue ones and strongly amplifies de violet channel. This nonlinearity provides the possibility for selective removal of useless wavelengths. Particular attention is given to the amplification coefficient weights, which allow taking into account the wavelength background effects when a band needs to be filtered from a wider range of mixed signals, or when optical active filter gates are used to select and filter input signals to specific output ports in WDM communication systems. A truth table of an encoder that performs 8-to-1 multiplexer (MUX) function is presented.

Introduction

Optical communication in the visible spectrum usually interfaces with an optoelectronic device for further signal processing. There has been much research on semiconductor devices as elements for optical communication [START_REF] Connelly | Semiconductor Optical Amplifiers[END_REF][START_REF] Petit | Photodiodes with integrated optical filters for passive optical network applications[END_REF][START_REF] Ibrahim | Fully reconfigurable silicon photonic lattice filters with four cascaded unit cells[END_REF].

Multilayered structures based on amorphous silicon technology are predictable to become reconfigurable to perform WDM optoelectronic logic functions and provide photonic functions such as signal amplification and switching [START_REF] Yang | OFDM Radio-over-Fibre Systems Employing Routing in Multi-Mode Fibre In-Building Networks[END_REF]. They will be a solution in WDM technique for information transmission and decoding in the visible range [START_REF] Yang | 47.4 Gb/s transmission over 100m graded-index plastic optical fiber based on rate-adaptive discrete multitone modulation[END_REF]. Amplification and amplitude change are two key functionality properties outcome of a balanced interaction between wavelength of the optical signal and background wavelength and placement within a WDM link. Any change in any of these factors will result in filter readjustments. An algorithm to decode the information is presented and support new optoelectronic logic architecture.

Internet of Things

Future advances in computer speeds are contingent on augmenting intra-and interchip bandwidth. Increases in power efficiency per bit of data is projected to be achieved by replacing electrical interconnects with their optical counterparts in the near future. The implementation of photonic circuits for the purpose of chip-level communications requires devices such as modulators, detectors and wavelength division multiplexers (WDM).

Reconfigurable multi-rate next generation optical networks are currently investigated to handle the ever increasing growth of the Internet traffic. Reconfigurable wavelength selectors that allow for operation on a large number of wavelength channels, with dynamic response, are essential sub-systems for implementing reconfigurable WDM networks and optical signal processing.

In this paper, we demonstrate that is possible, using the amorphous Si/C technology, to produce reconfigurable SiC multilayer devices acting simultaneously as photodetectors, selectors and active filters. Therefore, the theoretical approach presented in this research is useful in examining a particular aspect of the optical communications, configuring firm activities towards innovation and combining unique resources and capabilities. Devices based on new materials for signal (de)multiplexing in the visible spectrum are an important contribute to the future development of the Internet of Things.

Device Configuration and Operation

The device consists of a p-i'(a-SiC:H)-n/p-i(a-Si:H)-n heterostructure with low conductivity doped layers (Fig. 1). The thicknesses and optical gap of the front í'-(200 nm; 2.1 eV) and back i-(1000 nm; 1.8 eV) layers are optimized for light absorption in the blue and red ranges, respectively [START_REF] Vieira | Three Transducers Embedded into One Single SiC Photodetector: LSP Direct Image Sensor, Optical Amplifier and Demux Device[END_REF][START_REF] Louro | Use of a SiC:H Photodiodes in Optical Communications Applications[END_REF]. As a result, both front and back pin structures act as optical filters confining, respectively, the blue and the red optical carriers. Several monochromatic pulsed lights, separately (λ R,G,B input channels) or in a polychromatic mixture (multiplexed signal) at different bit rates illuminated the device from the glass side. Steady state optical bias with different wavelength are superimposed (400nm-800 nm) from the front or from the back sides and the generated photocurrent measured at -8 V. The device operates within the visible range using as input color channels (data) the wave square modulated light (external regulation of frequency and intensity) supplied by a red (R: 626 nm; 51 µW/cm 2 ), a green (G: 524 nm; 73 µW/cm 2 ) and a blue (B: 470 nm; 115 µW/cm 2 ) LED. Additionally, steady state red, green, blue and violet (background) was superimposed by different LED's (R: 625 µW/cm 2 , G: 515 µW/cm 2 , B: 400 µW/cm 2 , V: 2800 µW/cm 2 ).

Photonic Active Filters

The spectral sensitivity was tested through spectral response measurements under different frequencies, with and without steady state optical bias applied either from the front or back side. In Fig. 2, the spectral photocurrent (symbols) is displayed under red, green, blue and violet background and without it. In Fig. 2a the steady state optical bias was applied from the front side and in Fig. 2b from the back side. For comparison the normalized spectral photocurrent for the front, p-i'-n, and the back, p-i-n, photodiodes (dash lines) are superimposed.

Data shows that the front and back diodes, separately, presents the typical response of single p-i-n cells with intrinsic layers based on a-SiC:H or a-Si:H materials, respectively. The front diode cuts the wavelengths higher than 550 nm while the back one rejects the ones lower than 500 nm. The overall device presents an enlarged sensitivity when compared with the individual ones.

Under front irradiation the sensitivity is much higher than under back irradiation. Under front irradiation (Fig. 2a) the violet background amplifies the spectral sensitivity in the visible range while the blue optical bias only enhances the spectral sensitivity in the long wavelength range (>550 nm) and quenches it in the others. Under red bias, the photocurrent is strongly enhanced at short wavelengths and disappears for wavelengths higher than 550 nm. Under green the sensitivity is strongly reduced in all the visible spectra. In Fig. 2b, whatever the wavelength of the backgrounds, the back irradiation strongly quenches the sensitivity in the long wavelength range (>550 nm) and enhances the short wavelength range. So, back irradiation, tunes the front diode while front irradiation, depending on the wavelength used, can tune the back one.

In Fig. 3 the spectral gain, defined as the ratio between the spectral photocurrents under red (α R ), green (α G ) blue (α B ) and violet (α V ) illumination and without it is plotted at 3500 Hz. The optical bias is applied from the front side, in Fig. 3a and from the back side, in Fig. 3b. Under front bias and red irradiation, the gain is high at short wavelengths and strongly lowers for wavelengths higher than 500 nm, acting as a short-pass filter. Under green background, the device behaves as a band-stop filter that screens out the medium wavelength range (475 nm-550 nm) enhancing only the photocurrent for wavelengths outside of that range. Under blue and violet light the device works as a long-pass filter for wavelengths higher than 550 nm, blocking the shorter wavelengths. Back light, whatever the frequency, leads to a short-pass filter performance. Results show that by combining the background wavelengths and the irradiation side, the short-, medium-and long-spectral region can be sequentially tuned.

Encoder and Decoder Device

To analyze the device under information-modulated wave and uniform irradiation, three monochromatic pulsed lights separately (red, green and blue input channels, Fig. 4) or combined (multiplexed signal, Fig. 5) illuminated the device at 6000 bps. Steady state violet optical bias was superimposed separately from the front (solid lines) and the back (dash lines) sides and the photocurrent generated measured at -8 V. The transient signals were normalized to their values without background. Results show that, even under transient conditions, the background side affects the signal magnitude of the color channels. Under front irradiation, it enhances mainly the light-to-dark sensitivity in the medium-long wavelength ranges. Violet radiation is absorbed at the top of the front diode, increasing the electric field at the least absorbing cell, the back diode. Taking into account Fig. 4, the red and green collections are strongly enhanced (α V G =2.2, α V R =3.1) while the blue one stays near its dark value (α V B =1.1). Under back irradiation the small absorption depth of the violet photons across the back diode quenches the electric field and so, the red collection almost disappears (α V R =0.2). Here, blue channel is absorbed across the front diode where the electric field is enhanced resulting in an increase collection of the blue channel (α V B =1.6). Since the green channel is absorbed across front and back diodes its collection is balanced by the increased collection in the front diode and its reduction at the back one (α V G =0.7). For an optoelectronic digital capture system, opto-electronic conversion is the relationship between the optical inputs and the corresponding digital output levels. In Fig. 5 the normalized multiplexed signal under front (symbols) and back (lines) violet irradiation is displayed. On the top of the figure the signals used to drive the input channels are shown to guide the eyes into the ON/OFF channel states. 0,0 0,2 0,4 0,6 0,8 1,0 1,2 1,4

0,0 0,3 0,7 1,0 1,4 0 0 0 0 0 B 0 G 0 0 G B R 0 B R 0 0 R G 0 R G B S G S B S R
x 0 x 1 x 2 x 3 x 4 x 5 x 6 x 7 MUX signal (a.u.) Time (ms) Fig. 5. Normalized multiplexed signal under front (symbols) and back (lines) violet irradiation.

On the top the signals used to drive the input channels are shown to guide the eyes into the ON/OFF channel states.

Under front irradiation (pin 1 ;

α V R >>1, α V G >1 and α V B ~1
) the 2 3 levels can be grouped into two main classes due to the high amplification of the red channel. The upper four levels are ascribed to the presence of the red channel ON, and the lower four to its absence, allowing the red channel decoder (4-to-1 multiplexer; long-pass filter function). Since under front irradiation the green channel is amplified the two highest levels, in both classes, are ascribed to the presence of the green channel and the two lower ones to its lack.

Under back irradiation (pin 2 ; α V R <<1, α V G <1 and α V B >1, Fig. 4), the blue channel is enhanced; the green reduced the red almost suppressed. The encoded multiplexed signal has the eight sublevels grouped also into two main levels, the higher where the blue channel is ON and the lower where it is OFF (4-to-1 multiplexer; short-pass filter function). In each main level the four existent sublevels are grouped in two classes, with and without the channel green ON. Each of those sublevels split into two near ones, attributed to the presence or absence of the red channel. If we consider this red output bit "not significant" only four separate levels (2 2 ) are considered and the logic MUX function is converted into a logic filter function. The blue channel is then decoded. This decoding algorithm is mapped in the right side of the multiplexed signal using a RGB code for each level (Fig. 5).

Like regular binary numbers, the binary RGB code is an arithmetic code and so, it is weighted, i. e. there is specific weights assigned to each bit position. Under front violet irradiation, the most significant digit, the left most bit, in the RGB code, is the red (α V R >>1). Going from the left to right, the next is the green (α V G >1) and the last is the blue (α V Bpin1 ~1). Under back violet irradiation, the left most bit is the blue, the next is the green and last the red.

The truth tables of both encoders of Fig. 6, that perform 8-to-1 MUX function, are shown in Fig. 6. In the inputs (x 0 ….x 7 ), the index of each bit, is related to the first (highest) nonzero logic input. Here, the MUX device selects, through the violet background, one of the eight possible input logic signals and sends it to the output (y=x S ). The output is a three-bit [S 2 S 1 S 0 ] binary RGB number that may identify one of eight possible inputs.

x 0 x 1 x 2 x 3 x 4 x 5 x 6 x 7 0 0 0 0 0 0 0 1 0 0 0 0 0 0 1 0 0 0 0 0 0 1 0 0 0 0 0 0 1 0 0 0 0 0 0 1 0 0 0 0 0 0 1 0 0 0 0 0 0 1 0 0 0 0 0 0 1 0 0 0 0 0 0 0

S 2 S 1 S 0 1 1 1 1 1 0 1 0 1 1 0 0 0 1 1 0 1 0 0 0 1 0 0 0 Inputs Outputs Y MUX DEMUX a)
x 0 x 1 x 2 x 3 x 4 x 5 x 6 x 7 0 0 0 0 0 0 0 1 0 0 0 1 0 0 0 0 0 0 0 0 0 1 0 0 0 1 0 0 0 0 0 0 0 0 0 0 0 0 1 0 0 0 1 0 0 0 0 0 0 0 0 0 1 0 0 0 1 0 0 0 0 0 0 0

S 2 S 1 S 0 1 1 1 1 1 0 1 0 1 1 0 0 0 1 1 0 1 0 0 0 1 0 0 0 Inputs Outputs Y MUX DEMUX S 2 S 1 S 0 1 1 X 1 1 X 1 0 X 1 0 X 0 1 X 0 1 X 0 0 X 0 0 X Y' b)
Fig. 6. Truth tables of the encoders that perform 8-to-1 multiplexer (MUX) function, under (a) front and (b) back violet irradiations (x means " not significant").

Results show that the correspondence between the on/off state of the input channels S R , S G , S B in Fig. 5 and S 2 , S 1 , S 0 , outputs in Fig. 6, are obvious. In Fig. 6, for the input x 7 and output S 2 , the first nonzero logic input is 7 (2 2 +2 1 +2 0 ), which corresponds an output [111]. Those OR gates are expressed, respectively, as: S 2 =x 7 +x 6 +x 5 +x 4 under front irradiation (Fig. 6a) and S 2 =x 7 +x 3 +x 5 +x 1 under back light (Fig. 6b).

Conclusions

Tandem WDM a-SiC:H pi'n/pin devices are analyzed under different front and back optical bias wavelengths.

Results show that those devices, under appropriated optical bias act as reconfigurable active filters that allows optical switching and optoelectronic logic functions development. Depending on the wavelength of the external background the device acts either as a short-or a long-pass filter or as a band-stop filter.

An optoelectronic digital capture system based on this device is analyzed and the relationship between the optical inputs and the corresponding digital output levels established. A truth table of an encoder that performs 8-to-1 multiplexer (MUX) function is presented.

Fig. 1 .

 1 Fig. 1. Device configuration and operation.

Fig. 2 .

 2 Photocurrent (nA)

Fig. 3 .

 3 Fig. 3. Spectral gain under red (α R ), green (α G ), and blue (α B ) optical bias, applied from the front (a) and the back (b) sides at 3500Hz.

Fig. 4 .

 4 Fig. 4. Normalized red, green and blue transient signals at -8V with violet (400 nm) steady state optical bias applied from the front and the back sides.

Acknowledgements. This work was supported by PTDC/EEA-ELC/111854/2009, PTDC/EEA-ELC/115577/2009 and PTDC/EEA-ELC/120539/2010.


Dora Gonçalves

L Miguel Fernandes

Paula Louro

Manuela Vieira

Alessandro Fantoni

Measurement of Photo Capacitance in Amorphous Silicon Photodiodes

Keywords: capacitance, photodiode, amorphous silicon

This paper discusses the photodiode capacitance dependence on imposed light and applied voltage using different devices. The first device is a double amorphous silicon pin-pin photodiode; the second one a crystalline pin diode and the last one a single pin amorphous silicon diode. Double amorphous silicon diodes can be used as (de)multiplexer devices for optical communications. For short range applications, using plastic optical fibres, the WDM (wavelength-division multiplexing) technique can be used in the visible light range to encode multiple signals. Experimental results consist on measurements of the photodiode capacitance under different conditions of imposed light and applied voltage. The relation between the capacitive effects of the double diode and the quality of the semiconductor internal junction will be analysed. The dynamics of charge accumulations will be measured when the photodiode is illuminated by a pulsed monochromatic light.

Introduction

Research Question and Motivation

The research question that supports the study presented in this paper is: "Is it possible to control the photo capacitance of the double pin-pin photodiode?"

The methodology used to give answer to this research question consists on the comparison of a double pin-pin photodiode based on a-Si:H/a-SiC:H with an amorphous silicon pin device and a crystalline pin diode through the measurement of capacitance-voltage and photocurrent-voltage characteristics. We are interested in the influence of the material and device structure on the photodiode capacitance in order to improve the performance of these devices for WDM applications in the visible range.

State of the Art and Related Literature. Methodological Approach

In a crystalline silicon pn junction there are two types of capacitances, related to the charge stored in the depletion layer (junction capacitance) and to the diffusion of carriers along the junction (diffusion capacitance) [1].

When we consider the pn junction under reverse bias conditions, there is an increase in the width and charge of the depletion layer; as the voltage across the device changes, the charge stored in the depletion layer changes as a result. We can observe a correspondence between a capacitor and the depletion layer of a pn junction; the charge stored on either side of the layer is a non linear function of the applied reverse voltage. Under a small-signal approximation the depletion capacitance, or junction capacitance, is the gradient of this curve, at the bias point. On the other hand, we can consider the depletion layer as a parallel-plate capacitor and obtain the expression for its capacitance, using the traditional procedure. When the pn junction is under forward-bias conditions, the depletion layer narrows and the depletion barrier voltage reduces; the diffusion current increases until equilibrium is achieved. In the steady state, a certain amount of excess minority carriers charge is stored in each of the bulk pn regions. If the terminal voltage changes, this charge will have to adjust, before a new steady state is achieved. This charge storage occurrence introduces another capacitive effect, which is directly proportional to the diode current: we are referring the diffusion capacitance.

Nowadays, hydrogenated amorphous silicon and hydrogenated amorphous silicon carbide have found a large application in electronic devices. In this type of pn junctions, we must have special attention to the midgap density of states and junction properties, to obtain the correct expression of the capacity. The pin configuration is frequently used in photo devices; the i-layer may be alloyed with carbon to optimize the band gap for a given application. May be the simplest and most widely applied techniques to characterize mid gap states (i.e. deep levels) in semiconductors are admittance measurements. The imaginary part of the admittance, i.e. the capacitance, is directly related to the charge trapped in or released from mid gap states, as a consequence of time-varying excitation. State occupancy is manipulated by varying the band bending associated with a Schottky barrier or pn junction. The spectral and energetic distribution of states within the space-charge region is probed by varying the frequency, temperature and bias voltage, as seen in [START_REF] Hegedus | Characterization of a-Si:H and a-SiGe:H pi-n and Schottky Junctions by Admittance Circuit modelling[END_REF].

A possible application of these type of photodiodes is in short range optical communications using the multilayered a-SiC:H heterostructure as a wavelength division demultiplexing device [START_REF] Vieira | SiC multilayer photonic structures with self optical bias amplification[END_REF]. Nowadays optical communications demand the transmission of a huge amount of information. To increase the transmission capacity and to allow bidirectional communication over one strand fibre, wavelength division multiplexing (WDM) is used. This technique consists on combining multiple wavelength optical signals on a single optical fibre. To perform such task the WDM system uses a multiplexer at the transmitter to mix the signals and a demultiplexer at the receiver to split them apart [4,5]. The pin-pin diode used in this analysis was designed to work as an optical filter, with selective wavelength sensitivity. Both pin structures of this device were optimized for selective collection of photo generated carriers. Band gap engineering was considered to adjust the photo generation and recombination rates of the intrinsic region of each diode, taking into account wavelength absorption and carrier collection in the visible range.

Photo capacitive effects influence the device frequency response under transient conditions. As a WDM device is intended to be used to increase the bandwidth of the optical communication system the transmission rate must be kept as high as possible. Capacitive effects result in signal degradation and induce errors at the reception end [START_REF] Hegedus | Characterization of a-Si:H and a-SiGe:H pi-n and Schottky Junctions by Admittance Circuit modelling[END_REF], which is an important limitative aspect for the device performance. In order to minimize this effect it is important to understand the mechanisms responsible for the device photo capacitance.

Capacitance values are evaluated considering the impedance of the device in study.

Contribution to Internet of Things

The Internet of Things is a "vision" that is being created these days. It concerns the idea of expanding communication among things. This will bring the ability of objects performance to change due to what they contact through the internet. In the particular case of optical communications in the visible range, recent research and new sensor devices, for instance WDM (wavelength-division multiplexing), will contribute to innovation and new resources and capabilities. And that will contribute to the future development of the Internet of Things.

Research Contribution and Innovation

Measurements of capacitance were made with different photodiodes, under different illumination conditions and changing the applied voltage. Three different devices were used in these experiments: a commercial crystalline silicon pin photodiode (Vishay Semiconductors BVP10), an amorphous silicon pin photodiode and a pin-pin structure based on amorphous silicon and amorphous silicon carbide. The a-Si:H devices were produced by PECVD (Plasma Enhanced Chemical Vapour Deposition) The pin-pin device structure consists of a p-i'(a-SiC:H)-n/p-i(a-Si:H)-n heterostructure with low conductivity doped layers. Our goal was to point out the different behaviours of the devices and to relate these differences to the structure composition and to the material used. The light incident onto the diodes was obtained by a white lamp and a colour filter covering the light source. The measurements were performed using a pulsed monochromatic light with a frequency ranging from 10 Hz up to 1 kHz. From the WDM application point of view, these frequencies are indeed very low. We have chosen this measurement condition for permitting a clear analysis about the influence of the materials and interfaces properties on the device capacitance.

Discussion of Results and Critical View

In Fig. 1 it is reported the CV measurement of an a-Si:H pin photodiode, under dark conditions and using illumination with different wavelengths (red: 626 nm, green: 510 nm, and blue: 430 nm). The device is biased in the range -1V up to +1V.

Results show that in dark the capacitance does not change with the applied voltage, either under reverse and forward bias. Under illumination, a different behaviour is observed, as the capacitance changes with the voltage and in the analysed bias range it takes higher values. The highest capacitances are observed under red illumination; the medium capacitance values are observed under green light; and at last the smallest ones under blue light.

Under reverse bias (-1V) the capacitance under blue and green light match the dark capacitance, while the capacitance under red illumination is slightly higher. However, as the voltage increases the capacitance under illumination increases at different rates. This is in agreement with the classical theory of the junction capacitance. In Fig. 2 we can see the CV measurement referred to a pin-pin photodiode. The value of the light induced capacitance depends slightly on the applied bias, on the light intensity. And it varies with light wavelength greatly; the red colour corresponds to the higher values of capacitance. Comparing the pin diode with the pin-pin device we can see that both capacitances vary with applied bias and with light wavelength, being the capacitance in the pin-pin structure much higher than the pin one.

Considering imposed red light, the pin diode has a major increase in the value of capacitance, as the value of bias voltage gets higher; in the case of the pin-pin diode, the capacitance value is very high, but remains constant, regardless the bias voltage value. In the case of green and blue light, the capacitance changes are very similar.

In Fig. 3 it is reported the comparison of the transient response of the crystalline silicon diode (c-Si diode) under a pulsed red light, used as reference term, and the hydrogenated amorphous silicon diode (a-Si:H pin diode) under red and blue pulsed light, with different frequencies: 10 Hz, 100Hz and 1 kHz, namely a), b) and c).

In Fig. 3 -a), b) and c), the inferior trace of each plot shows the behaviour of the crystalline diode, under red pulsed light. We can see a similar pulsed photocurrent as a function of time, as the diode response. In the classical theory, capacitive effects of the device were not expected.

In Fig. 3 -a), b) and c), the central image in each one show the response of an amorphous pin diode under red pulsed light. A transient photocurrent, as a function of the time, is the response of this experiment. A peak value of current can be recognized; for small values of frequency, 100 Hz and specially 10 Hz, the amorphous photodiode photo current has a peak of intensity, in the beginning of each time period. It repeats the behaviour of a RC circuit in transient regime.

Using a different colour of pulsed light (violet), as we can see in the superior images of each figure 3 -a), b) and c), the amorphous silicon diode response is similar to the applied pulsed light. In Fig. 3 -c) the frequency used is 1 kHz. The crystalline diode keeps its response stable and the amorphous photodiodes have different responses, similar to the previous plots, but on a smaller scale.

In Fig. 4 we present the time response of a double amorphous silicon device, a pin-pin diode, for different wavelengths of pulsed light. The red, green and blue imposed lights are transformed in transient photocurrents, with a maximum and a minimum values, during a time period, whose rising and decaying times are a little different. The peak values are obtained under the red colour, and the smaller ones under the blue colour, recalling the behaviour of a RC circuit in transient regime. The charging and discharging times to and from the offset photocurrent value are different. This may lead to conclude that the two processes are made on different paths with different resistance or/and capacitance characteristics.

Observed values for the capacitance of the pin-pin are higher than for the pin device, which may be related to the presence of two double pin structures stacked together. From this configuration results an additional internal pn junction causing a separation of the trapped charge within the two active layers of the device. Also a higher series resistance is expected in the double diode configuration.

Conclusions and Further Work

Diode photo capacitance varies with different parameters. Considering the crystalline diode, the photocurrent follows the imposed pulsed light showing no capacitive effect; it is independent of its frequency and colour.

In the case of an amorphous pin diode the photocurrent presents a transient behaviour that is consistent with charge and discharge of a capacitive circuit. Its highest value is present when the frequency value is low, and when the pulsed light is red.

Considering the double pin-pin diode, the time response has two peaks in a time period, one positive and the other, smaller, negative. They expose the capacitive effects in this device; capacitance values also depend on light wavelength, the highest values for the red colour. These results permitted to relate the photodiode behaviour with the device characteristics of capacitance and resistance.

In future work it will be useful to make measurements of different devices with varying thickness of the absorbing layers. Also the influence of the defect densities in the intrinsic layer should be studied. These results should be confirmed by electronic simulation based on an equivalent model of the device that is being tested. And the study will be extended to higher values of frequency, up to 1 GHz.

Fig. 1 .

 1 Fig. 1. -Capacitance-voltage characteristics of an a-Si:H photodiode, under dark condition and under illumination with different wavelengths (red, green, and blue).

Fig. 2 .

 2 Fig. 2. -Capacitance-voltage characteristics of a pin-pin photodiode, under dark condition and under illumination with different wavelengths (red, green, and blue).

Fig. 3 .

 3 Fig. 3. -Comparison of the transient response of the c-Si diode under a pulsed red light and the a-SI:H pin diode under red and blue pulsed light: a) 10 Hz, b) 100Hz and c) 1 kHz.

Fig. 4 .

 4 -Time response in a pin-pin diode, showing the photo current for different wavelengths (red, green and blue colours).


Vitor Silva

Manuela A Vieira

Paula Louro

Manuel Barata

Optoelectronic Digital Capture Device Based on Si/C Multilayer Heterostructures

Keywords: SiC hetrostructures, Optical sensors, Optical active filters, Numerical and electrical simulations, Optoelectronic model

Combined tunable WDM converters based on SiC multilayer photonic active filters are analyzed. The operation combines the properties of active long-pass and short-pass wavelength filter sections into a capacitive active band-pass filter. The sensor element is a multilayered heterostructure produced by PE-CVD. The configuration includes two stacked SiC p-i-n structures sandwiched between two transparent contacts. Transfer function characteristics are studied both theoretically and experimentally. Results show that optical bias activated photonic device combines the demultiplexing operation with the simultaneous photodetection and self amplification of an optical signal acting the device as an integrated photonic filter in the visible range. Depending on the wavelength of the external background and irradiation side, the device acts either as a short-or a long-pass band filter or as a bandstop filter. The output waveform presents a nonlinear amplitude-dependent response to the wavelengths of the input channels. A numerical simulation and a two building-blocks active circuit is presented and gives insight into the physics of the device.

Introduction

An optoelectronic device converts light photons to electrons that mimic the light signal in such way that data transmitted by the light beam can be received and processed further with electrical circuits. These devices have one surface over which the optical signal shines. On our device, a multilayered Si/C heterostructure [START_REF] Vieira | Three Transducers Embedded into One Single SiC Photodetector: LSP Direct Image Sensor, Optical Amplifier and Demux Device[END_REF][START_REF] Louro | Use of a SiC:H Photodiodes in Optical Communications Applications[END_REF], light can shine on the two surfaces, namely back and front. This device acts as an optical filter when other fixed wavelengths superimpose the incident light data signal on the surface it shines with. By selecting a wavelength on either the red or blue part of the spectrum the device can be tuned as a filter and used as a wavelength division multiplexing-demultiplexing technique, WDM [START_REF] Iguchi | Novel rear-illuminated 1.55 µm-photodiode with high wavelength selectivity designed for bi-directional optical transceiver[END_REF]. This device has been characterized with a model with its experimental verification. This paper continues this work by analyzing the digital signals which modulate the incident light beam and performs logic functions [START_REF] Bas | Fiber Optics Handbook, Fiber, Devices and Systems for Optical Communication[END_REF][START_REF] Randel | Advanced modulation techniques for polymer optical fiber transmission[END_REF].

Internet of Things

Internet is a well known concept that reflects the whole connection of computers forming a single international network. Behind computers are users and their communication skills looked upon this network as an excellent communication media for human interaction and thus the social network bruited. People while communicating with one another usually share ideas, emotions and objects as photographs, text, sound, video, and much more things could be handled. The Internet of Things is a growing concept since Kevin Ashton brought it forward in 1999 [START_REF] Ashton | That 'Internet of Things' Thing[END_REF]. Objects are easily identified when an RFID tag is attached to them, and its whereabouts can be known. But objects don't actually communicate. Were this possible and an Internet of physical objects can emerge. People and objects interacting; whoever leaves its home keys inside the house would not be locked outside but instead a warning signal from the keys would call to ones attention. The umbrella would also call to its attention if the weather report expects rain. An occasional party could be foreseen in the agenda and tagged clothes would shine a few leds indicating possible matches inside the cupboard and eventually for the couple, even if they were in different homes. The device we propose would certainly be integrated in objects of design which interact with people via colored visible light. Spectral response measurements without and under different optical bias and frequencies were performed in order to test the devices sensitivity.

Device Optimization and Characterization

The device operates within the visible range using as input color channels (data) the wave square modulated light (external regulation of frequency and intensity) supplied by a red (624 nm; 51 µW/cm 2 ), a green (526 nm; 73 µW/cm 2 ) and a blue (470 nm; 115 µW/cm 2 ) LED. Additionally, steady state violet (400 nm, 2800 µW/cm 2 ), red (624 nm, 652 µW/cm 2 ), green (526 nm, 580 µW/cm 2 ) and blue (470 nm, 680 µW/cm 2 ) illumination (optical bias) was superimposed from the front (pin 1 ) and back (pin 2 ) sides, in LEDs driven at different current values. The spectral sensitivity was analyzed by applying different wavelengths optical bias from the front and back sides of the device (see Fig. 1). Under front irradiation, in Fig. 2 Results show that the spectral response depends strongly on the bias control wavelength and frequency. As the bias control wavelength increases the spectral sensitivity shifts to the low wavelength spectral regions and decreases with the frequency, suggesting capacitive effect across the device.

In Fig. 3 it is displayed the spectral photocurrent at 3500 Hz, under red, green, blue and violet background irradiations (color symbols) and without it (black symbols) applied from the front (a) and back (b) diodes. For comparison the spectral photocurrent (right axis) for the front, p-i'-n and the back, p-i-n, photodiodes (dash lines) are superimposed. Results show that the spectral sensitivity, under steady state irradiation, depends on its wavelength and on the impinging side.

λ R =624nm λ G =526nm λ B =470nm λ V =400nm
λ R =624nm λ G =526nm λ B =470nm λ V =400nm

Back optical bias

Under front irradiation, the back diode photoresponse is tuned and the sensitivity strongly increases for wavelengths higher than 500 nm when compared with its value without optical bias. Here, as the background wavelength decreases the spectral response increases.

Under back irradiation the front diode photoresponse is selected. The sensitivity strongly increases in the short wavelengths range and collapse in the long wavelength region.

Light Filtering Effects

In Fig. 4, at 3500 Hz, it is displayed the ratio between the photocurrent under different optical bias and without it (gain) under front (symbols) and back (lines) irradiations: red (α R ), green (α G ), blue (α B ) and violet (α V ). Under back irradiation (lines) the gain does not depend on the wavelength of the background. The spectral sensitivity in the low wavelength range is enhanced and the device acts, always, as a short-pass filter.

Under front irradiation (symbols) the filter properties of the device depend on the background wavelength. Under red irradiation (Fig. 2e) the gain is high in the short wavelengths range acting the device as a short-pass filter. Under violet (Fig. 2b) and blue (Fig. 2c) irradiations the transfer function presents an enhanced gain in the long wavelength range acting as a long-pass filter. Under front green background (Fig. 2d) the device, for frequencies higher than 2000 Hz, is a band-rejection active filter that works to screen out wavelengths that are within the medium range (475nm-550nm), giving easy passage at all wavelengths below and above.

Results confirm that under controlled wavelength backgrounds it is possible to fine-tune the spectral sensitivity of the device. Its sensitivity is strongly enhanced (α>1) in a specific wavelength range and quenched (α<1) in the others, tuning or suppressing a specific band. The sensor is a wavelength current-controlled device [START_REF] Vieira | Double Pin Photodiodes with Two Optical Gate Connections for Light Triggering: A Capacitive Two-phototransistor Model[END_REF] that makes use of changes in the wavelength and impinging side of the optical bias to control the power delivered to the load. Self optical bias amplification or quenching under uniform irradiation is achieved. By using background lights with complementary light penetration depths across the pi´n/pin device and changing the irradiation side, it is possible to control the spectral response and to filter a specific wavelength band.

Research Question and Hypothesis

How many channels in the visible light range can be transmitted by WDM with a a-Si:H and a-SiC:H sensor selecting the desired channel with visible light?

Light shining on one of the surfaces, the selector, can be used to distinguish between two signals with different wavelengths. For n different selecting frequencies there would be 2 n different signals to be identified. Value of n is then limited by the bandwidth of each channel. By changing the wavelength of the selector the output changes due to the non linear gain of the device which will reduce the maximum number of distinguishable channels. This limitation can be overcome and the number of identifiable channels may increase if the output of the selecting wavelengths are combined with an algorithm and signal modulation. Experimental results are presented in this paper with four different visible leds which are commonly available in the market. These results were obtained with a maximum bit rate of 6000bps. Although this is a very low throughput we are not at this stage worried with this aspect, but to study the recovery of several input signals with only one sensor. In this paper we present the gains of each of the four channels. This represents the influence of each of the selectors upon each input signal wavelength.

Photonic Active Filters

To analyze the device under information-modulated wave and uniform irradiation, three monochromatic pulsed lights separately (red, green and blue input channels) or their combination (MUX signal) illuminated the device. Steady state red (652µWcm - 2 ), green (515µWcm -2 ), blue (680µWcm -2 ) and violet (2800µWcm -2 ) optical bias was superimposed separately from the front (pin 1 ) and the back (pin 2 ) sides and the photocurrent measured at -8 V.

In Table 1 the gains (α) of the individual channels, defined as the ratio between photocurrents under irradiation (ON state) and without it (OFF state), for the red, green, and blue input channels are presented. Here, the superscripts are related to the background wavelength (R, G, B, V) and the subscripts (Rpin 1,2 ,G pin 1 , 2 B pin 1 , 2 ) to the channel color and irradiation side.

Results show that, even under transient conditions, the effect of the background wavelength and impinging side presents the same nonlinear dependence as in Fig. 4. The morphology of filter results from the interaction of the electric field under applied optical bias (red, green, blue, violet) and the transient electric field induced by the input channels (red, green, blue and violet). Under back irradiation the small absorption depth of the violet photons across the back diode quenches the electric field there and so, the red collection (Red ON) almost disappears (α V R,pin2 <<1). Blue channel is absorbed across the front diode where the electric field was enhanced resulting in an increase collection of the blue channel (α V B,pin2 >1). Since the green channel is absorbed across front and back diodes its collection is balanced by the increase collection in the front diode and its reduction at the back (α V G,pin2 ~1). The front violet background is absorbed at the surface of the front diode, increasing the electric field at the back diode, where the red and part of the green channels generate optical carriers. So, the collection is strongly enhanced (α V R,pin1 >>1, α V G,pin1 >1) while the blue collection stays near its dark value (α V B,pin1 ~1). Polychromatic combinations of the same red, green, blue and violet input channels whose gain is presented in Table I was used to generate a multiplexed (MUX) signal.

In Fig. 5 the filtered signals under front (pin 1 ) and back (pin 2 ) violet light control are displayed. The signals were normalized to the maximum intensity under violet front irradiations to suppress the dependence on sensor and LEDs positioning. The bit sequences used to transmit the information are shown at the top of the figures. Table 1. Gains (α R, G,B,V R,G.B,pin1,2 ) at the input red, green, and blue channels wavelengths. Different gains for the RGB channels were obtained (Table 1). Due to this wavelength non-linearity under front violet background, the encoded multiplexed signal presents as many levels as the possible RGB combinations, in a maximum of 2 3 (eight-level encode).Those levels can be grouped into two main classes due to the high amplification of the red channel (α V R,pin1 >>1). The upper levels are ascribed to the presence of the red channel and the lower to its absence allowing the red channel decoder. Since under front irradiation the green channel is amplified (α V G,pin1 >1), the highest levels, in both classes, are ascribed to the presence of the green channel and the lower ones to its lack (long-pass filter). Under back irradiation the red channel is suppressed (α V R,pin2 <<1), the blue enhanced (α V B,pin2 >1) and the green reduced(α V G,pin2 <1), so the encoded multiplexed signal presents a maximum of four separate levels (2 2 ). The highest levels correspond to the presence of the blue channel ON with or without the green ON respectively, and the other to its absence. The blue channel is then decoded using this simple algorithm (short-pass filter).

Channels

α R α G α B α V α R,pin1 0,
Results show that by means of violet optical bias control, the MUX photonic function may be modified, giving reconfiguration. This new method, based on wavelength background processing techniques enables the optical routing.

Conclusions

Integrated photonic filters based on SiC multilayer devices are analyzed and its transfer functions presented showing that the light-activated pi'n/pin a-SiC:H devices combine the demultiplexing operation with the simultaneous photodetection and selfamplification of an optical signal. Reconfiguration is provided by background wavelength control. Results show that the background wavelength and irradiation side control the output signal. By using background lights with complementary light penetration depths and changing the irradiation side, it is possible to change the spectral response and to filter a specific wavelength band acting either as a short-or a long-pass band filter or as a band-stop filter.

Fig. 1 .

 1 Fig. 1. Device configuration and operation. The sensor element is a multilayered heterostructure based on a-Si:H and a-SiC:H. The configuration shown in Fig. 1 includes two stacked p-i-n structures sandwiched between two transparent contacts. The thicknesses and optical gap of the front í'-(200nm; 2.1 eV) and back i-(1000nm; 1.8eV) layers are optimized for light absorption in the blue and red ranges, respectively.Spectral response measurements without and under different optical bias and frequencies were performed in order to test the devices sensitivity.The device operates within the visible range using as input color channels (data) the wave square modulated light (external regulation of frequency and intensity) supplied by a red (624 nm; 51 µW/cm 2 ), a green (526 nm; 73 µW/cm 2 ) and a blue (470 nm; 115 µW/cm 2 ) LED. Additionally, steady state violet (400 nm, 2800 µW/cm 2 ), red (624 nm, 652 µW/cm 2 ), green (526 nm, 580 µW/cm 2 ) and blue (470

Fig. 2 .

 2 Photo curre nt(µA)

  it is displayed the spectral photocurrent for different frequencies without (a) (b), blue (c), green (d) and red (e) light bias control at -8V applied voltage and different frequencies (250 Hz-3500Hz).

  Photocurrent (µA)V=-8V 3500 Hz

Fig. 3 .

 3 Photocurrent (µA)V=-8V 3500 Hz

Fig. 4 .

 4 Fig.4. Gain under front (symbols) and back (lines) irradiations: red (α R ), green(α G ), blue (α B ), violet (α V ).

Fig. 5 .

 5 Fig. 5. Filtered output signals under front (pin 1 ; lines) and back (pin 2 ; symbols) violet irradiation. a) Red, Green and Blue channels. b) Red, Green, Blue and violet channels. On the top of the figures the optical signals used to transmit the information guide the eyes.

Acknowledgements. This work was supported by PTDC/EEA-ELC/111854/2009, PTDC/EEA-ELC/115577/2009 and PTDC/EEA-ELC/120539/2010.


Jelena Radic
email: jelenar_@uns.ac.rs

Alena Djugova
email: alenad@uns.ac.rs

Laszlo Nagy

Mirjana Videnovic Misic

Ljiljana Zivanov

Comparison of Feedback Influence on Ring Oscillator Performance for IR-UWB Pulse Generator in 0.13µm and 0.18 µm CMOS Technologies

Keywords: CMOS process, impulse radio ultra wideband, PMOS transistor feedback, pulse generator, resistive feedback, ring oscillator

A CMOS three-stage ring oscillator is examined in UMC 0.13 µm and 0.18 µm technologies. The influence of PMOS transistor and resistor, as inverter feedbacks, on the ring oscillator frequency and the peak-to-peak amplitude is investigated in both technologies. Furthermore, as the ring oscillator usually drives a buffer in pulse generator/transmitter chain, dependence of its Figures of Merit on the buffer feedback is presented in the paper. Simulation results showed that the ring oscillator frequency is strongly dependent on the inverter feedback. The presented techniques can be used to increase (resistive feedback) and control (PMOS transistor feedback) the ring oscillator frequency. As the ring oscillator is a part of an IR-UWB (Impulse Radio Ultra Wide Band) pulse generator, its oscillating frequency determines the spectrum central frequency and has significant effect on spectrum fitting within UWB FCC mask.

Introduction

Since the Federal Communications Commission (FCC) allocated the 3.1 -10. [START_REF] Siwiak | Ultra-Wideband Radio Technology[END_REF] GHz unlicensed spectrum for commercial ultra wideband (UWB) application in February 2002[1], several technologies have been developed to satisfy the communication market requirements such as Multi-Band Orthogonal Frequency Division Multiplexing (MB-OFDM), Direct Sequence (DS)and Impulse Radio (IR) [2 -5]. As a carrier-less approach which makes use of ultra-short duration pulses (pulse duration is less than 1 ns) that can yield ultra-wide bandwidth signals, IR-UWB technology stands out from the rest by providing very simple (without mixer and power amplifier), low-cost and low-power UWB transmitter realizations [START_REF] Siwiak | Ultra-Wideband Radio Technology[END_REF]. Besides wide bandwidth, it offers a great resistance to multipath fading that usually plagues for narrow-band systems, and allows multiple access and robustness to interference [START_REF] Fernandes | Recent Advances in IR-UWB Transceivers: An Overview[END_REF]. Another advantage of the IR-UWB radios is high precision ranging [START_REF] Lee | Ranging in a dense multipath environment using an uwb radio link[END_REF], with a potential for centimeter accuracy in indoor environments. Consequently, the approach appears to be good candidate for very high data rate short-range communications, low data rate communications related to localization or/and positioning systems, sensor networks, biomedical and many other cutting edge Internet of Things applications.

A pulse generator plays the core role in an UWB system as it produces pulse train which spectrum has to fully comply with and efficiently use the FCC mask. Therefore, it is extremely challenging to design UWB transmitter that satisfies such demanding requirements while achieving wide bandwidth, low-power, low cost and simple architecture. Besides, it is very desirable to provide pulse generator spectrum tuning to enable compensation due to process, voltage and temperature (PVT) variations, and antenna and communication channel performance changes.

As an essential part of analogue-digital pulse generator, a ring oscillator is investigated in this work. In addition to the standard switched three-stage ring oscillator, topologies with resistors and PMOS transistors, introduced as feedbacks in the ring oscillator inverters, are examined in 0.13 µm and 0.18 µm UMC CMOS technologies. A novel approach that uses the resistive feedback in the buffer stage to increase ring oscillator frequency without degrading its peak-to-peak amplitude is proposed. Influence of negative feedback on the ring oscillator performance and comparison of results obtained using considered processes are presented.

Relation to Internet of Things

After the computer in the 1940s and the Internet in the 1970s, the Internet of Things (IoT) represents the third revolution in information and communication technology (ICT). With main vision of "Everything connects", its development depends on dynamic technology evolutions in a set of multidisciplinary and interdisciplinary fields, ranging from the material and devices, sensor technology, to electronic system design and wireless communications. To fulfill the IoT demands, future microdevices must provide a variety of advanced futures such as sensing, processing, communication, positioning, and capability to be integrated into everyday objects. To accomplish this, reliable-operating and maintenance-free wireless networks with lowpower and low-cost radio transceivers are essential. Many advantages of the IR-UWB systems provide them leading role in wireless communications and sensor networks, and thus precedence in IoT applications. Wideband signals offer robust communications and high-precision positioning; duty-cycled operations allow link scalability and energy-efficient solutions; and baseband-like architecture facilitates extremely simple and low-power transmitters that can be integrated in various objects.

Thought it is not required, ability of UWB transmitter spectrum tuning represents very desirable quality that allows compensation for PVT variations and communication channel performance changes, and furthermore interference avoidance. Methods proposed in this work provide control and increase in the ring oscillator frequency. Since this parameter determines the center frequency of the ring oscillator-based pulse generator output spectrum, such ideas can be used to enable the IR-UWB transmitter spectrum adjustment.

Standard Three-stage Ring Oscillator Design

It is already mentioned that the pulse generator is one of the most important parts of the IR-UWB system. The oscillator-based transmitter that contains a switched ring oscillator represents the most commonly used topology, presented in Fig. 1 [9]. The glitch generator turns on/off the ring oscillator approximately defining duration of its oscillation and so the width of the transmitter output pulse. The pulse generator spectrum center frequency is determined by the ring oscillator frequency. The buffer stage isolates the switched oscillator from the output filter, and improves the UWB transmitter current driving capability. The pulse shaping, usually the band-pass filter additionally accommodates spectrum fitting within the FCC mask. 4 Ring Oscillator in 0.13 µm and 0.18 µm CMOS Technologies

The proposed design has been simulated in mixed mode/RF 0.13 µm and 0.18 µm UMCCMOS technologies using Cadence's Spectre RF Simulator. Supply voltages V dd of these technologies are 1.2 V and 1.8 V, respectively.

The ring oscillator working frequency depends directly on transistors sizes. The period of the oscillation T rises proportionally with increase in transistors sizes, while the oscillating frequency decreases (f 0 =1/T). For the same NMOS and PMOS ring transistors size (channel width/length W/L=3.6 µm/0.12 µm) in 0.13 µm CMOS technology, the oscillation frequency of 7.65 GHz has been obtained. However, for the smallest NMOS transistors (W/L=25 µm/0.18 µm) and approximately two times larger PMOS transistors (W/L=45 µm/0.18 µm) in 0.18 µm CMOS technology, the f 0 parameter of 3.77 GHz has been achieved. It should be noted that in inverter topology PMOS transistor is usually equal or two times larger than NMOS transistor to compensate the difference in PMOS and NMOS carrier mobility and set V TH closer to V dd /2. Although former technology allows considerably higher ring oscillator frequency, 0.18 µm technology is much more cost effective [10], and attractive from the IC fabrication costs point of view. Nevertheless, obtained values are not high enough to utilize the UWB higher-band (6 -10 GHz) more effectively (the center frequency of at least 8 GHz is required).

Influence of the Inverter Resistive Feedback

To increase the oscillation frequency at a given DC current, the resistive feedback is used in each inverter stage, as proposed in [START_REF] Sim | A CMOS UWB pulse generator for 6-10 GHz applications[END_REF]. Resistors were connected between nodes A-A', B-B', and C-C', as shown in Fig. 3. Resistors R influence on the performance (the frequency f 0 and peak-to-peak amplitude V pp ) of the ring oscillator 1. It can be noted that with decrease in the resistor value the oscillation frequency increases, but the peak-to-peak amplitude V pp of the ring oscillator output signal is reduced, as expected. This behavior is caused by the reduction of the inverter closed loop gain with decrease in the feedback resistor value. Furthermore, the V pp parameter and the time for reaching the peak signal values are reduced resulting in f 0 parameter increase. Moreover, it can be seen that the oscillation frequency was considerably increased in both technologies, while higherf 0 valuesare obtained in the scaled UMC technology, as expected. But, the V pp parameter values achieved for the lowest resistors values (4 kΩ in case of 0.13 µm, and 0.5 kΩ in 0.18 µm CMOS process)are not large enough to drive properly the subsequent stage (usually a buffer) in the pulse generator chain. This is the main reason why higher feedback resistor values are used in 0.13 µm CMOS architecture (otherwise V pp values even lower than 440 mV would be obtained). On the other hand, if higher resistor values are used in0.18 µm technology (the same as in 0.13 µm) the ring oscillator frequency would not be increased comparing to the initial value of 3.77 GHz. Therefore, it should be emphasized that the resistor value needs to be selected considering the trade-off between the oscillation frequency and the required V pp value.

Although the f 0 parameter can be significantly increased by proposed technique, this method does not allow the ring oscillator frequency to be tuned. It is already mentioned that even if it is not required in the IR-UWB technology, the ability of the frequency adjustment is very desirable advantage.

Influence of Inverter PMOS Transistor Feedback

To enable electrical tuning, the topology with PMOS transistors as inverter feedbacks has been proposed, Fig. 4.Influence of the PMOS gate control voltage V ctrl on the performance of the ring oscillator architectures in 0.13 µm and 0.18 µm technologies is summarized in Table 2.

In addition to higher f 0 values, the 0.13 µm technology offers noticeably wider tuning range (3.15 GHz) compared to frequency range (1.46 GHz) available in 0.18 µm CMOS process. For V ctrl parameter value equal to zero, the same situation as in the previous method can be noticed; the achieved V pp values are below acceptable

Buffer Feedback Influence

The method of increasing the ring oscillator frequency by using the inverter resistive feedback is already known in literature, [START_REF] Sim | A CMOS UWB pulse generator for 6-10 GHz applications[END_REF]. To the best of our knowledge, the influence of the buffer feedback on the ring performance is still not examined. The two-stage buffer topologies with resistive and PMOS transistor feedback are proposed in Fig. 5. The first stage provides control/increase in the ring oscillator frequency, while the second buffer stage prevents this effect to change the output filter shaping characteristics. Dependences of performance of the ring oscillator designs in 0.13 µm and 0.18 µm UMC technologies on the buffer resistive and PMOS transistor Varying R from 10 kΩ to 2.32 kΩ in 0.13 µm technology, the oscillation frequency is increased from 8.10 GHz to 9.05 GHz, followed with decrease in V pp parameter from 0.85 V to 0.72 V. Reducing R from 4.0 kΩ to 0.25 kΩ in 0.18 µm CMOS ring oscillator design, the f 0 parameter rises from 3.82 feedback resistor values lower than in method presented in section 4.1 because obtained V pp values are higher than V dd /2. It can be noticed that comparing to the mentioned approach the proposed technique allows smaller increase in the ring oscillator frequency. However, it should take into account that this method uses only one resistor in contrast to three additional resistors utilized in the inverter feedback architecture, and at the same time provides higher V pp parameter values for the same ring oscillator frequency. The three-stage ring oscillator period T is determined by the propagation time of a signal transition through the complete oscillator chain (T=6×t p , where t p is the gate propagation delay). An inverter signal propagation time is largely determined by the strength of the driving gate, and the load presented by the output node itself, which sums the contributions of the gates and the wiring parasitic. Varying the buffer feedback resistor changes its feedback impedance, which further modifies the buffer input impedance (Miller's theorem) leading to the change in the ring oscillator load, propagation time and the ring oscillator period/frequency. Approach presented in Fig. 5b shows similar influence on the ring oscillator FOM. Reducing V ctrl parameter from V dd to zero the oscillation frequency can be tuned in the frequency ranges narrower than in technique presented in section 4.2 (especially in 0.18 µm CMOS process), but with higher obtainable V pp parameter values. Moreover, the method requires only one resistor. Additionally, it can be noticed that the frequency range available in 0.13 µm CMOS technology is extensively wider than in the other considered process.

Discussion and conclusion

Dependences of the three-stage ring oscillator performance on resistors and PMOS transistors, introduced as feedbacks in the oscillator inverters and buffer, are investigated in 0.13 µm and 0.18 µm UMC CMOS technologies. Simulation results in both technologies confirm strong dependency of the ring oscillator frequency and the peak-to-peak amplitude on the inverters and buffer feedbacks. Considering the presented methods, higher f 0 parameter values and wider available frequency range can be obtained with topologies that use feedbacks in the ring oscillator inverters. However, these architectures require additional three resistors/PMOS transistors that occupy more area and thus increase chip cost. The topology with buffer feedback would be the best choice in case of the medium operating frequency range applications as it uses only one additional component and gives higher V pp parameter values. Research presented here shows that the 0.13 µm process offers much more promising performance. Moreover, the layout designs dimensions are approximately two times smaller than in0.18 µm technology. Nevertheless, as the main drawback of the former technology is higher IC fabrication cost, the later process is chosen as optimal solution in this work. The ring oscillator-based pulse generator designs in 0.18 µm technology that combine proposed methods are presented in [START_REF] Radic | A Low Power 3.1-7.5 GHz Tunable Pulse Generator for Impulse Radio UWB[END_REF], [START_REF] Nagy | Ultra Low-Power Low-Complexity Tunable 3-10 GHz IR-UWB Pulse Generator[END_REF].

Fig. 1 .

 1 Fig. 1. An IR-UWB transmitter based on ring oscillator as a part of pulse generator. The switched ring oscillator topology is shown in Fig. 2. It consists of the standard three-stage ring oscillator (inverter stages M 1 -M 3 ) and oscillation-enabling switches (transistors M 4 and M 5 ). The ring oscillator is turned on by transistor M 4 at the on-off signal (produced by the glitch generator) rising edge. The inverter stages output voltage values are determined by the size ratio of the corresponding PMOS and NMOS transistors. During the on-off signal low level, the transistor M 4 is switched off, and no signal is generated at the oscillator output. Furthermore, transistor M 5 connects the M 1 transistor output (the M 2 transistor input) to V dd providing the oscillation start from the same initial state at the rising edge the on-off signal.

Fig. 2 .

 2 Fig. 2. The standard three-stage ring oscillator architecture.

Fig. 3 .

 3 Fig. 3. Architecture of the ring oscillator with the resistive feedbacks.

Fig. 4 .

 4 Fig. 4. Topology of the ring oscillator with PMOS transistor feedbacks.

Fig. 5 .

 5 Fig. 5. Two-stage buffer with: (a) Resistive feedback (b) PMOS transistor feedback.

Table 1 .

 1 Ring oscillator performance dependence on the inverter feedback resistor value.

	Tech.		0.13µm CMOS			0.18µm CMOS
	R (kΩ)	4	6	8	10	∞	0.5	1	2	4	∞
	f 0 (GHz) 9.80	9.10	8.60	8.20	7.65	5.60 4.47	4.09	3.92	3.77
	V pp (V)	0.44	0.62	0.69	0.73	0.88	0.60 1.18	1.38	1.48	1.57

Table 2 .

 2 Ring oscillator performance dependence on the gate control voltage of the PMOS transistor in the inverter feedback. This can be explained by the fact that low resistance of PMOS feedback transistor, operating in deep triode region, reduced significantly inverter closed loop gain. As the presented topology has the PMOS transistor feedback in each inverter stage, the effect is much more intense resulting in the unsuitably low V pp value. It should be pointed out that the maximum oscillation frequency and the available frequency range depend on the ring oscillator transistors sizes. Increase in the transistors sizes reduces the observed Figures of Merits (FOM).

	Tech.		0.13µm CMOS			0.18µm CMOS
	V ctrl (V)	0	0.3	0.6	0.9	1.2	0	0.3	0.6	1.2	1.8
	f 0 (GHz) 8.75	6.70	5.85	5.60	5.60	4.29 3.40	3.13	2.84	2.83
	V pp (V)	0.11	0.72	0.90	0.94	0.94	0.46 1.09	1.34	1.56	1.57
	limits.

Table 3 .

 3 Ring oscillator performance dependence on the buffer feedback resistor value.

*Minimal resistance value for the RNHR resistor model.

Table 4 .

 4 Influence of the buffer feedback transistor V ctrl parameter on the ring oscillator performance.

	Tech.		0.13µm CMOS			0.18µm CMOS
	V ctrl (V)	0	0.3	0.6	0.9	1.2	0	0.3	0.6	1.2	1.8
	f 0 (GHz) 8.90	8.25	7.70	7.45	7.4	3.94 3.74	3.63	3.55	3.55
	V pp (V)	0.71	0.82	0.87	0.87	0.88	1.17 1.27	1.37	1.47	1.47

  GHz to 4.27 GHz simultaneously with reduction in V pp amplitude from 1.53.85 V to 1.19 V. This technique uses

	Tech.		0.13µm CMOS			0.18µm CMOS
	R (kΩ)	2.32 *	4.0	6.0	8.0	10.0 0.25	0.5	1.0	2.0	4.0
	f 0 (GHz) 9.05	8.50	8.30	8.15	8.10	4.27 4.11	3.98	3.87	3.82
	V pp (V)	0.72	0.79	0.82	0.84	0.85	1.19 1.35	1.44	1.50	1.53

Acknowledgments. This work was supported in part by the Ministry of Education and Science, Republic of Serbia, on the project no. TR-32016, and by FP7 project no. 256615.