Abdelmoumin Allioua
email: abdelmoumin.allioua@lea.tu-darmstadt.de

Prof. Dr.-Ing Gerd Griepentrog
email: gerd.griepentrog@lea.tu-darmstadt.de

Martin Vögel
email: martin.voegel@liebherr.com

Julian Eitler
email: julian.eitler@liebherr.com

Dr.-Ing Nejat Mahdavi
email: nejat.mahdavi@liebherr.com

Design of PCB-based planar coil inductive coupler

Keywords: Planar coil, Inductive coupling, Coupling efficiency, Transmission gain, Parasitic components, Channel transfer function (TF), PCB design, PLC, QPSK Modems, Latency, BER, HDX (half-duplex), RTCA

The coupling method used to superimpose and separate the data signal to and from the powerline plays a crucial role concerning the channel frequency response, the coupling efficiency, and also preserving the signal integrity against the peaks and notches caused by reflections, which will cancel certain carrier frequencies at the receiver. Furthermore, the coupling unit should be bi-directional to allow the signals coupling regardless the direction of data flow, likewise it needs to accomplish galvanic isolation together with transient protection, and offers more degrees of freedom to impedance-matching, as well as allocating less space and achieve high bandwidth for broadband coupling. This paper discusses an approach to pursue toward designing a printed-circuit board (PCB)-based planar coil inductive coupler with wide bandwidth for data transfer.

The designed coupler will eventually be used to implement powerline communication (PLC) adequately for airborne use, where a real-case context study was investigated, also PLC implementation and tests were conducted as well as discussed from latency, bit error rate (BER) and electromagnetic compatibility (EMC) compliance perspective with the RTCA DO-160G standard requirements.

I. INTRODUCTION

Replacing the aircraft's existing hydraulic and pneumatic systems via more flexible electrical ones, will notably reduce their operating and maintenance costs. However, this concept leads to an increase of the wiring length, complexity and space allocation, which affects the airborne weight and consequently impacts their polluting emissions.

A potential solution to reduce the total harness mass, is to use PLC technology in order to transmit data over the already The work presented in this paper has been financially supported by Liebherr-Aerospace Lindenberg GmbH, in the frame of the ATIKA research project, and funded from Bundesministeriums für Wirtschaft und Technologie under 20Y1708E.

existing power cables, with a special respect to their voltage level and noise margin, also with regards to the used coupling method which plays an important role on implementing PLC successfully.

Different coupling methods for broadband and narrowband PLC transceivers have been addressed and reviewed in [START_REF] Da | Coupling for Power Line Communication: A Survey[END_REF] and [START_REF] Bilal | Designs of broadband coupling circuits for powerline communication[END_REF].

In this work, we investigated several possible coupling solutions, such as low-cost capacitive coupling by inserting one or more capacitor in direct contact with the powerline for the parallel signal injection as shown in Fig. 1 (a), in addition to the inductive coupling using toroidal magnetic cores for serial signal injection as shown in Fig. 1 (b), which can offer low insertion and return loss along with covering wide frequency bandwidth. Nevertheless, such solutions need a high number of components to achieve symmetrical and reciprocal high bandwidth, also distortion and signal loss can be caused due to the magnetic materials non-linearity, which can be problematic from aeronautic safety and reliability standards point of view. However, a highly effective planar coil coupler with a wide bandwidth seems to be a promising solution to be implemented for airborne uses, and offers more flexibility in terms of geometry shapes, size and turns number. More advantages of the coreless planar inductors can be found in [START_REF] Tang | Coreless planar printed-circuitboard (PCB) transformers-a fundamental concept for signal and energy transfer[END_REF]- [START_REF] Tang | Optimal operation of coreless PCB transformer-isolated gate drive circuits with wide switching frequency range[END_REF].

The planar coupler efficiency represent a major factor in preserving the PLC signal integrity. This efficiency is highly dependable on the position between the transmitting (Tx) and the receiving (Rx) coil, and inversely proportional to the distance between them. Furthermore, an important limiting component of the planar coupler operating frequency bandwidth is the parasitic capacitance, which includes the intertrace capacitance, and the coupling capacitance between the Tx and Rx inductor. Thus, the right compromise need to be found between both coupling efficiency and the parasitic capacitances, with an optimal model for the designed coupler.

The design and prototyping of such a coupling approach for operating at high frequencies will be investigated in this work, while considering the coupling efficiency, along with the inductors parasitic effects. The PLC channel frequency response will be also evaluated, as well as the signal integrity, while being compliant with the DO-160G aeronautic standards.

The paper is structured as follows. In the next section, the standalone planar coil model is described. In section III, we present the inductive coupler optimal design. In section IV, we discuss the coupler implementation into the PLC channel, furthermore we present the obtained results with regard to the bitrate, the latency, as well as the BER and the EMCcompliance. Finally, section V concludes the paper.

II. STANDALONE SINGLE LAYER PCB INDUCTOR

In this part, the main physical and electrical key parameters to design a standalone planar coil are discussed, from the desired inductance value, together with the stray capacitances. The winding shape and turns number were also evoked, along with the coil outer and inner dimensions, as well as the spacing between traces, in addition to their width and thickness.

A. Inductance value

To reduce impedance mismatching and to decrease the reflection over the PLC channel, an adaptive stage was designed as detailed in [START_REF] Allioua | Aircraft DC networks characterization and adaptive stage design for PLC use[END_REF] and showed in Fig. 2. To avoid effecting the matching network effectiveness, the planar coil inductance value need to be considered carefully, where the coil reactance need to be greater than the matching network impedance value as shown in Eq. (1). Considering that the lowest operating frequency is 1 MHz, the inductance value need to be in the range of 8 to 13 µH at least, this inductance values are possible to achieve with small compact geometries.

B. Coil shape and turns number

In this section, based on Wheeler method [START_REF] Derkaoui | Modeling and simulation of an integrated octagonal planar transformer for RF systems[END_REF] [5] represented in Eq. ( 2), a trade-off between the coils inductance value L s and the number of turns N s was made for different shapes as illustrated in Fig. 3.

L s = k 1 µ 0 N 2 s d avg 1 + k 2 A m (2)
Where d avg is the average diameter, A m is the fill factor, µ 0 is the vacuum permeability, and the coefficients k 1 and k 2 depend on the different coil shapes as shown in TABLE I.

Regardless the coils shape, the used outer and inner diameters in this trade-off were 30 mm and 10 mm respectively. The square shape coil presents a higher inductance value compared to the circular and octagonal coil shapes. However, for prototyping simplicity and to avoid manufacturing constraints, the circular planar coil shape will be chosen to build the targeted prototype.

C. Planar coil electrical model

High-frequency inductors can be modeled as an impedance network [START_REF] Lee | Printed Spiral Winding Inductor With Wide Frequency Bandwidth[END_REF], with the lumped-element equivalent circuit shown in Fig. 4.

The self-resonance-frequency F srf of the inductor is calculated in Eq. (3) as follow. The inductor's self-capacitance C s is computed using Eq. ( 4), where it is divided into C pc the capacitance between two traces through air or coating isolator, and C ps which represent the capacitance through the substrate as shown in Fig. 5.

F srf = 1 2π √ C s • L s (3)
C s = C pc + C ps = (α • ε rc + β • ε rs ) • ε 0 t s • l t (4)
ε rc and ε rs represent the relative dielectric constants of the coating and the substrate materials respectively as detailed in [START_REF] Ashraf | Design and Optimization of Printed Circuit Board Inductors for Wireless Power Transfer System[END_REF], and shown in TABLE II.

l t represent the total length of the coil, s is the space between traces and t is the trace thickness. In case of using a ground plane to reduce the noise and interference, C p need to be considered as well among the parasitic capacitances, which represent the capacitance to the ground in a single layer PCB inductor, where the conductor are on one side, and the ground layer is on the other side, as detailed in Fig. 5 and calculated in Eq. [START_REF] Zolog | Characterization of spiral planar inductors built on printed circuit boards[END_REF].

C p = ε 0 ε rs • w • l t t rs ( 5
)
Where t rs is the substrate dielectric thickness, and w is the traces width.

The AC resistance of the coil R s which takes into consideration the skin effect as detailed in [START_REF] Zolog | Characterization of spiral planar inductors built on printed circuit boards[END_REF], is computed using Eq. ( 6) as follow.

R s = ρ cu • l t w • t eff = ρ cu • l t w • δ • (1 -e -t δ ) (6)
Where t eff is the effective thickness, δ is the skin depth computed using Eq. ( 7), and ρ cu is the annealed-copper electrical resistivity.

δ = ρ cu π • f • µ 0 • µ r (7)

D. Simulation model and measurement results

Based on the equivalent circuit analytical model shown previously in II. A, B and C, a standalone planar coil geometry parameters were identified at first as shown in TABLE III. To validate the analytical model, a lumped element π-model simulation was conducted, for the three different planar coil shapes shown in Fig. 6, using CST-Studio suite. The shown single layer prototype in Fig. 7 was built to verify the accuracy of the calculated characteristics as detailed in TABLE IV, and also to confirm the simulated results as shown in Fig. 8, where both the magnitude and phase of the coils were compared, we can notice that the results were within the same range. Compared to the long computing time of a 3D numerical coupling model, the analytical calculation can provide fast and acceptable accuracy results. Hence, the analytical model will be used next to identify empirically an optimal planar coupler design, which will be verified with experimental results.

III. DOUBLE LAYER PLANAR COIL INDUCTIVE COUPLER

In this section, a double layer planar coupler will be designed, where different geometry arrangements were tested, to identify an optimal geometry combination regarding the turns number, and the coils inner and outer diameters, also a compromise was determined between traces width and space along with the distance between the Tx and Rx coils.

A. Geometry parameters

With respect to the coupling efficiency and the desired operating frequency bandwidth, coil positions, turn numbers, and geometry combinations were investigated, along with the possibility of adding flexible ferrite sheets to enhance the coupling efficiency.

Based on the analytical model, the design parameters of the planar coupler presented in Fig. 9 were empirically optimized, and identified as represented in TABLE V, where 150 µm and 450 µm were chosen as traces width and spacing respectively, to keep C s as low as possible. To ensure a high coupling efficiency between the Tx and Rx coil, the substrate thickness h 2 between them was kept to 254 µm, however C k was slightly reduced, by skewing the Rx coil traces with regards to the Tx ones as shown in Fig. 10.

B. Electrical parameters

Based on the illustrated π-model in Fig. 4 for the single layer inductor, we propose the electrical model shown in Fig. 11, for the double layer inductive coupler.

The coils inductance L s -Rx/Tx , resistance R s -Rx/Tx , along with the stray capacitances C s -Rx/Tx and C p , were computed using the previous formulas in Eq. ( 2), ( 4), ( 5) and [START_REF] Ashraf | Design and Optimization of Printed Circuit Board Inductors for Wireless Power Transfer System[END_REF].

Furthermore, the capacitive coupling represented with C k was included in calculating the F srf by considering the distance between the two coils, and the overlapping surface area between them. The computed results are shown in The capacitive coupling, along with the open and short circuit inductances were measured using the vector network analyser (VNA) as shown in Fig. 12. Two tests were preformed, the first one by shorting the transmitting and receiving coil as shown in Fig. 12 (a) to identify C k as represented in Fig. 13 (a).

The second one was done to identify L short and L open , by shorting or not P 1 and P 2 , shown in Fig. 12 (b). The results are shown in Fig. 13 (b) and(c) respectively. Both tests, with and without ferrite sheets option were investigated.

C. Coupling efficiency and transmission gain

The coupling coefficient represents the magnetic strength between the Tx and Rx coils, which corresponds to the induced amount of power in the Rx coil based on the generated electromagnetic field in the Tx coil, and it can be computed as shown in Fig. 14 using Eq. ( 8) with the open and short circuit inductance measured previously [START_REF] Durate | Analysis of the Coupling Coefficient in Inductive Energy Transfer Systems[END_REF].

k = 1 - L short L open (8)
Fig. 14: Coupling coefficient k of the planar coupler.

The scattering parameter S21, which refers to the transmission gain, represents the ability to transfer power and, the matter of interest, data from one coil to another [START_REF] Lim | Fully Symmetrical Monolithic Transformer (True 1 : 1) for Silicon RFIC[END_REF]. This parameter was measured and the results are shown in Fig. 15. The prototype with the ferrite sheets shows an improvement of the coupling coefficient with k = 0.95, compared to the one without them, where k was 0.84. However, both investigated models were able to reach the same transmission gain S21 < 6 dB over the desired frequency bandwidth.

D. Channel TF

With the use of a two-port vector network analyzer (VNA) as illustrated in Fig. 16, a direct PLC channel TF measurement was done, to evaluate the channel flatness, along with the coupling efficiency between the transmitting and the receiving side. For investigation purpose, the channel TF magnitude measurement was made with two cable types, the shielded-twisted pair (STP) and the parallel pair (PP), also the effect of adding flexible ferrite sheet on the planar inductive coupler was evaluated. In Fig. 17 We can observe that the STP-PLC channel with the ferrite sheets has -20.78 dB coupling efficiency over 50% of the entire bandwidth, compared to the STP-PLC channel without the ferrite sheets, which has -23.34 dB coupling efficiency over 50% of the entire bandwidth. This leads to a 2.5 dB PLC coupling efficiency enhancement.

IV. PLANAR COIL COUPLER IMPLEMENTATION AND TESTS

In this part, the planar coil inductive coupler was used to test its effectiveness, and to validate its performance. Two narrowband PLC modems, adopting single carrier quadrature phase-shift keying (QPSK) modulation scheme were chosen to set the PLC channel, and run the tests, in order to get an overview on the coupling module efficiency, the latency and reliability, as well as the EMC compliance.

A. Latency and bitrate measurement

For latency and bitrate measurement, a carrier frequency of 19.4 MHz was chosen to transmit a predefined PLC Frame from the load side to the power supply (PS) side.

When the modems operate at 1.4 Mbit/s, the transmitted ASCII pattern, which are equivalent to 208 overall bits, take 419 µs to be received at the PS side, which leads to an actual 495 kbit/s bitrate. This time-delay is due to the approach in which the data is packed and protected before being transmitted over the powerline.

The modulated superimposed PLC signal over the powerline was measured, during the data transmission from the load side to the PS side as shown in Fig. 18

B. BER Analysis

To identify the carrier frequencies with the highest level of fidelity, a BER sweep test was done, where a defined PLC frame was transmitted in half-duplex mode, over different carrier frequencies, between 5 to 30 MHz frequency range.

This parametric sweep allows a full BER analysis of the PLC channel, to get data error rate for each carrier frequency, which can be manifested as received faulty bytes or as missed bytes at the receiver side. These results are evaluated in order to decide on which frequency range data can be transmitted reliably and safely with a low BER.

The performed tests were conducted over the STP channel, using the planar coil inductive coupling. As represented in Fig. 19, the bandwidths with the lowest BER can be identified mainly between 5 MHz to 11 MHz, 13 MHz to 23 MHz, and from 27 MHz to 30 MHz.

C. EMC compliance assessment

In this section, only the conducted emissions will be measured, at a lab test rig which was built and adapted to the typical one mentioned in [START_REF]Environmental conditions and test procedures for airborne equipment[END_REF], except for the RF absorbers of the shielded enclosure.

In Fig. 20 the EMC compliance measurements were made from both sides of the channel over the PP cable, using pearson electronics current probe (model 6600) connected to the digital signal oscilloscope (DSO), when transmitting the PLC frames as a differential mode signal, from the load to the PS. Furthermore, the noise floor was measured as well to differentiate the dc/dc converters switching noise.

Both switching noise level and data signal level exceed the Cat B-powerlines of the conducted emissions standard limitation, which makes the setup under investigation compliant with the Cat B-interconnecting bundles only, where the interference should be kept to a tolerable level.

The data signal level appearing at the used 19.4 MHz carrier frequency can be reduced to below 40 dBµA, when opting for a lower peak-to-peak data signal amplitude, which makes this solution susceptible to be compliant with the Cat Bpowerlines, as well as Cat L,M,H-interconnecting bundles with respect to the data signal level effect only.

V. CONCLUSION

In this paper an optimal PCB planar transformer was designed, to build a symmetrical PLC coupler that can achieve high coupling efficiency and transmission gain, over relatively wide operating frequency bandwidth, with a reduced size and weight, also with less signal degradation and with respect to the previously designed channel matching network.

The PLC planar coupler model was designed, investigated and confirmed with measured results, also it was successfully implemented to operate bidirectionally over point-to-point PLC channel.

The PLC implementation tests using this coupler were conducted within a lab environment in half-duplex mode, to operate with a latency of 419 µs.

Using the BER analysis, the frequency range with the highest reliability was identified.

With special respect to the DO-160G EMC-compliance standard, it is mainly dependable on the data signal level as well as the PS noise.

Fig. 1 :

 1 Fig. 1: (a) Parallel signal injection (b) Serial signal injection.

Fig. 2 :

 2 Fig. 2: Matching network with the PCB planar coil coupler.

Fig. 3 :

 3 Fig. 3: Trade-off between the inductance and the turns number for different coil shapes.

Fig. 4 :

 4 Fig. 4: π-model of a single layer PCB inductor.

Fig. 5 :

 5 Fig. 5: Side view of the PCB single layer inductor.

Fig. 6 :

 6 Fig. 6: CST-models of the tested planar coil shapes, (a) Square (b) Octagonal (c) Circular.

Fig. 7 :

 7 Fig. 7: Top view of the planar coils PCB-prototype.

Fig. 8 :

 8 Fig. 8: π-model simulation and measurement of the single layer inductors.

Fig. 9 :

 9 Fig. 9: Top view of an optimal circular shape double-layer planar coupler.

Fig. 10 :

 10 Fig. 10: Side view of the double-layer planar coupler.

Fig. 11 :

 11 Fig. 11: Double-layer PCB coupler equivalent circuit.

Fig. 12 :

 12 Fig. 12: Measurement setup for (a) capacitive coupling and (b) open and short circuit inductance.

Fig. 13 :

 13 Fig. 13: Measurement of the coupling capacitance, open and short circuit inductance.

Fig. 15 :

 15 Fig. 15: Transmission gain-S21 of the planar coupler.

Fig. 16 :

 16 Fig. 16: Block diagram of the PLC test bench.

Fig. 17 :

 17 Fig. 17: (a) PLC channel transfer function with planar coupler over the STP and PP (b) CDF on the PLC channel transfer function.

  (a), we can notice less severe peaks and notches over the STP case compared to the PP one. The cumulative distribution function (CDF) of the measured transfer function was computed and represented in Fig.

17

 17

  (b), to easily compare the results based on their coupling efficiency, so a decision can be made regarding the use of the ferrite sheet.

  (a) and (b) respectively, while the DC voltage was filtered.

Fig. 18 :

 18 Fig. 18: Latency measurement when transmitting a PLC frame from the load side (a) to the PS side (b) using the planar coupler.

Fig. 19 :

 19 Fig. 19: BER Sweep testing.

Fig. 20 :

 20 Fig. 20: Do-160G Conducted emissions standard compliance when transmitting data from the load side.

TABLE I :

 I Empirical values of the coefficients k 1 and k 2 [4] [5].

	Form	k 1	k 2
	Square	2.34	2.75
	Octagonal 2.25	3.55
	Circular	2.35	3.25

TABLE II :

 II Physical parameters.

	Parameter Value
	εrc	1
	εrs	4.3

TABLE III :

 III Geometry parameters of a potential compact design.

	Dimensioning parameters	Values
	Outer diameter dout	30mm
	Inner diameter d in	10mm
	Thickness of traces t	35µm
	Conductor's skin depth δ	20.6µm
	Width of traces w	273µm
	Space between traces s	150µm
	Total length of coil lt	1.83m
	Turns Number Ns	24
	Substrate thickness h	254µm

TABLE IV :

 IV Comparison of the measured and calculated parameters. Shape Calc Meas Calc Meas Calc Meas Calc Meas Square 14.26 14.36 3.31 3.72 5.05 3.22 18.80 23.61 Octagonal 11.73 12.68 3.31 2.29 5.05 2.88 20.75 26.31 Circular 12.96 12.71 3.31 2.48 5.05 2.23 30.18 29.28 a Measured/calculated at 10 MHz.

Parameters Ls(µH) a

Rs(Ω) a Cs(pF) a F srf (MHz)

TABLE V :

 V Geometry parameters of the double-layer planar coupler.

	Dimensioning parameters	Values
	Outer diameter dout	30mm
	Inner diameter d in	2mm
	Thickness of traces t	35µm
	Conductor's skin depth δ	20.6µm
	Width of traces w	150µm
	Space between traces s	450µm
	Total length of coil lt	1.41m
	Turns Number Ns	24
	Substrate 1 thickness h 1	254µm
	Substrate 2 thickness h 2	71µm
	MnZn sheet thickness h 3	200µm

TABLE VI below .

 VI.

TABLE VI :

 VI Calculated electrical parameters.

	Parameters	Calculated Values
	Ls	8.22 µH
	Rs	4.59 Ω
	Cs	4.10 pF
	C k	52.74 pF
	F srf	27.38 MHz

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ACKNOWLEDGMENT

This work was carried out under the frame of the ATIKA research project supported by Liebherr-Aerospace Lindenberg GmbH, and funded from Bundesministeriums für Wirtschaft und Technologie under 20Y1708E.


Abdelmoumin Allioua
email: abdelmoumin.allioua@lea.tu-darmstadt.de

Prof. Dr.-Ing Gerd Griepentrog
email: gerd.griepentrog@lea.tu-darmstadt.de

Martin Vögel
email: martin.voegel@liebherr.com

Julian Eitler
email: julian.eitler@liebherr.com

Dr.-Ing Nejat Mahdavi
email: nejat.mahdavi@liebherr.com

Aircraft DC networks characterization and adaptive stage design for PLC use

Keywords: PLC, Channel analysis, Noise characterization, Matching and coupling network, Reflection and VSWR, QPSK Modems, Latency, BER, RTCA DO-160G

Aerospace industry is moving towards a more electric aircraft (MEA) or even an all-electric aircraft (AEA), which will incite the focus on reducing the total harness mass and complexity, thus employing powerline communication (PLC) technique seems to be a potential solution to be implemented for airborne uses. This paper explores the feasibility to implement PLC technology over aircraft DC powerline, fed by switchedmode power supply. First, a channel characterization and analysis were conducted, also an adaptive stage was discussed and designed to transmit the data over an already existing powerline. Furthermore, the channel capacity was estimated along with the required data signal level. This study investigates also a realcase context with the use of switched-mode dc/dc converter which effect the PLC performance and signal integrity. Eventually, PLC implementation and tests were conducted as well as discussed from latency, bit error rate (BER) and EMC compliance perspective with the RTCA DO-160G standard requirements.

I. INTRODUCTION

Even not made for, the powerline can be relatively convenient for data transmission, depending on its voltage level and noise margin. The various types of harnesses and the presence of different connected branches, such as the powerelectronic devices, will also have an influence. With a beforehand accurate channel characterisation, an early prediction can be made on the best-case scenario for data transmission, and more adequate adaptation can be provided, to avoid impedance mismatches which causes multiple reflections and lead to high voltage standing wave ratio (VSWR), that can affect the data signal integrity by cancelling certain frequencies at the receiver. Furthermore, based on the channel noise characterisa-

The work presented in this paper has been financially supported by Liebherr-Aerospace Lindenberg GmbH, in the frame of the ATIKA research project, and funded from Bundesministeriums für Wirtschaft und Technologie under 20Y1708E. tion, the data signal level and the frequency bandwidth needed to achieve the desired bitrate can be identified.

The paper is structured as follows. In the next section, an approach to pursue towards characterizing aircraft DC networks is focused on, along with designing an adaptive solution, to implement powerline communication adequately for airborne use. In section III, we discuss about the channel capacity along with the minimum signal-to-noise ratio (SNR), and the frequency bandwidth (FBW) needed to achieve the desired throughput. In section IV, we present some results we obtain in our implementation tests with regard to the bitrate and latency, as well as the BER and the EMC-compliance. Finally, Section V concludes the paper.

II. CHANNEL CHARACTERISATION AND ADAPTATION

The channel under investigation represent a point-to-point interconnection between a power distribution and conversion module (PDCM) on one end, which represent a dc/dc converter that converts the 270 VDC to 28 VDC. In addition to a remote electronic unit (REU) feeding and controlling an actuator at the other end. For investigation purposes both devices are connected with a 25 m long special cable, either with an unshielded parallel pair (UPP) or a shielded twisted one (STP), where only one pair can operate sequentially.

Each element of the channel represented in Fig. 1, was characterised, modelled and measured in the frequency range from 1 to 30 MHz, to identify a suitable path for the data signal injection, which will be either by transmitting the data signal as a differential mode (DM), or as a common mode (CM) signal where the return path will be through the cable shield along with the PDCM and REU housing.

A. Cable Modelling

The investigated transmission line (TL) cross-section shown in Fig. 2, represent a 25 m long DC cable composed of two pairs, UPP and STP, where both pairs are joined together with protective sleeve to provide mechanical and thermal protection. The reason behind using a multi-optional cable, where only one pair can operate sequentially, is to compare their performance in terms of wave impedance, attenuation and propagation factor.

To reflect effectively the coupling between the operating pair and the floating one, a 3D CST studio frequency domain numerical model of the whole cable bundle was built and analysed based on its geometrical parameters, where the per unit length (p.u.l) RLCG parameters were extracted, together with the propagation constant, as well as the wave impedance.

B. Wave impedance

To extract the cable p.u.l parameters, two tests are required: the short-circuit to obtain R and L as shown in Eq. [START_REF] Idir | Skin Effect and Dielectric Loss Models of Power Cables[END_REF], and the open circuit determines the C and G in Eq. ( 2), as detailed in [START_REF] Idir | Skin Effect and Dielectric Loss Models of Power Cables[END_REF]. The wave impedance can be then calculated using Eq. ( 3) based on the already obtained results.

Z sc = R + jωL (1)
Z oc = 1 G + jωC (2)
Z w = R + jωL G + jωC = Z sc • Z oc (3)
Both open and short tests along with the wave impedance results over the UPP and STP for DM and CM path are shown in Fig. 3. As it has pointed out in black, the measured wave impedance is not completely flat, but it fluctuates over the frequency. To avoid the cancellation of certain frequencies at the receiver side, the suitable path is the DM one over the STP, which provides less severe peaks and notches as presented in Fig. 3 (a) with a mean impedance value of 53 Ω, compared to the DM-UPP and CM-STP paths as shown in Fig. 3 (b) and (c) respectively. Furthermore, using a symmetric TL DM path is less susceptible to crosstalk, and can handle well electromagnetic interference (EMI) potential attacks.

C. Power Source and Load Modelling

For the dc/dc converter, a passive two-port representation is adopted as shown in Fig. 4, where its output stage was simulated in CST STUDIO SUITE while considering the parasitic effects of each component, as well as measured using a E5061B vector network analyser (VNA). The DM impedance response was identified between port 1 and 2, however the CM impedance response was done between the two-ports shorted and the ground. The same strategy was used to model the REU input stage with its equivalent circuit in Fig. 5, to identify accordingly the impedance response for both devices in DM (inductive behaviour) and CM (capacitive behaviour) in the frequency band of interest, as represented in Fig. 6.

D. Signal attenuation and reflection coefficient

Based on the REU and the PDCM impedance response, along with the open and short transmission line impedance, the data signal attenuation and reflection coefficient were calculated for both CM and DM using Eq. ( 4) and Eq. ( 5). The results are represented in Fig. 7, and Fig. 8 respectively.

γ = α + jβ = 1 l c tanh -1 ( Z sc Z oc ) ( 4
)
Where γ is the propagation constant, α is the attenuation constant, β is the phase constant and l c is the cable length. Also Γ is the reflection coefficient, Z w represent the wave impedance, and Z L is the load impedance which is either the PDCM or the REU depending on the wave propagation sense. The attenuation constant refers to the loss of the signal amplitude per TL length, and as represented in Fig. 7, a slightly higher attenuation can be noticed on the STP path compared to the UPP path. This difference is due to the presence of additional insulating material in the shielded case compared to the unshielded one, which leads to a higher conductivity and permittivity that induces more electromagnetic field attenuation [START_REF] Knobloch | Shielded or Unshielded Twisted-Pair for High Speed Data Transmission ?[END_REF]. In Fig. 8, the reflection coefficient Γ fluctuates between -1 and 1, which reveals that the transmitted wave is completely reflected when |Γ|= 1, and leads to a high VSWR as shown in Eq. ( 6), that can damage the source and the load, and also affect the PLC signal integrity.

Γ = Z L -Z w Z L + Z w (5)
V SW R = 1 + |Γ | 1 -|Γ | (6)
The propagation velocity factor ϑ calculated in Eq. ( 7), refers to the propagation speed of the electrical signal through the cable compared to the speed of light.

ϑ = λ • f c ( 7
)
Where f is the first resonance frequency when the TL is one quarter-wavelength long, c is the speed of an electromagnetic wave in free space and λ is the 100m wavelength. In TABLE III, a comparison summary is shown between the UPP and the STP, where the wave impedance value, 1st resonance frequency as well as their propagation velocity, the attenuation also along with reflection and VSWR were compared.

E. Channel transfer function

To analyse the channel transfer function (TF) flatness as well as the coupling efficiency, a direct channel TF measurement was done using the VNA over both cable types, when two kind of bidirectional inductive couplers were designed and employed:

• Ferrite (MnZn) core toroidal coupler.

• Nanocrystalline (NC) core toroidal coupler. Each coupler was composed of two toroidal magnetic core with the same geometry, connected in series to improve the coupling efficiency, as shown with more details in [START_REF] Bilal | Design of broadband coupling circuits for power line communication[END_REF].

Fig. 9 (a) represents the measured channel TF for different cable and coupler configuration, where "STP-NC core" refers to the configuration with the STP cable and the inductive coupler with the NC cores.

Fig. 9 (b) shows the cumulative distribution function (CDF) of the channel TF, for example with the UPP-NC configuration the coupling efficiency is higher than -27dB and -31dB over 50 % and 90% of the entire bandwidth respectively. A high coupling efficiency provided by Nanocrystalline coupler compared to the Ferrite one can be observed. However, the channel TF with the Ferrite coupler has relatively less severe notches and peaks, which preserve the signal integrity at the receiver, also they are less sensitive to vibration in harsh environment compared to the Nanocrystalline one.

Regardless the acceptable results shown by the UPP-MnZn coupler configuration, and due to weight and space allocation reasons, also magnetic materials sensitivity and cost, the capacitive coupling will be used to implement and test PLC over the channel under investigation.

F. Matching network

To reduce impedance mismatching, there exist different approaches as detailed in [START_REF] Wang | A Review of Impedance Matching Techniques in Power Line Communications[END_REF] to decrease the reflection and the VSWR.

In order to avoid large electronic components number which leads to large circuit size and cost, the equal impedance matching method was chosen to be used, where an equivalent resistance value to the TL wave impedance was terminated from both sides of the cable. The used matching network is presented in Fig. 10, where decoupling inductors were utilized to block the AC data signal from leaking into the REU and the PDCM, also to improve the matching network performance, where their values must be equal or larger than the inductance magnitude of the REU and the dc/dc converter.

The inductance value effect on the matching network impedance response in shown in Fig. 11, and explained with more details in [START_REF] Grassi | Channel Characterization and EMC Assessment of a PLC System for Spacecraft DC Differential Power Buses[END_REF].

Two capacitors were added from both sides of the matching resistor to provide a low pass path for the data AC signal, and to block the 28 VDC across the matching resistor where the data signal will be injected to and separated from the powerline.

An impedance sweep was done and shown in Fig. 11, to confirm the effectiveness of the matching network, from the load and source side.

The impedance response with 3.3 µH and 22 µH blocking inductors reaches the desired impedance matching magnitude starting from 3 MHz and 800 kHz respectively.

The reflection coefficient and the VSWR were reduced accordingly.

III. CHANNEL CAPACITY

Based on Shannon's channel capacity theorem Eq. ( 8), and the channel conducted noise characterisation shown in Fig. 12, a trade-off was made between the channel capacity and the SNR for different frequency bandwidth,to identify the data signal level and the frequency bandwidth needed to achieve the desired bitrate.

C max = f bw . log 2 (1 + S N ) (8)
The theoretical channel capacity C max is the maximum achievable bitrate which is susceptible to decrease gradually due to BER. Thus, a safety margin is taken into account during the design phase.

Additionally, these trade-offs holds irrespective of the applied modulation scheme, which means that the throughput can be increased with regards to the used modulation order, or more precisely based on the number of transmitted bits per modulation symbol.

A. Ripple and noise measurement

The voltage measurement at the REU side was done to identify the AC content, the noise level over both the UPP and the STP, in Fig. 12

B. Signal-to-noise ratio (SNR) and frequency bandwidth trade-off

In spite of the fact that the practically achievable throughput is lower than the theoretical one, reaching 20 Mbit/s channel capacity with 5 MHz bandwidth for instance requires at least an SNR of 24 dB as shown in Fig. 13 (a), or injecting a minimum data signal RMS amplitude of 288 (mV) as detailed in Fig. 13 (b).

IV. PLC IMPLEMENTATION AND TESTS A. Off the shelf product review

A wide variety of PLC modems exist already on the market, most of them are used mainly for outdoor applications such as energy monitoring, smart grid communication and also to cover the so called 'last mile', as well as smart metering, solar renewable energy management and street light automation and control.

Indoor application modems are also widely used for automation in home, heating ventilation and air conditioning. However, there are still many fields which are not fully covered, such as aeronautic and automotive industry, where existing solutions need to be adapted to the desired application.

In TABLE IV, a summary on the possible suitable solution was made, based on the modems frequency range, data bitrate and modulation method. The two DCB1M narrowband PLC modems which adopting single carrier quadrature phase shift keying (QPSK) modulation scheme were chosen to set the PLC channel, and run the first experiments, in order to get an early overview on the coupling module efficiency, the latency and reliability, as well as the EMC compliance tests. In Fig. 14, a block diagram is shown to present the PLC implementation test bench, where defined data bit-stream will be transmitted from the REU side to the PDCM side, the CM noise will be measured from both sides of the channel using pearson electronics current probe (model 6600) connected to the digital signal oscilloscope (DSO), and also differential probes were used to measure the latency on the modems RXI and TXO pins.

B. Latency and bitrate measurement

To measure the latency and bitrate, a 19.4 MHz carrier frequency was chosen to transmit a PLC Frame of an A2Z ASCII pattern (equivalent to 208 overall bits) from the REU side to the PDCM side as shown in Fig. 15 An amplitude attenuation of the transmitted data signal can be noticed, along with a time-delay of a 446 µs for both idle and operation mode of the REU.

C. BER Sweep

In this part, a BER Sweep test was done to identify the carrier frequency with the highest BER, where A2Z ASCII pattern was transmitted in half-duplex mode on 250 different carrier frequencies, over 5-30 MHz frequency range, with 100 kHz step.

This mode allows a full sweep BER measurement, performed over the powerline, to identify the carrier frequency over which the data can be transmitted reliably and safely with low BER.

During this analysis, 1344 bytes were transmitted, which is equal to 10752 bits, and leads to a BER of ≈ 10 -5 .

However, transmitting a larger number of bits over a safe carrier frequency can achieve a lower BER.

The tests were performed without and with the matching network on the UPP channel with capacitive coupling as shown in Fig. 17

D. EMC Pre-compliance assessment

Both conducted and radiated RF emissions measurements require a shielded enclosure, however for the pre-compliance tests, a Lab test rig was built and adapted to the typical one, except for the RF absorbers since only the conducted emissions will be measured.

In Fig. 18 (a) and (b) the EMC pre-compliance measurement was made from both side of the channel over the UPP and the STP, when transmitting the PLC frames as a DM mode signal from the REU to the PDCM, the noise floor was also measured to differentiate the dc/dc converter switching noise.

Both switching noise level and data signal level exceed the 40 dBµA of the conducted emissions standard limitation, which make the under-investigation solution compliant with the Cat B-interconnecting bundles only, where the interference should be kept to a tolerable level.

The data signal level appearing at the used 19.4 MHz carrier frequency can be reduced to below 40 dBµA, when opting for 1V peak-to-peak instead of 2V peak-to-peak data signal amplitude.

V. CONCLUSION This paper has analyzed several aspects regarding PLC implementation for airborne use, and also demonstrated the necessity of a beforehand channel characterisation and adaptation stage design for avionic networks, to improve the reliability, safety and the compliance with the aerospace standard requirement. The investigated solution was successfully implemented in a lab environment over point-to-point network, to operate with a latency of 446 µs in half-duplex mode up to 1.4 Mbit/s, which is equivalent to 1.4 MHz bandwidth.

Regarding the carrier frequencies with the highest reliability, they were identified using the BER sweep, to avoid the signal loss at the receiver, caused by the channel transfer function non-flatness, which is essentially due to the medium, the used coupling method and the number of paths or the connected devices over the channel.

With special respect to the DO-160G standard, the EMCcompliance is essentially dependable on the power supply noise as well as the data signal level, which can be adjusted in future work to certify the solution for other categories.

Adopting PLC technology on-board aircrafts is highly dependable on the targeted communication systems as well. For example in the case of aircraft lighting systems, such as the branding lights or the wing inspection lights, the achieved results in terms of latency, bitrate and EMC compliance can be sufficient and adequate.

However, for time-critical and real-time aerospace application such as flight control systems (FCS), advanced requirements need to be considered during the design stage in terms of achieving higher bitrate (10 Mbit/s), and ensuring a lower BER of 10 -12 as on the existing avionic data buses such as the Avionics Full-DupleX switched ethernet (AFDX) [START_REF] Larhzaoui | Feasibility Study of a PLC System for Avionic Safety-Critical Systems[END_REF].

As a complementary future work, a more complicated pointto-multipoint network consisting of numerous communication participants or ports, including branching points shall be considered. So we can investigate the probable impedance mismatching which causes reflections and other destructive effects, such as the multi-path propagation with frequencyselective fading, to evaluate the suitability of PLC-technique in a more realistic airborne application scenario.

Fig. 1 .

 1 Fig. 1. PLC Channel under investigation.

Fig. 3 .

 3 Fig. 3. UPP and STP Wave impedance.

Fig. 4 .

 4 Fig. 4. PDCM output equivalent circuit.

Fig. 5 .

 5 Fig. 5. REU input equivalent circuit.

Fig. 6 .

 6 Fig. 6. PDCM and REU DM and CM impedance response.

Fig. 7 .

 7 Fig. 7. DM and CM attenuation per meter on the UPP and STP.

Fig. 8 .

 8 Fig. 8. Reflection coefficient on the UPP and the STP DM and CM path.

Fig. 9 .

 9 Fig. 9. Channel TF and CDF over UPP and STP using two cores inductive coupler.

Fig. 10 .

 10 Fig. 10. Matching network equivalent circuit.

Fig. 11 .

 11 Fig. 11. Matching network frequency response with different blocking inductors values.

  (a) and (b) the 109 kHz switching frequency components weight and repetition rate were clearly identified.

Fig. 12 .

 12 Fig. 12. AC content Measurement with filtered DC voltage.

Fig. 13 .

 13 Fig. 13. Trade-off between the Channel capacity, SNR and the required signal level.

Fig. 14 .

 14 Fig. 14. Block diagram of the PLC test bench.

  (a) and (b), where the DCB1M was at 1.4 Mbit/s operation mode which is equivalent to 1.4 MHz bandwidth as shown in Fig. 16.

Fig. 15 .

 15 Fig. 15. Latency measurement when transmitting PLC frame from the REU side.

Fig. 16 .

 16 Fig. 16. QPSK bandwidth and carrier frequency.

  (a) and (b) respectively. The matching network effect can be noticed, where the bandwidth over which the BER is low has increased significantly.

Fig. 17 .

 17 Fig. 17. BER Sweep testing w/o (a) and with (b) matching network while REU in idle Mode.

Fig. 18 .

 18 Fig. 18. Do-160G Conducted emissions standard compliance when transmitting from the REU side.

TABLE III SUMMARY

 III TABLE COMPARING THE UPP AND STP PARAMETERS AT 10 MHZ

	Cable	Wave	1st resonance Propagation
	Type	impedance	frequency	velocity	α	|Γ | SWR
		(Ω)	(MHz) b	(ϑ)	(m -1 )
	UPP Simu	83.50	2.13	0.710	0.0061 0.987 799
	UPP Meas	125	2.14	0.713	0.0040 0.995 445
	STP Simu	53.83	1.85	0.616	0.0086 0.996 512
	STP Meas	56.87	1.83	0.610	0.0151 0.984 128

b when the wavelength λ is 100m.

TABLE IV SUMMARY

 IV TABLE OF THE POSSIBLE EXISTING PLC MODEM SOLUTION

	Product	Frequency	Data	Modulation
		range	bitrate	scheme
	STMicroelectronics	[9-95 kHz]	Up to 128	OFDM-PHY
	ST7590		kbps	half-duplex
				(HDX) mode
	Maxim Integrated	[10-490 kHz] Up to 100	OFDM-PHY
	MAX79356 ZENO™		kbps	HDX mode
	Maxim Integrated [4.49-20.7 MHz] Up to 14	OFDM-PHY,
	MAX2982 ZENO™		Mbps	DQPSK, DBPSK
	EVSYS			HDX mode
	Yamar	[5-30 MHz]	Up to 1.4	QPSK
	DCB1M EVB		Mbps	HDX mode

Authorized licensed use limited to: ULB Darmstadt. Downloaded on November 23,2021 at 07:43:53 UTC from IEEE Xplore. Restrictions apply.

ACKNOWLEDGMENT

This work was carried out under the frame of ATIKA research project supported by Liebherr-Aerospace Lindenberg GmbH, and funded from Bundesministeriums für Wirtschaft und Technologie under 20Y1708E.

Julian Eitler is an electrical engineer working at Liebherr-Elektronik GmbH in Lindau, Germany since 2021. After studying electrical engineering (subject communications technology) at university of applied siences Ravensburg-Weingarten he graduated with a bachelor's degree of Engineering in 2019. Before working for Liebherr-Elektronik GmbH he was employed at Liebherr-Aerospace Lindenberg GmbH as an engineer for electrical systems and was part of the ATIKA team.


COMMISSARIAT A L'ENERGIE F Dacquait

CEA / DEN France C Atomique

M Andrieu

Electricite Berger

De

C Andrieu

M Berger

CORROSION PRODUCT TRANSFER IN FRENCH PWRs DURING SHUTDOWN

Physicochemical conditions significantly vary during cold shutdown in French PWRs: power and temperature decrease, chemical conditions are modified (pH decrease and change from reducing to oxidizing conditions). These changes involve corrosion product releases in the primary water (dissolving and crud bursts) which can be seen by a considerable corrosion product increase, from 2 to 3 orders of magnitude, in the activity concentrations (about a hundred GBq/t of 58 Co, about one GBq/t of 60 Co…) and in the chemical species quantity (several kilograms of nickel, several hundred grams of chromium…). The 58 Co peak activities which are lower than the French PWR average are generally due to delayed or incomplete oxygenations or unscheduled shutdowns with oxygenation during operating cycles. An unscheduled shutdown with oxygenation in the last six months of an operating cycle must reduce the 58 Co activity peak during refueling shutdown, more especially as the unscheduled shutdown occurs near the end of the cycle. Thus, when the shutdown happens in the three months prior to the end of the cycle, it is very likely that the 58 Co peak during the refueling shutdown will be two times lower than the expected peak.

The large quantity of corrosion products released during shutdown mainly come from the dissolution of the incore deposits due to primary system oxygenation, in particular the dissolution of nickel metal or nickel oxide, leading to the dissolution of the 58 Co derived from the nickel sites in the crud. Thus, the total 58 Co activity released during oxygenation is equal to, or even higher than, the out-of-core total surface activity. Therefore, the cold shutdown procedure must prevent the transfer of this considerable activity from the in-core to the out-ofcore surfaces. Deposited activity measurements carried out in French PWR primary circuits have never shown any visible contamination reduction during cold shutdown. On the contrary, recontamination can occur because of a temperature plateau or an oxygenation at a temperature over 80°C.

That is the reason why:

• Primary circuit oxygenation at 80°C, • Rapid temperature decrease and • Improvements on primary fluid purification, during cold shutdowns, have, up to this day, been the main points in the EDF strategy to reach the required dosimetric goals (ALARA), all the while optimizing unit outage schedules (availability). This paper thus proposes a description of corrosion product behaviour during French PWR shutdowns in order to optimise cold shutdown procedures.

INTRODUCTION

Physicochemical conditions significantly vary during cold shutdown in French PWRs: power and temperature drops, chemical condition changes of the primary coolant. These changes lead to corrosion product releases in the primary coolant (dissolving and crud bursts), which can be seen by considerable corrosion product increase in the radiochemical and chemical species. This is why the cold shutdown procedure must meet the following objectives:

• Minimize corrosion product release when this does not constitute an out-of-core contamination decrease, • Lower primary coolant activity as quickly as possible,

• Avoid contaminating the primary circuit, • Facilitate fuel-handling operations while maintaining reactor cavity water transparency and reducing the surface dose rate during core unloading.

To meet these objectives, studies have been carried out in France to optimise the shutdown procedure for many years. These studies are founded on the chemical and radiochemical analysis of primary coolant samplings and on the measurements of the activities deposited on the circuits during cold shutdowns.

COLD SHUTDOWN DESCRIPTION

The physicochemical parameter variations during a cold shutdown on a French 1300 MWe PWR (Reactor A) are described in Figure 1. After an H 2 decrease to a content lower than 3 cc/kg STP, in order to oxygenate the primary circuit, 15 litres (900 MWe PWR) or 23 litres (1300 MWe PWR) of hydrogen peroxide (30%) are injected and the vapor space of the CVCS volume control tank is scavenged. The hydrogen peroxide injection is carried out when the primary coolant temperature is at 80°C, if possible with all the primary coolant pumps in operation.

Because of the oxygenation, corrosion product activity concentrations significantly increase in the primary circuit, especially the

ACTIVITY AND CHEMICAL ELEMENT CONCENTRATIONS

To assess corrosion product behaviour, primary coolant samplings are made during PWR cold shutdown. During the reactor A shutdown described in the previous paragraph, 250 to 1000 ml samplings were filtered through 0.45 µm millipore filters and ion exchange papers. The volume activities of the corrosion products in solution and suspension were measured by gamma spectrometry (. Figure 2 and Figure 3), the chemical concentrations by X-ray fluorescence (. Figure 4). Thanks to these measurements, the total released nickel mass, the 58 Co total released activity and the 58 Co specific activity during the shutdown can be evaluated (. Figure 5).

Co total release and specific activity

Moreover, during another 1300 MWe PWR (Reactor B) shutdown, particle size analyses were determined by 1 litre samplings through a succession of millipore filters with decreasing meshes: 10 µm, 5 µm, 1.2 µm and 0.45 µm ( Figure 6). Power reduction is characterized by a rapid increase in volume activity. However, this increase is slight, about 100 GBq of 58 Co, versus the total activity released during shutdown. This activity mainly consists of particles, part of which deposits very rapidly. The radionuclide particle size is between 5 and 10 µm for reactor B before shutdown, while it is mostly greater than 10 µm during hot shutdown. Co activity ratio, about 20 (this varies between 2 and 6 for out-of-core deposits), and the 58 Co specific activity, about 30 GBq/g (this is about 45 GBq/g at in-core equilibrium), show that these particles originate from an in-core deposit detachment, due to thermal and mechanical stress variations during control rod insertion.

The primary coolant boration and temperature decrease (from 290 to 180°C and then from 180 to 80°C) result in a fast increase in the activity and chemical concentration mainly by dissolution. This corrosion product dissolution is related to the solubility increase due to temperature and pH decreases. The 58 Co/ 60 Co ratio, about 20, and the 58 Co specific activity, less than 10 GBq/g, indicate that this dissolution affects in-core and out-of-core deposits. The activities and masses released in the primary coolant come from deposit decomposition (nickel ferrite). During the temperature plateau at 180°C, the similar variation of the nickel and iron concentrations and the low 58 Co specific activity (less than 5) show that this nickel ferrite decomposition especially takes place on out-of-core surfaces. As the soluble radionuclide ratio decreases after the dissolution due to the falling temperature or pH, part of these radionuclides must precipitate. For reactor B during the temperature plateau, approximately 25% of the particulate activity is in the 5-10 µm range and about 75% greater than 10 µm. During the temperature decrease from 180 to 80°C, half of the particulate activity is equally distributed between the 1.2-5 µm and 5-10 µm ranges, the other half being greater than 10 µm. The 58 Co activity concerned during this phase is about several thousands of GBq, a low activity compared to the total activity released during shutdown.

Primary circuit oxygenation causes a large dissolving of nickel (a concentration of about 8 ppm), and, to a lesser extent, of cobalt. On the other hand, iron precipitates to form particles (Fe 2+ oxidized to Fe 3+ ), the 59 Fe activity being mainly in particulate form. Chromium, in particulate form under reducing conditions, dissolves in oxidizing conditions. The particle size distribution is not greatly modified by oxygenation. The high 58 Co/ 60 Co ratio (> 200) and the 58 Co specific activity (approximately 25 GBq/g) show that an in-core nickel deposit dissolves. It is an in-core nickel metal or nickel oxide deposit that is stable in reducing conditions at high temperature and that dissolves in oxidizing conditions at low temperature. This is confirmed by an experiment in the SOZIE loop [START_REF] You | Corrosion product transfer in PWR primary circuits during cold shutdowns[END_REF] Co released activity increase which lasts during several shutdowns after the SGR cycle [START_REF] Dacquait | Engler -Impact of Steam Generator Replacement on PWR primary circuit contamination by radioactive corrosion products[END_REF]. For the 1300 MWe PWRs, the higher peaks occurred during the first cycles, probably because of a larger release of Steam Generator (SG) tubing at the beginning of life. Note that the SG tubing material in the French PWRs is alloy 600 for the oldest or alloy 690 for the most recent.

The low peak values are generally due to delayed, incomplete oxygenations or unscheduled shutdowns with oxygenation in the course of cycle. The unscheduled shutdowns allow nickel and 58 Co deposited on fuel to be solubilized and purified. These deposits do not have time to reform before the refueling shutdown.

Figure 7 shows the ratio between peaks during refueling shutdowns after an unscheduled shutdown with oxygenation and the expected peaks if there had not been any unscheduled shutdown, versus operating time after an unscheduled shutdown with oxygenation.

It appears that an unscheduled shutdown with oxygenation of about 6 months before the refueling shutdown allows the 58 Co volume activity released during the refueling shutdown at the end of the cycle to be reduced, particularly as this unscheduled shutdown occurs near the end of the cycle. Thus, when the shutdown occurs three months before the end of the cycle, it is extremely likely that the peak during the refueling shutdown will be reduced by at least half compared to the expected peak.

However, before the increase in the primary circuit temperature during an unscheduled shutdown, the corrosion product volume activities must be sufficiently low (EDF criteria: 58 Co < 7 GBq/t and γ total < 14 GBq/t) to prevent their precipitation, due to temperature and pH increase, from causing their deposition on out-of-core surfaces. This risk of primary circuit recontamination would lead to a dose rate increase during the refueling shutdown at the end of the cycle.

PRIMARY COOLANT PURIFICATION

The previous paragraph showed that, during a cold shutdown, a great quantity of corrosion products was transported by primary coolant. This huge quantity of corrosion products is trapped in the filters and demineralizers of CVCS as shown in Figure 8. The CVCS Decontamination Factor (DF = upstream filtration activity / downstream filtration activity), given during a 1300 MWe PWR shutdown, shows that purification efficiency is high for radioactive corrosion products: from 100 to 10 6 . It would seem that the higher the decontamination factor, the more significant the upstream filtration activity. But this is perhaps also due to the particular behaviour of the ion exchange resins with respect to radioelement retention. Ag), two complementary research projects are under development and design engineering at EDF: increase the purification flow rate and optimise the purification supports [START_REF] Berger | Rocher -Policy and Good Practices in Electricité de France PWRs, for Primary Coolant Purification during Refuelling Cold Shutdown[END_REF].

Volume activity A t (at moment t) in the primary circuit is :

)] 1 1 ( [ 0 exp DF M t Q t A A - ⋅ ⋅ - ⋅ =

with

• A 0 : initial volume activity • Q : CVCS letdown flow rate • M : total water mass to be purified (including CVCS volume)

The decontamination factors being higher than 100, this expression is simplified in:

] [ 0 exp M t Q t A A ⋅ - ⋅ =
Thus, the purification flow rate increase allows an increase in the CVCS purification constant to be obtained (Q/M). One can discount a significant savings of time in obtaining the primary pump shutdown criteria (availability) or, for the same scheduled purification time, in obtaining the prescribed activity criteria (ALARA) below. Moreover, the increase in the CVCS purification flow rate helps meet the reactor cavity filling criteria more quickly.

Consequently, the increased CVCS letdown flow rates retained by EDF are:

SURFACE ACTIVITIES

In order to evaluate the cold shutdown impact on contamination, the activities deposited in the PWR primary circuit have been measured for many years by means of the EMECC gamma spectrometry system [START_REF] Eimecke | Anthoni -Ensemble de Mesure et d[END_REF]. These measurements are carried out before and after oxygenation, mainly on hot legs and SG tube bundles.

Figure 9 shows the variation in the Co activities deposited on the primary circuit during oxygenation has never been observed. However, a strong increase in 58 Co deposited activity can occur on out-of-core surfaces if the effective oxygenation temperature of the primary circuit is higher than 80°C ( Figure 10) [START_REF] Anthoni | Caramel -Behaviour of the corrosion products during PWR shutdowns, Chemistry in Water Reactors: operating experience & new developments[END_REF]. Recontamination then occurs very rapidly (less than one hour) and at the time of the activity peak. On the contrary, the oxygenation temperature does not seem to have an impact on 60 Co contamination ( Figure 9). This is due to the fact that the 60 Co released quantity, suitable to be deposited on the out-of-flux surfaces, is probably not significant enough to be measured. Let us note that at identical oxygenation temperatures, 58 Co SG recontamination, when significant, is roughly the same as in the procedures using hydrogen peroxide or CVCS venting ( Figure 10).

Figure 1 :

 1 Figure 1 : Cold shutdown -Parameter variation

Figure 3 :

 3 Figure 3 : Dissolved radionuclide proportion

Figure 6 :

 6 Figure 6 : Radionuclide particle size distribution before and during shutdown

Figure 7 :

 7 Figure 7 : Impact of unscheduled shutdowns with oxygenation

Figure 9 :

 9 Figure 9 : Surface activity variation during oxygenation On approximately 80 analysed shutdowns, a visible reduction in the 58 Co and 60

Figure 10 :

 10 Figure 10 : Impact of the effective oxygenation temperature

Table 1 : Peaks during refueling shutdown

 1 . The nickel dissolution involves the dissolution of

	58 Co, derived from the nickel

Table 2 : Activity criteria

 2 Thus, the activities and the masses of the "traditional" corrosion products are trapped by the prefilter and by the ion exchange resins of the CVCS. On the other hand, other undesirable radioelements such as The activity criteria currently specified for corrosion products at EDF during cold shutdowns in French PWRs are defined in Table2.

		1.E+6 1.E+7	H 2 O 2 injection	0 200	Flowrate (t/h) Temp. (°C)		1.E+5 1.E+6	H 2 O 2 injection	0 200	Temp. Flowrate (t/h) (°C)
	Activity (MBq/t) -DF	1.E+4 1.E+5		-400 -200 58Co 58Co -DF 60Co 60Co -DF CVCS flowrate	Activity (MBq/t) -DF	1.E+3 1.E+4		-400 -200 54Mn 54Mn -DF 51Cr 51Cr -DF CVCS flowrate
		1.E+3		-600 Temperature		1.E+2		-600 Temperature
		1.E+2		-800			1.E+1		-800
		25/09	26/09	27/09			25/09	26/09	27/09
				Figure 8 : Purification efficiency
	as well as	122 Sb and Ag, are not systematically properly trapped by filters and demineralizers and these can significantly 124 Sb, 110m
	add to dosimetry [5].
			Hold points		Activity criteria relating to corrosion products during purification by CVCS
			Primary pump shutdown		58 Co < 50 GBq/t and γ total < 100 GBq/t
			Reactor cavity filling			58 Co < 2 GBq/t and γ total < 4 GBq/t
	In order to quickly meet these goals by means of purification, while taking into account specific problems (presence of 122 Sb 124 Sb and 110m

Co volume activity.

The increases in the 58 Co activities deposited on the hot legs and the SG tubes during oxygenation presented in Figure 9 are due to an effective oxygenation temperature higher than 80°C, in particular for the most significant variation rates. However, these increases can also be due to prolonged temperature plateaux.

Thus Figure 11 shows the evolution of the 58 Co activity deposited on a SG tube bundle during a cold shutdown characterized by a temperature plateau at 120°C during 5 days. Although the volume activity during this plateau is not very significant (between 15 and 20 GBq/t), the SG 58 Co contamination increases appreciably during the plateau at 120°C. Note that since part of the 58 Co activity is released during the temperature plateau, the activity peak due to oxygenation at 80°C is not very high (33 GBq/t). However, the total released activity is equivalent to what it would have been without a temperature plateau, but with a more significant peak at the time of oxygenation. Finally, let us note that the H 2 O 2 injection temperature being 80°C, oxygenation did not lead to primary circuit recontamination. During reactor A shutdown ( § Cold shutdown description), about 95% of the total 58 Co released is due to oxygenation. This activity is equal to, or even higher than, the total activity deposited on the out-of-core primary surfaces. Indeed, the 58 Co total activity released during the shutdown is about 150000 GBq, whereas the 58 Co out-of-core total surface activity is about 75000 GBq before and after oxygenation, that is to say two times less. This confirms that the activity released during oxygenation comes from in-core deposits and not out-of-core deposits. This activity is due to the dissolution of nickel oxide or metal nickel in-core deposits in an oxidizing medium, leading to the dissolution of the 58 Co derived from the nickel sites in the crud. Although the released activity is primarily in soluble form, crud bursts occur during cold shutdown, and these particle sizes can be relatively significant: larger than 10 µm.

Gamma spectrometry measurements carried out by the CEA have never shown a visible reduction in the 58 Co and 60 Co contamination on the out-of-core primary surfaces. On the other hand, a significant 58 Co recontamination of the primary circuit can occur if the effective temperature of the primary circuit oxygenation is higher than 80°C. Moreover, 58 Co recontamination is probably accompanied by a cobalt and nickel re-deposition. These elements are redistributed during the following cycle and can be activated in core, thus creating an additional source of radioactive corrosion products.

In the paragraph "Activity and chemical element concentrations", it was stated that the nickel ferrite deposits were to decompose during the temperature plateau at 180°C. But this dissolution of the out-of-core deposits is negligible, since it represents less than 1% of the out-of-core total surface activity. Moreover, the measurements of the activities deposited on the primary circuit have never shown a reduction in the activities ([H 2 ] < 3 cc/kg). According to [START_REF]PWR Primary Water Chemistry Guidelines[END_REF], the dissolved hydrogen content should be higher than 10 cc/kg and the temperature should range between 120 and 150°C during at least 16 hours, in order to decompose nickel ferrites. However, the reaction kinetics is not known and a temperature plateau can lead to out-of-core surface recontamination (. Figure 11). The hypothetical benefit is thus likely to be masked, all the more so as the acidreducing conditions can also decompose the in-core nickel ferrites and thus recontaminate out-of-flux surfaces. In order to evaluate its impact, an acid-reducing chemistry will be applied and monitored during a French PWR shutdown this year.

In France, the current procedure for the cold shutdown prescribed by EDF does not aim at decontaminating the out-of-core surfaces, but, at, as soon as possible, reaching the criteria of the reactor cavity filling by avoiding recontaminating the primary circuit out-of-core surfaces. That is why:

• The primary coolant temperature is reduced as quickly as possible,

• The oxygenation temperature of the primary circuit is 80°C,

• The high flow rate purification is being evaluated so as to be applied to EDF PWRs. The corrosion products are effectively trapped on the CVCS filters and demineralizers, except for specific pollutions (antimony and silver) for which studies on the optimisation of the purification media are in progress.

Although penalizing, among other things, to availability, an unscheduled shutdown with oxygenation can have a beneficial effect on the refueling shutdown at the end of the cycle. Indeed, an unscheduled shutdown with oxygenation in the last six months of a cycle allows the 58 Co activity peak to be decreased during the refueling shutdown. As for purification, an unscheduled shutdown makes it possible to more quickly reach the criteria of reactor cavity filling or, for the same purification duration, to obtain activity values below the criteria. This is why EDF recommends oxygenating the primary circuit if the unscheduled shutdown lasts more than 3 days. Moreover, it allows the reduction of out-of-core contamination, since part of the in-core corrosion product deposits is removed and trapped by CVCS filtration. Let us recall that it is advisable to purify the primary circuit sufficiently to avoid recontaminating the out-of-flux surfaces, because of the temperature and pH increase at the end of the unscheduled shutdown (EDF criteria: 58 Co < 7 GBq/t and γ total < 14 GBq/t). These criteria are to be respected all the more so as, since the unscheduled shutdown occurs near the end of the cycle, the radionuclides have less time to decrease.

CONCLUSION

Cold shutdown in French PWRs leads to considerable corrosion product increase in the primary coolant (2 to 3 orders of magnitude). This is highlighted by high activity peaks (about a hundred GBq/t of 58 Co, about one GBq/t of 60 Co) and high chemical concentrations (several ppm of nickel…). This abrupt increase is due to the changes in the thermal and chemical conditions of the primary circuit: temperature and pH decrease, change from oxidizing to reducing conditions. This significant quantity of corrosion products primarily comes from the dissolution of in-core deposits during primary circuit oxygenation, in particular, the dissolution of metal nickel or nickel oxide, involving the 58 Co formed from nickel. Thus, the total 58 Co activity released during oxygenation is equal to, or even higher than, the total activity deposited out-of-core. It is thus advisable that the cold shutdown procedure be adapted to avoid transferring this considerable activity from in-core surfaces to out-of-flux surfaces, particularly as the measurements of primary circuit surface activities in French PWRs have never shown a visible reduction in contamination during cold shutdown. On the contrary, recontamination can occur during temperature plateaux or oxygenations at a temperature higher than 80°C.

That is why:

• Primary circuit oxygenation at 80°C, • Rapid temperature decrease and • Improvements on primary fluid purification, during cold shutdowns, have, up to this day, been the main points in the EDF strategy to reach the required dosimetric goals (ALARA), all the while optimizing unit outage schedules (availability).


Abdelmoumin Allioua
email: abdelmoumin.allioua@lea.tu-darmstadt.de

David Krause

Andrea Zingariello

Gerd Griepentrog

Reduction of DC/DC Converters EMI Emission Using Bi-and Unidirectional QR-ZVS Topologies

come

Introduction ZVS Condition in QRCs

Conclusions

Contact

Load Regulation Dynamics

Converter Suitability

•

VCR analysis with normalized loads (0.1, 0.2, 0.4) indicated HW topology's high load-dependence, while FW topology demonstrated robust stability, enhancing component lifespan.

•

FW leads to faster initial condition recovery and shorter turn-off times, especially beneficial at higher output currents.

This analysis determines the ZVS boundaries.

• Permissible duty cycle variation for ZVS: ±12.6% (full-wave) and ±12.3% (half-wave).

• Corresponds to time windows of 31.51 ns (full-wave) and 30.79 ns (half-wave).

• This range ensures operational flexibility and ZVS capability under varying input voltages and loads.

QR-ZVS Converter

Introducing a 70W buck converter for 48V/24V conversion, incorporating a hardswitching benchmark and two QR-ZVS variants for half-wave (HW) and full-wave (FW) soft switching.

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VCR concept, influenced by normalized switching frequency and normalized load, is essential in QRCs operation and components selection [START_REF] Tabisz | Zerovoltage-switched quasiresonant buck and flyback converters -Experimental results at 10 MHz[END_REF] [START_REF] Wuti | Analysis and Design of a Zero-Voltage-Switched (ZVS) Quasi Resonant Buck Converter Operating in Full-Wave Mode[END_REF].

• For ensuring ZVS across various loads, the condition 𝑟 ≤ 𝑀 must be satisfied [START_REF] Wuti | Analysis and Design of a Zero-Voltage-Switched (ZVS) Quasi Resonant Buck Converter Operating in Full-Wave Mode[END_REF].

• GaN-HEMTs offer superior performance with lower on-resistance and output capacitance, enabling efficient multi-MHz switching and reduced losses [3][4]. Soft-switching converters, utilizing ZVS, improve efficiency by reducing turn-on losses. Hard-switching (HS) efficiency is at 72.49%, while soft-switching FW operates at 80.30% and HW at 83.44%, with the FW's efficiency slightly reduced due to additional 𝐷 s conduction losses.

𝑓 s 𝑓 R = 2𝜋 1 -𝑀 𝛼 + 𝑟 2𝑀 + 𝑀 𝑟 1 -𝑐𝑜𝑠 𝛼 , 𝛼 = 𝜋 + 𝑎𝑟𝑐𝑠𝑖𝑛 𝑟 𝑀 Switch Configuration D f S C out U in R L L f D r

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Results indicated that hard-switching converter showed higher CM noise under 3.5 MHz, whereas soft-switching converters had more noise from 3.5 to 19 MHz due to transients and resonant components. Above 19 MHz, soft-switching significantly reduced EMI.

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CDF analysis, soft-switching converters maintained noise levels under 40 dBµA, showing better EMI performance than hardswitching converters, which reached over 60 dBµA within the same frequency bandwidth. This study investigates the use of QR-ZVS and GaN-HEMTs in MHz-range SMPSs to improve efficiency and mitigate EMI. By employing soft-switching QRCs with ZVS topology, we address the challenges of increased stress and instability from HSF operation, optimizing voltage control and cost-effectiveness while reducing losses and EMI.

•

Potentially suitable for automotive applications, especially in heavy-duty vehicles and trucks.

Possible applications: Starting systems, lighting, ignition controls, auxiliary power for onboard electronics.

Potential Improvements: Enhanced EMI filtering can further align with the stringent demands of automotive applications.

Our study introduced QR-ZVS converter approach to minimize EMI noise and enhance efficiency at 4 MHz SF using GaN-HEMT. The soft-switching prototypes outperformed the hardswitching design in efficiency and EMI noise reduction, demonstrating less sensitivity to load variations, particularly in full-wave topology. However, HSF posed thermal challenges, indicating a need for careful thermal management in future designs. Additionally, EMI filter refinement is needed to meet specific EMC standards such as CISPR-25.

Figure 1 .

 1 Figure 1. Buck converter configurations.

Figure 6 .

 6 Figure 6. Frequency spectrum of the converters input CM noise.

Figure 5 .

 5 Figure 5. Schematic of the conducted EMI measurements setup.

Figure 3 .

 3 Figure 3. VCR analysis for different 𝑟 values in (a) HW, and (b) FW.

Figure 4 .

 4 Figure 4. State locus curve of 𝑈 𝐶 R and 𝑍 R in HW and FW operation. (a) Measured, and (b) Simulated.

Figure 2 .

 2 Figure 2. Measurement representing the switching signal 𝑈 SG and the resonant capacitor voltage 𝑈 𝐶 R . In (a) FW and (b) HW operation.