This thesis demonstrates the feasibility of using gallium nitride (GaN) technology in reconfigurable RF systems. GaN-based varactor diodes and switch circuits are pursued as promising candidates for high-power/high-frequency applications. The first part is devoted to active GaN device development. Active components were realized using the Canadian National Research Council (NRC) GaN HEMTs process. Based on three process, such as, GaN150v0 (gate length of 0.15um), GaN500v1 and GaN500v2 (both with gate length of 0.5um), many varactor diodes with size different have been manufactured and characterized via DC and RF smallsignal and large-signal measurements. Then, the varactor diodes were modeled by analytic equations containing empirical coefficients. These expressions have been introduced for the first time for the voltage dependency of equivalent capacitance (CEq) and series resistance (REq) and can be used as a general model to represent the nonlinear behavior of GaN based varactors. For small-signal operation, all of the developed equations describing REq and CEq are only bias voltage and device geometry dependent, while for large-signal operation, the influence of RF-power must be taken into account. In addition, different size single stand-alone switches were fabricated using GaN500v2. By analyzing the small-signal measurements, it was observed that the isolation is high at low frequencies but quickly drops with increasing frequency. Also, it was observed that the larger the device the lower the insertion loss and the poorer the isolation will be. Moreover, based on these small-signal measurements, a model was introduced.

The second part addresses the integration and design aspects of the reconfigurable proposed circuits, such as tunable phase shifter, reconfigurable 3-dB 90° hybrid coupler, tunable frequency oscillator, beam switching antenna array and matching reconfigurable patch antenna operating below 10 GHz. Here the developed GaN varactors and switches are used for achieving the tenability purpose.

List of tables

x Also, special thanks are reserved for fellow colleagues Tyler Ross, Nathan JESS, Shiecheng YANG and Alexis MARTIN, who continue to set an example for graduate students with their thirst for knowledge and enthusiasm to share it.

On a personal note, I would like to express my sincerest gratitude to my brother Bouchaib and his wife, Bouchra, for their financial and moral support since I came to France for study in October 2006. I would also like to express my deepest gratitude to my parents, brothers, and sisters (meaning also nieces) for the continuous encouragement which helped me endure tough moments. My gratitude also goes to my friends in Rennes and Ottawa, who encouraged me when needed. The name of best friends in particular should be highlighted. Tarik DANKIR for his encouragements and moral support and Imad ABDELJAOUAD who has been an inspiration and example for me since I first moved to Canada for my PhD degree. Over the past decade, several changes have drawn more attention to RF/microwaves power devices. Increasing demands of high power handling capabilities and worries about capabilities of functioning in harsh environments have created a need for alternatives to traditional semiconductors technology. Due to its high critical breakdown field associated with wide band gap energy as well as its high saturated electron velocity, gallium nitride (GaN) is a potential solution to satisfy these simultaneous requirements for RF, microwave and mm-w transmitters for communications and RADAR [START_REF] Runton | History of GaN: High-Power RF Gallium Nitride (GaN) from Infancy to Manufacturable Process and Beyond[END_REF]. On one hand, GaN-based high electron mobility transistors (HEMTs) power devices are considered as an excellent candidate for future power devices due to its high breakdown voltage and high saturation drain current [START_REF] Runton | History of GaN: High-Power RF Gallium Nitride (GaN) from Infancy to Manufacturable Process and Beyond[END_REF]. On the other hand, besides being used for high power purposes, GaN

HFETs devices are also a strong candidate to be used in harsh environment receiver applications due to the robust nature and good low noise performance of these devices [START_REF] Ross | Gallium nitride hybrid microwave circuits for low-noise applications[END_REF], where others semiconductors can suffer.

With the advent of this wide band gap material, enormous progress has been made until now especially in the development of GaN technology. Consequently, more and more industries and academic research labs in the field of semiconductors have introduced different GaN HEMTs processes (i.e United Monolithic Semiconductors (UMS), TriQuint, Cree, NRC (National Research Council Canada), etc). These devices are intended to fulfill the growing demands for high power at high frequency electronic components as well as for high voltage power switches operating at higher frequencies. Furthermore, GaN HEMTs based MMIC technology is now challenging the other traditional semiconductor technologies such as GaAs mostly by offering the highest level of integration and the smallest form factor while keeping to operate with high voltage and to handle high power levels.

However, some challenges remain with GaN technology. The most serious problem is the cost of a GaN substrate in which the power devices are fabricated and the pace of maturation of their technological processes. Another issue is the lack of bulk GaN source material which leading to the need for GaN growth by using mismatched substrates such as Si, SiC and sapphire.

The recent trend in RF/microwaves technology is toward circuits that are highly reconfigurable compacted and reliable with more functionalities, with low power consumption, and reduced cost. In particular, the downsizing demands in RF/microwaves wireless communication systems are growing exponentially as the number of multi-band and multi-service systems has increased along with the need for high power handling capabilities at higher frequency. The tunable and reconfigurability aspect of modern RF/microwaves front-ends transceivers (Tx/Rx) is also an interesting subject concerning GaN HEMTs technology. Research into the usage of GaN based high power RF switches and varactors provide further opportunities for GaN to achieve these demands.

GaN technology is emerging as a major promising candidate for reconfigurable high power circuits for RF front-end Tx/Rx. Moreover, there is a wide consensus that the antenna system should be integrated in the same package with the front-end integrated circuit [START_REF] Johannsen | Modular antenna array concept for millimeter-wave beam-steering applications[END_REF]. As an example, mobile communications (including SATCOM) imply that reconfigurable radiation pattern antennas are now of interest for such applications to steer the antenna beam. So antenna array system with a controlled gain and phase between single elements are often considered. In particular, for mobile applications, the antenna array must exhibit the capability to align its beam in the direction of the strongest signal. Hence, an antenna array with beam-steering capability is imperative and could be integrated on the same package with its two crucial components, phase shifter and gain amplifier to produce the desired amplitude and phase.

Fulfilling the in-depth development of GaN HEMTs technology for tunable high power circuits, it has become the motivation of this research to investigate novel highly compact reconfigurable active microwave circuits operating up to X-band frequency by using developed GaN varactor and switch models.

The main aim of this thesis research is to investigate and characterize GaN-based varactor diodes and switches that will be integrated to realize

II. Objective of Research

The proposed overall objective of this research is to study reconfigurable integrated microwave circuits using developed GaN varactor and switch models for use, for example, in X-band systems. It is expected that this project will demonstrate a new concept of realizing reconfigurable large-signal microwave circuits based on GaN technology, which will be used for the first time as active varactors and switches co-integrated in low cost microstrip technology to achieve the tunability of circuits.

The proposed specifics objectives are:

1. Explore possibilities of emerging GaN technology for integrated circuits that are miniaturized such as phase shifter and frequency reconfigurable hybrid couplers where the control functionality is achieved by using varactor diodes.

2. Investigate possibilities for RF reconfigurable circuits using active GaN components with a low-cost substrate. For this a reconfigurable oscillator, a beam switching antenna array and a matching reconfigurable patch antenna operating below 10 GHz were designed.

3. Demonstrate a working prototype that combines active GaN devices and passive components so as to experimentally validate the proposed concepts.

It will be shown that the developed GaN varactor and switch models can be used along with passive circuit designs in low cost microstrip technology so as to achieve, in the same circuit, reconfigurability and good RF performance. Additionally, this research project will demonstrate the great potential of GaN for use in other tunable microwave high power applications such as oscillators and reconfigurable antennas.

III. Thesis Organisation

The thesis begins with the chapter II dealing with a review of gallium nitride technology. Then, chapter III presents the characterization and modeling of gallium nitride varactor and switch devices manufactured with NRC process. The final chapter IV covers contributions and realized applications over this thesis research.

Chapter II [START_REF] Kolias | Recent advances in Ga N MMIC technology[END_REF]. In the early 1990s, GaN was revealed as a potential solution to satisfy the simultaneous requirements of high frequency and power [START_REF] Runton | History of GaN: High-Power RF Gallium Nitride (GaN) from Infancy to Manufacturable Process and Beyond[END_REF]. GaN material has the capability of satisfying system device requirements from low-frequencies [START_REF] Saito | 600V AlGaN/GaN power-HEMT: design, fabrication and demonstration on high voltage DC-DC converter[END_REF]- [START_REF] Delaine | High frequency DC-DC converter using GaN device[END_REF] through millimeter-waves [9]-[10]- [START_REF] Micovic | W-band GaN MMIC with 842 mW output power at 88 GHz[END_REF]- [START_REF] Milligan | SiC and GaN Wide Bandgap Device Technology Overview[END_REF] and was deemed an excellent material for being useful in high power applications. Because of its special properties, GaN technology is truly recognized as a revolutionary wide band gap technology resulting in dramatic afforded power levels that exceed many times (> 5 times) those levels achieved with other traditional technologies such as, indium phosphide (InP), gallium arsenide (GaAs), and silicon (Si). As a result, current densities in excess of 2 A/mm have been reported [START_REF] Saito | High breakdown voltage AlGaN-GaN power-HEMT design and high current density switching behavior[END_REF]- [START_REF] Zimmermann | AlN/GaN Insulated-Gate HEMTs With 2.3 A/mm Output Current and 480 mS/mm Transconductance[END_REF]. High power densities of 40 W/mm have also been reported at 4 GHz with 1.2 A/mm by Cree using field-plated devices [START_REF] Wu | 40-W/mm Double Field-plated GaN HEMTs[END_REF]. Watt-level output power has been demonstrated and reported at W-band in [START_REF] Brown | W-band GaN power amplifier MMICs[END_REF]- [START_REF] Nakasha | E-Band 85-mW Oscillator and 1.3-W Amplifier ICs Using[END_REF]- [START_REF] Micovic | W-band GaN MMIC with 842 mW output power at 88 GHz[END_REF] by HRL, Raytheon and Fujitsu. In [START_REF] Milligan | SiC and GaN Wide Bandgap Device Technology Overview[END_REF] Cree has demonstrated power density higher than 25 W/mm of gate periphery with unity current gain frequencies, Ft, greater than 40 GHz reported. As GaN offers very high breakdown field, high electron mobility, and saturation velocity. It is also known that GaN has an excellent thermal conductivity and ability to operate at high temperatures which make it an ideal candidate for highpower and high-temperature microwave applications [START_REF] Runton | History of GaN: High-Power RF Gallium Nitride (GaN) from Infancy to Manufacturable Process and Beyond[END_REF] All of these presented factors indirectly improve the efficiency of GaN devices. Furthermore, possible epitaxial growth on SiC substrate, which has excellent thermal properties, makes this device optimal to operate at high power. Later in this chapter, the possibility of growing GaN over other substrates will be more detailed.

--------------------------------------------------------------------------------------------------------------------- A Review
However, despite the superior characteristics of GaN device, there still remain some issues at multiple levels with respect to fabrication (process, device, circuit and system) that must be overcome before being widely adopted in future.

These challenges that are impacting the development of GaN technology can be summarized in two main challenges such as technological (device processing) and commercial tasks.

The first technological (device processing) challenge is that the GaN HEMTs are affected by two important phenomena, namely the trapping effect and thermal effect (or self-heating), which can directly cause current dispersion such as Most reported works, explaining this effect, conclude that trapping effect can mainly reside at the surface [START_REF] Vetury | The impact of surface states on the DC and RF characteristics of AlGaN/GaN HFETs[END_REF]- [START_REF] Mitrofanov | Mechanisms of gate lag in GaN/AlGaN/GaN high electron mobility transistors[END_REF], in the AlGaN barrier layer, at the twodimensional electron gas (2DEG) interface, and in the GaN buffer layer [START_REF] Klein | Photoionization spectroscopy of deep defects responsible for current collapse in nitride-based field effect transistors[END_REF]- [START_REF] Faqir | Mechanisms of RF Current Collapse in AlGaN #x2013;GaN High Electron Mobility Transistors[END_REF]. This effect is produced when a large bias voltage is applied between the source and drain, which induces a quick acceleration of the electrons in the conducting channel. These fast random moving electrons can be injected into an adjacent region of the device structure and became trapped if this region contains a significant concentration of traps which is the case for the GaN buffer layer.

On the other hand, thermal and self-heating effects exist in every power semiconductor device. Especially when the device is working in the high voltage and high power region, the self-heating causes electron carriers to accelerate in random directions instead of following the drain to source channel. Some of the electrons eject  Barrier: This semiconductor layer isolates the gate and the channel so that very low current flows between the channel and the gate. In GaN FETs, the barrier is typically made of aluminum gallium nitride (AlGaN).

 Channel: The channel or 2-DEG is the region where electrons flow from the source to the drain. The channel is typically made of high-quality GaN.

 Buffer: The role of the buffer is to restrict the movement of the electrons within the channel. In other words, the buffer acts as a barrier that prevents the electrons from wandering into the substrate. In a GaN FET, the buffer is typically made of GaN doped with carbon (C) or iron (Fe).

 Nucleation: The growth of GaN structures on a foreign substrate requires the careful matching of the GaN lattice to this substrate, involving the engineering of suitable nucleation layers between the substrate and the buffer layer.

 Substrate: The substrate provides mechanical support, heat spreading, and electromagnetic confinement. GaN FETs use a foreign substrate (that is, Sapphire, Si or (SiC), but not GaN). Because the substrate is not GaN, it has a different crystal lattice than the buffer. This creates dislocations in the material and ultimately reduces the electrical isolation between gate and channel.

Even though there have been remarkable improvements in growth and device fabrication using GaN material, trapping effects is still a challenge and cannot be eliminated perfectly, which still an active key research topic in AlGaN/GaN HEMT applications. Additionally, an accurate HEMT model including trapping effects is still needed for development of implementation of AlGaN/GaN HEMT-based circuits.

The second aspect of GaN HEMTs that is a significant disadvantage is the cost of fabrication. Reducing the production cost is an important factor for power GaN HEMTs commercialization. This big commercial challenge is due to a general lack of availability. There are more and more companies over the world develop GaN devices, so it is possible to buy some GaN HEMTs from distributors today, but the selection is limited and still expensive compared to others technologies. Additionally, the lack of device standardization between these distributors means that there are no real second sources for any of the devices on the market today. No major manufacturer is going to use a GaN transistor in a high-volume application until interchangeable devices are available from multiple manufacturers. Furthermore, the unavailability of bulk GaN source material led to the need for GaN growth on mismatched substrates such as Si, SiC and sapphire, this also poses an issue which leads to poor quality epitaxial films with dislocation densities.

Choice of the Substrate

Choosing an appropriate substrate to grow GaN material is still an issue for developing GaN circuits. Beside Si or GaAs circuits for example, a Si or GaAs substrate is normally used; while GaN circuits have to be grown over another substrate because it is difficult to manufacture a GaN substrate due to the lack of the material. Since GaN is not easy found in nature, it must be grown on another substrate, typically Si, SiC or Sapphire. There are three techniques to grow GaN over a mismatched substrate. These techniques have been employed widely in the past to reduce the high dislocation density and for the realization of GaN substrate with large size and high quality. One of the common methods is done by using metal organic chemical vapour deposition (MOCVD), after which the substrate is removed [START_REF] Nakamura | Continuous-wave operation of InGaN/GaN/AlGaN based laser diodes grown on GaN substrates[END_REF]. The success of this technique is resided in reducing dislocation density [START_REF] Nakamura | Continuous-wave operation of InGaN/GaN/AlGaN based laser diodes grown on GaN substrates[END_REF]; however it leads to selective area epitaxy due in large part to mechanical stresses and mechanical deformations, thus resulting on a limitation of GaN wafer. Typically GaN wafer are limited to two or three inches in diameter and thicknesses of about 100μm.

A second way to grow GaN, which is currently the most commonly used technique, is by hydrid vapour phase epitaxy (HVPE) [START_REF] Paskova | GaN Substrates for III-Nitride Devices[END_REF] which is initially used in the growth of GaAs and InP. This is faster and has lower defect density than MOCVD (5x10 4 /cm 2 for HVPE, compared to 8x10 7 /cm 2 for MOCVD). Additionally, it has been demonstrated that larger wafer sizes can be realized [START_REF] Hageman | Thick GaN layers grown by hydride vaporphase epitaxy: hetero-versus homo-epitaxy[END_REF]. A third method for growing GaN is by ammonothermal growth [START_REF] Dwilinski | Bulk ammonothermal GaN[END_REF]. This technique offers even better defect densities (1x10 3 /cm 2 ). However, growth of the crystal is much slower: 0.4-16 μm/h [START_REF] Hashimoto | Improvement of crystal quality in ammonothermal growth of bulk GaN[END_REF], compared to 100-500 μm/h for HVPE [START_REF] Łucznik | Deposition of thick GaN layers by HVPE on the pressure grown GaN substrates[END_REF].

While GaN substrate is still unused, the decision of choosing a foreign substrate is crucial and based on several factors such as lattice constant, thermal conductivity, thermal expansion coefficient, wafer size and, of course, cost. The lattice parameter and the coefficient of thermal conductivity are the two crucial important properties that should closely be matched between the GaN and the mismatched substrate. Any mismatch between these two properties can result in defects in the film such as misfit and threading dislocations due to lattice mismatch and cracking or bowing due to the thermal mismatch. Table II.2 summarizes these parameters that are used when a substrate is selected. [START_REF] Tang | The ammonia-MBE approach for high electron mobility in GaN epilayers and AlGaN/GaN HFET structures[END_REF]. Currently both standard processes, GaN500 and GaN150, use 3-inch diameter SiC wafers of 75 μm thickness. More details about the wafer growth can be found in [START_REF] Tang | The ammonia-MBE approach for high electron mobility in GaN epilayers and AlGaN/GaN HFET structures[END_REF]- [START_REF] Bardwell | AlGaN/GaN Transistors and Circuits[END_REF].

As shown in Figure II.5, where the complete process sequence is given, the first step in the GaN500 process fabrication as well as in GaN150 process is to deposit the Ohmic contact using the Ohmic mask (OHMIC Layer). The contacts are then annealed at a high temperature to ensure that a low-resistance contact is made to the two-dimensional electron gas (2DEG), where the carriers are located. The next step is to define the active area of the devices using the mesa mask (MESA Layer), and then it is isolated by etching it down to the buffer layer.

Gate metals are then formed by depositing a passivation dielectric layer in the channel area and etching a slot for the gate contact. The gate contact metal (GATEMETAL layer) is deposited and a second dielectric encapsulation is applied.

Openings are etched into the dielectric (VIA1 layer), where required, to allow connection to the ohmic and gate contacts.

Then, a thin layer of nichrome (NICHROME Layer) is deposited where resistors are required. The first level of interconnect metal (1 μm of gold) is next deposited using the 1ME mask (1ME Layer). The 1ME provides contacts to the ohmic metal and gate pads, as well as the nichrome layer. This layer is also used to form a field plate, which is connected to the source and overlaps the gate.

Next, the 1ME layer is isolated with a dielectric deposition and via openings are defined (VIA2 Layer) where required to define the vias. This layer of dielectric is used as the insulator in the MIM capacitors.

Finally, the areas where a second level of interconnect is defined (2ME Layer) to bridge over other layers is defined with a temporary layer of a special resist using the bridge mask (BRIDGE Layer). The 2ME layer is usually 1 μm of gold and is used to make the top layer of the MIM capacitors.

These process steps and GaN500v2 offers transistors with a unity current gain frequency, Ft, of 13

GHz and a maximum oscillation frequency, Fmax, of 60 GHz, and it is suitable for 30V maximum drain voltage bias, and yields power levels of ~7 W/mm (measured at 8

GHz). While GaN150v1 offers transistors with Ft of 35 GHz and a Fmax of 75 GHz.

Furthermore, GaN500v2 HFETs devices exhibit breakdown voltages (measured with channel pinched-off) generally greater than 200V, however a maximum drain voltage of 40V is currently recommended. While a breakdown voltage VDS_BD (VDS measured at channel pinched-off) greater than 100V and a maximum drain bias voltage of 30V is currently recommended for GaN150v1 devices. Finally, the gate source voltage can operate from -8V to 2V, with at least a threshold voltage of -4.6V needed to turn these devices ON [START_REF]Gallium Nitride MMIC Foundry Design Manual GaN500v2[END_REF]. Additional details about the technology are provided in NRC's GaN design manual process [START_REF]Gallium Nitride MMIC Foundry Design Manual GaN500v2[END_REF].

As is traditionally known, as a part of any technology fabrication process, proper and accurate models are usually set and provided in the process design kit (PDK) to achieve any circuit's functionality. In the used GaN fabrication process, there is still work to be done as it lacks essential components desired for reconfigurable applications, such as the lack of varactor models, and also the lack of common gate (CG) configured switch models, which are vital for revealing switch operation capabilities of these FETs. Moreover, the current version of the PDK only supports a limited number of active device sizes. This thesis will concentrate on proposing equivalent circuit models, firstly, for varactor diodes and then for switch in common gate configuration. Equivalent circuit models analysis for varactor diodes will be conducted in small signal and large signal while only small signal analysis will be conducted for switch devices. These developed models will be used in RF designs to verify them and validate their abilities of use in some RF applications.

III. Use of GaN in Reconfigurable RF/Microwave

Systems

Monolithic RF/microwaves integrated active reconfigurable systems are necessary for applications which require lightweight, low cost, high reliability, ease of integration, and high volume. The recent development of high electron mobility transistor (HEMT) makes it possible to realize a high level reconfigurable RF/microwaves design, such as power amplifiers, phase shifters, low noise oscillators, and reconfigurable antenna array designs. The high efficiency power amplifier and the phase shifting constitute the important parts of a RF/microwaves transmitter block.

Due to the mature development of GaAs technology, beam-former circuitry could be integrated into the same package as the antenna array system [START_REF] Edwards | Semiconductor Technology Trends for Phased Array Antenna Power Amplifiers[END_REF]. There have been intensive developments of these listed components, which have relied on high performance semiconductor technologies especially GaAs HEMT technology. In order to achieve higher RF output power with higher efficiency, GaN technology is a very promising candidate for realizing designs, such as high efficiency tunable circuits, highly reconfigurable low noise power oscillators and integrated active beam forming transmit antenna arrays [START_REF] Ohira | Megalithic microwave signal processing for phased-array beamforming and steering[END_REF]. The current studies of this technology reveal that GaN has the potential to be the technology of choice for the next generation of transmitter/receiver (Tx/Rx) modules where high power/frequency applications [START_REF] Eastman | The toughest transistor yet [GaN transistors[END_REF]- [START_REF] Heying | Reliable GaN HEMTS for high frequency applications[END_REF] and several functionalities on the same chip [START_REF] Oppermann | GaN/SiC MMICs and packaging for use in future transmit / receive modules[END_REF] are required.

Active reconfigurable circuitry for transmitting and/or receiving (Tx/Rx) modes has been widely used for many applications, such as space and defense. In addition, high efficiency compact Tx/Rx modules are increasly needed in RF/microwaves applications. Their development is driving the enhancement of HEMT MMIC technology, now extending to the promising GaN technology. Indeed, the trend in the last generation of Tx/Rx module systems is to reduce the number of MMICs in the module by increasing their power handling capabilities and efficiencies using GaN [START_REF] Lin | Application of GaN high power chips in T/R modules[END_REF]- [START_REF] Kolias | The development of T/R modules for radar applications[END_REF]. The current state of art shows that GaAs technology has been intensively used for these systems at many frequency bands [START_REF] Yukawa | Q-band gaas MMIC modules for active phased array antenna systems[END_REF]- [START_REF] Baccello | Miniaturised low cost solid state 4W TXRX common leg for 6 #x00F7;18 GHz phased array[END_REF]. Meanwhile for GaN technology, many reconfigurable power amplifiers for Tx/Rx modules have been reported in the last ten years. There has been some work on Tx/Rx module using GaN fox X-band [START_REF] Schuh | GaN MMIC based T/R-Module Front-End for X-Band Applications[END_REF], where three types of amplifiers (DA, HPA and LNA) were integrated on multi-layer low temperature co-fired ceramic (LTCC) substrates. In Another Tx/Rx module using GaN was reported [START_REF] Masuda | C-Ku band GaN MMIC T/R frontend module using multilayer ceramics technology[END_REF], which contains a GaN power amplifier (PA) with output power higher than 10 W over 6-18 GHz and a GaN In [START_REF] Jantarachote | Frequency band and operating class reconfigurable GaN HEMT power amplifier for sustainable wireless communications and energy applications[END_REF] using tunable capacitors in the input and output matching networks, a 2.45 GHz GaN HEMT operating class and frequency reconfigurable power amplifier is reported. More studies revealed that due to the drawbacks of GaAs MMIC HPA as low efficiency and limited output power level have made it not to be a competent candidate of power chips against the high efficiency of GaN MMICs technology [START_REF] Lin | Application of GaN high power chips in T/R modules[END_REF].

In this same reference, a GaN-based high power amplifier with output power of 85 W and operating at S-band is reported. In 2014, an investigation [START_REF] Oppermann | GaN/SiC MMICs and packaging for use in future transmit / receive modules[END_REF] revealed that the next generation of GaN MMICs will cover more functionalities on the same chip.

Multifunctional MMICs will allow very compact Tx/Rx module designs, and will reduce the number of active devices [START_REF] Oppermann | GaN/SiC MMICs and packaging for use in future transmit / receive modules[END_REF].

In this context, this thesis research is focused to demonstrate the use of GaN on highly efficient reconfigurable designs for broadband RF/microwave applications operating below 10 GHz. GaN components such as varactors and switches used for tunable purpose are firstly fabricated, characterized and then modeled. The proposed models are used to achieve the reconfigurability of RF/microwave circuits. Tunable phase shifter, reconfigurable 3-dB 90° hybrid coupler, tunable frequency oscillator, beam switching antenna array and matching reconfigurable patch antenna are developed along with the development of GaN modeling varactors and switches, and are presented as applications results of this thesis research. A current literature review of these reconfigurable cited circuits will be presented along with the realization of these designs later in chapter IV.

IV. Summary

In this chapter the properties of GaN technology have been reviewed, discussed and compared with the commonly used semiconductor technologies. It has been clearly shown that GaN technology is a great promising semiconductor candidate for applications that high power and high frequency are required as well as high temperature. Its disadvantages and difficulties are also studied presented and discussed. Additionally the features and limitations of the used AlGaN/GaN process throughout this thesis provided by NRC were summarized and presented.

Furthermore, an overview of the ability to use GaN technology for tunable purposes such as power amplifier discussed. We also reviewed some previous work done on the development of reconfigurable power amplifier circuits using GaN technology that are used in Tx/Rx modules.

Chapter III

---------------------------------------------------------------------------------------------------------------------

GaN Varactor and Switch Devices Modeling I. Introduction

A general purpose reconfigurable integrated RF/microwave circuit could be realized using a promising semiconductor technology for the active device, and an interconnection technique such as wire-bonding technique to compact all the circuits.

Recent technology trends favor greater integration of electronic devices and radiating elements, greater functionality and reconfigurability (on a single chip or via multichips), as well as high operating frequency/power with excellent efficiency and linearity. Such are the needs for next generation of Tx/Rx modules.

For the proposed thesis research, GaN-based varactor and switch components will be pursued; the main features of this type of semiconductor technology being its high power with high efficiency. Compact reconfigurable circuits, such as tunable phase shifter, reconfigurable 3-dB 90° hybrid coupler, tunable frequency oscillator, beam switching antenna array and matching reconfigurable patch antenna based on these developed varactors and switches will be studied. The first part of the work is devoted to active GaN device development and characterization. The second part addresses the integration and design aspects of the reconfigurable proposed circuits.

GaN technology to date has seen very little application to signal control devices such as varactors and switches. These two elements are key points to realize phase shifters (both continuous and discrete state), frequency reconfigurable circuits, and signal path selection circuits for beam switching or Tx/Rx selection.

Our investigation considers modeling in small-and large-signal of both

Applications

GaN HEMTs semiconductor devices are no longer limited to RF power amplifier applications, but having been extended to other applications such as LNAs [START_REF] Pengelly | Commercial GaN Devices for Switching and Low Noise Applications[END_REF], VCOs [START_REF] Lan | A V-Band Monolithic AlGaN/GaN VCO[END_REF], mixers [START_REF] Andersson | C-band linear resistive wide bandgap FET mixers[END_REF], and phase shifters [50]- [START_REF] Jin | A GaN Schottky Diode-based analog phase shifter MMIC[END_REF]. Furthermore, reconfigurable RF circuits, with high power/frequency, and excellent linearity, are in great demand.

Varactors are key components in the design of these reconfigurable circuits [START_REF] Marso | MSM diodes based on an AlGaN/GaN HEMT layer structure for varactor and photodiode application[END_REF] for microwave and millimeter wave applications. The varactor diodes can be fabricated from either p-n or Schottky junctions and both cases are under research today.

The superior characteristics (i.e high-temperature operation due to large energy band gaps, high breakdown field, high saturation velocities of electrons, etc) make GaN heterostructures promising candidates for realizing high power/tuning In [START_REF] Jin | A novel GaN-based high frequency varactor diode[END_REF], a novel GaN-based high frequency varactor diode was reported.

This main goal of this work is to investigate the high frequency performances Based on Schottky barrier, another work was reported in [START_REF] Lu | InGaN/GaN Schottky Diodes With Enhanced Voltage Handling Capability for Varactor Applications[END_REF]. In this work, a design of InGaN/GaN schottky diodes with enhanced voltage handling capability with vertical geometry was discussed. It is, therefore, illustrated that a higher voltage handling capability may be obtained with a suitable InGaN surface layer design.

44

The reported work shows that varactor diode made in GaN material are capable of bias voltages over 120V at a doping level of 1 ×10 17 cm -3 over a 1-μm-thick depletion region. Furthermore, at 1 GHz, a significantly high Q value superior of 35 was reported which corresponds to series resistance of this device (∼8 Ω) which was primarily from the underlying n + GaN and contact resistance. Two years later, on 2012, the same group researchers of [START_REF] Lu | InGaN/GaN Schottky Diodes With Enhanced Voltage Handling Capability for Varactor Applications[END_REF] reports a new work [START_REF] Lu | Critical design considerations for GaN-based microwave power varactors[END_REF] where the critical design considerations for achieving GaN-based high voltage, high Q factor and high linearity microwave varactors were summarized and were discussed. In addition, detailed performances for a real fabricated GaN-based anti-parallel varactor diode structure that simultaneously achieving high Q, highbreakdown voltage and high linearity were presented. This investigation was based on using the same GaN structure presented in Figure III.11. It was highlighted that heavily doped semiconductor material is required to minimize the series resistor and achieve a high Q-factor. However, in order to achieve high breakdown voltage, the semiconductor needs to be lightly doped to keep the electric field in the semiconductor below its critical electric field. Furthermore, the two anti-parallel diodes structure was used to investigate the varactor linearity. It was reported that by using this structure, the RF signal induced capacitance variation is minimized. The linearity of the varactor diode is thus significantly improved.

Realized GaN HEMT-based Varactor Diodes

These previous reported works present good performance in term of high breakdown voltage, high Q-factor, and high linearity; however, the limited capacitance tuning range is still a challenge. Furthermore, to achieve good performance, most of these reported works require some changes in epitaxial and mask layer of the GaN HEMT process which results in a complexity of fabrication and cost increasing. These challenges have limited the progress of developing varactor diodes. In order to overcome the two above-mentioned drawbacks, here, for this thesis work, the varactor diode is realized by connecting together the drain and source terminations thus resulting in a single Schottky junction which can be considered as planar schottky varactor diode (PSVD) [START_REF] Lucyszyn | Accurate millimeter-wave large signal modeling of planar Schottky varactor diodes[END_REF].

The bias potential VG is then applied across the drain/source and gate terminations. This kind of varactor diodes do not require any additional selective changing in the epitaxial layers of the HEMT process. As a result, a standard GaN HEMT process can be used which minimize their cost production compared to the other reported type of varactor diodes. However, GaN-based PSVDs have not been widely used because of the unavailability of accurate large and small signal models which become our motivations to develop them. We investigate the equivalent capacitance CEq and the equivalent series resistance REq of the varactor, when the overall impedance is considered as a simple series RC circuit. the experimental results and the simulation model results is generally achieved. For GaN150v0, agreement is nearly perfect, while for GaN500v1 there are some issues with the larger device at high frequencies for -6 V bias. Here the channel is fully depleted and some parasitic inductances are not captured in the equivalent model.

GaN500v1 and GaN150v0 Varactor Devices

For VG of -3 V, -4 V and -6 V, the parameter values of the equivalent model are presented in Table III Thus the REq-VG characteristics are taken to be:

Equation (4) 𝑅𝑒𝑞𝐺𝑎𝑁500 = 44 A [ ( 1 √ 𝑊𝜋 Y 8. 10 -5 ) (𝑒 -2(𝑉𝑔+4.35) 2 ) + 𝑌] Equation (5) 𝑅𝑒𝑞𝐺𝑎𝑁150 = [ ( 1 6 √ 𝐴𝜋 Y 8. 10 -5 ) (𝑒 -2(𝑉𝑔+4.35) 2 ) + 𝑌]
Where A is the finger area. The parameter Y can be adjusted to properly fit the REq equation to the measured characteristics of an arbitrary varactor size.

According to Matlab simulations, the value Y can be fixed to 14 for GaN500v1 devices and as 4.75 for GaN150 devices. All of these equations are dependent only on finger geometry (number, width and length of fingers) and on VG. Equation (4) describes REq for GaN500v1 devices, while Equation ( 5) describes REq for GaN150v0 devices. These equations describe the experimental results adequately under small signal excitation. Due to these resistance values, a part of the RF available power is consumed by the devices and losses will occur in the RF signal. More REq is high more losses are high. A good agreement is obtained, except for REq in range (1) of the GaN150v0 devices. A slightly more complicated model would be required here to capture the higher measured REq.

Analysis of Q-Factor

In The Q factor decreases with frequency. It is notable that a minimum Q of ~25 and ~50 was achieved over a range of frequency below 2 GHz, respectively, for GaN500v1 and GaN150v0. Another most common used figure of merit for varactor is the cut-off frequency. The cut-off frequency is defined as the frequency at which the capacitive reactance is equal to the series resistance. Note that the cut-off frequency is also a function of voltage and that it achieves its maximum value at VB. This FOM indicates that the maximum operating frequency of the circuit is dependent on the series resistance and the equivalent capacitance. FOM should be as high as possible, which can be mathematically given as: Equation ( 7)

𝐹𝑐𝑢𝑡_𝑜𝑓𝑓 = 1 (2. 𝜋. 𝐶𝑒𝑞. 𝑅𝑒𝑞)
At f = 2.5 GHz, maximum values of 164.9 GHz and of 419.2 GHz were achieved, respectively, for W40L0.5 and W100L0.5 at VG = -5 V. While for W50L0.15 and W100L0.15, the maximum obtained values are 770.6 GHz and 335.9 GHz, respectively. These obtained values serve as figure of merit to compare with other varactor diode in terms of frequency operation capability. A comparison between the two GaN processes in terms of capacitor CEq is presented in Table III

GaN500v2 Varactor Devices

To conduct this second varactors modeling study, many samples with different sizes have been fabricated with the GaN500v2 process but only three of them are presented here (see Table III.4). The presented devices are W50L0.5, W100L0.5 and W200L0.5 (different fingers width).

Small Signal Measurement

The on-wafer RF measurements were performed with an Agilent VNA Network Analyzer over a frequency range of 0.1-20 GHz with a fixed RF power of -15dBm, while the control voltage at the gate of varactors, VG, was swept from 0 V to -10 V with a step of -0.1 V. Because it was observed that from 0 V to -3 V, the measured results are similar and also from -5 V to -10 V measured results are similar.

So only measurements from -2 V to -6 V will be treated and analysed here. III.5 summarizes the parameter values of the equivalent model at three bias voltage, such as, VG of -3 V (in range 1), -4 V (in range 2) and -6 V (in range 3). This means that at very high frequency, the parasitic gate inductance, Lg, and other effects become self-resonant. and rapidly over a small DC bias range of 1 V. This behavior is also observed for the devices using the GaN500v2.

Because we are essentially focused on the range [START_REF] Ross | Gallium nitride hybrid microwave circuits for low-noise applications[END_REF] where capacitance values change significantly, the x-axis of all C-V plots is delimited from -2 V to -6 V.

Values from 0 V to -2 corresponding to CMAX are stable as well as ones from -6 V to -10 V corresponding to CMIN. samples and from GaN500v2 samples, it was revealed that the achieved capacitance values from a device size manufactured with GaN500v2 are at least twice larger than those achieved with the same device size fabricated with GaN500v1 process.

However, the overall calculated CMAX/CMIN ratios for GaN500v2 devices are smaller (~1.7) than ones calculated from GaN500V1 devices (~2.4). This significant change in equivalent capacitance values is essentially due to the additional field plate layer added in GaN500v2. As discussed in the GaN500 design manual provided by NRC, it should be pointed out that all devices included in GaN500v1, do not employ field plate designs. However, the shape of the gates, using GaN500v1, results in better frequency performance, but slightly lower breakdown voltage compared to GaN500v2. For GaN500v1, values of 27 GHz and 75 GHz were reported by NRC, respectively, for fT and for fmax. The process is suitable for 20V maximum drain voltage bias. While for GaN500v2 process, a field plated design is included in this version, and devices exhibit much lower gate leakage than in GaN500v1. However, RF performance frequency decreased; a fT of 12 GHz and fmax of 24 values were measured and reported for this version. The process version is suitable for 40V maximum drain voltage, and yields power levels of ~5 W/mm. We assume that this additional field plate layer is the main parameter behind the significant increase in the equivalent capacitance values.

Analysis of Equivalent Series Resistance

The equivalent series resistor extracted from measurement versus bias voltage is depicted in Figure III.36 for all studied devices. This series resistor is not impacted by the frequency, so these extracted values are valid at all frequencies.

Here the same general behaviour observed before for GaN500v1 is also noticed for the devices fabricated with GaN500v2, expect that the values of the series resistor are significantly different from a process version to another one.

Unlike the GaN500v1 devices where an analytic expression was introduced to describe the behavior of REq versus VG, here for GaN500v2 devices, as can be clearly seen the equivalent series resistor values do not change significantly and this is observed for all devices, so a mean value can be adopted as a fixed value to describe the REq for a specific device size. In addition, it was observed than more the device size increases more the REq decreases. The maximum peak value of REq is observed on range (2) when the channel starts depleting because the depletion resistor (Rdep) becomes slightly important and increases until the channel is close to be fully depleted. This behavior has been also observed for GaN500v1 process' devices. It should be pointed out the overall series resistor values extracted from GaN500v2 devices are smaller than ones found with the previous process version (GaN500v1).

In our point of view, this significant drop in series resistor can be due to the development occurred out on the stack of epitaxial layers and masks of the process to reduce the dynamic on-resistance and the Ohmic contact resistance that are constituting the series equivalent resistor (REq).

As mentioned in chapter II, the trapping electrons phenomena in AlGaN/GaN HEMTs structure is directly reducing the sheet electron density in the channel, leading to increase dynamic Ron and provoking drain current dispersion.

Moreover, many efforts for high performance AlGaN/GaN HEMTs have been made to suppress this current dispersion issue and increased Ron, which are directly related to modification of surface/interface states in AlGaN/GaN heterostructures.

These modifications are surface passivation [START_REF] Arulkumaran | Surface passivation effects on AlGaN/GaN high-electron-mobility transistors with SiO2, Si3N4, and silicon oxynitride[END_REF], a thin AlN barrier layer in AlGaN/GaN interface [START_REF] Lee | Reduction of current collapse in AlGaN/GaN HFETs using AlN interfacial layer[END_REF], application of a field plate [START_REF] Wu | 30-W/mm GaN HEMTs by field plate optimization[END_REF], surface treatments with chemicals or plasma [START_REF] Edwards | Improved reliability of AlGaN-GaN HEMTs using an NH3 plasma treatment prior to SiN passivation[END_REF], and postgate-annealing [START_REF] Miura | Thermal annealing effects on Ni/Au based Schottky contacts on n-GaN and AlGaN/GaN with insertion of high work function metal[END_REF]. On the other hand, in spite of the rapid development and the studies of GaN-based HFETs, achieving a low Ohmic contact resistance value still one important issue concerning these devices [START_REF] Takei | Reduction of contact resistance on AlGaN/GaN HEMT structures introducing uneven AlGaN layers[END_REF]- [START_REF] Okamoto | An ohmic contact process for AlGaN/GaN structures using TiSi2 electrodes[END_REF] and has a great effect on the output signal during device operation. Low Ohmic contact resistance is crucial to realization of good RF and power performances of AlGaN/GaN HEMTs heterojunction devices. As a matter of fact, in terms of power consumption, high Ohmic contact resistance values cause high electrical power losses and increase self-heating problem. For example, if the contact resistor is decreased from 10 -5 to 10 -6 Ωcm 2 , a reduction of 60% of the power consumption could be achieved for devices with breakdown voltage of 600 V [START_REF] Saito | Theoretical limit estimation of lateral wide band-gap semiconductor power-switching device[END_REF].

Due to the restricted information, we could not have information on how NRC group researchers have improved GaN500v2 process to reduce the Ohmic contact resistance or/and Ron. However, it is reported, by NRC in design manual of GaN500v2 [START_REF]Gallium Nitride MMIC Foundry Design Manual GaN500v2[END_REF], that a field plate is applied to all devices fabricated with GaN500v2.

Large Signal Measurements

As we have just presented, the small signal analysis leading to propose a circuit model has been discussed. In addition, analytical expressions describing the general behavior of CEQ and REq for several DC bias were introduced. Following our original modeling and analysis of GaN-based varactor diodes, by using only a standard GaN HEMTs process; here, the effect of larger power signal is studied and presented to complete the analysis of GaN varactor small and large signal modeling.

Unlike to GaN500v1 devices, the newly developed version of GaN500v2 is used for manufacturing the samples which have been measured in large signal measurements.

Here the measurement calibration has been done at a fundamental to around 8 GHz, and because one of these studied varactor will be used for frequency reconfigurable antenna around 7 GHz. For this, large signal analysis will be conducted around 5 GHz and we assume that this analysis is valid for all frequencies below 8 GHz. In addition, the RF power was swept from -15 dBm to 18 dBm (maximum output power from the source) while the VG was also swept from -3 V to -6 with a step of -0.2 V.

The aim of this section is to propose an accurate analytical large signal model for GaN varactor diodes that could be used as general model, which offers the ability to describe the behavior of the device at high power levels and for various bias points. The large signal analysis of the devices, which includes the impact of RF power on the CEQ and REQ values is established and explained. In addition, the nonlinear model behavior of CEQ-VG characteristics is described by a simple analytical expression based on empirical coefficients which are derived from the large signal measurements. This developed analytical equation is valid for a linear size-scaling and is introduced for the voltage and RF power dependency of CEQ.

Analysis of Equivalent Capacitance

In order to describe the nonlinear behavior of CEq impacted by the injected RF power, an analytical equation based on empirical coefficients is derived from the measurements. This developed equation (see Equation ( 8)) is based on the one (Equation ( 3)) proposed before to describe CEQ-VG curve for small signal which was only VG dependency. However, the influence of RF power has been taken into account in Equation ( 8) to approximately describe CEq versus VG for various power levels. It is seen that Equation (8) accurately describes the dependency of junction capacitance upon applied reverse bias as well as upon RF power for all characterized devices. The CEq-VG curve can be expressed by the following analytic equation:

Equation (8) 𝐶𝑒𝑞 = ( 𝐵 𝐶 𝐴)[(1 + 1 𝐶 ) (𝑒 𝑡𝑎𝑛ℎ[(𝑉𝑔+𝐹)(𝐺-𝐻𝑃𝑖𝑛)+𝐼] ) + 𝑘]

Summary

The varactor devices structure studied here is based on NRC GaN processes. Three GaN HEMTs process, such as, GaN150v0 with a gate length of 0.15 um, GaN500v1 with gate length of 0.5 um and GaN500v2 which is the developed version of GaN500v1, have been used for manufacturing the varactors diodes.

Small signal analysis has been done for all fabricated devices. Equivalent capacitance and series resistor behaviors have been detailed, and it was revealed that all CEq versus VG exhibit the same general behavior for all devices. In addition, the capacitance tuning range is limited, and the region of variable capacitance exists only over a small reverse voltage of 1V. Furthermore, the capacitance value depends on the finger area. This means that larger is the device, higher are the capacitance values. Moreover, it was observed that the extracted capacitance values from varactors with GaN500v2 are at least twice larger than those extracted with the same device size fabricated with GaN500v1 process. This is due to the additional field plate layer in GaN500v2 process. It was, also, observed that the devices manufactured with The fabricated switch circuits will be integrated with antenna system to realize and to demonstrate the ability of using GaN devices for a steering beam antenna array that will be presented later in chapter IV.

As a promising high power handling alternative, GaN technology is expected to be a good candidate for the realization of high power HEMT-based switches [START_REF] Ciccognani | High isolation microstrip GaN-HEMT Single-FET Switch[END_REF]- [START_REF] Bettidi | High power GaN-HEMT SPDT switches for microwave applications[END_REF]. The HEMTs suitability for switch usage stems from the fact that the RF path between source and drain is considered as a voltage-controlled resistor [START_REF] Devlin | The design of integrated switches and phase shifters[END_REF]. The DC bias applied to the gate controls the two states of switching "ON and OFF states", with no DC bias applied to source or drain. In addition to this controlledresistance there is also a capacitance associated with the HEMT structure. In other words, the equivalent circuit of a HEMT-based switch could be a simple variable They demonstrated an extremely linear power handling of 43 W (10 times higher than that of typical GaAs-based switch ICs) at 1 GHz with excellent performance in terms of insertion loss (-0.26 dB) and isolation (-27 dB) [START_REF] Ishida | A high power Tx/Rx switch IC using AlGaN/GaN HFETs[END_REF]- [START_REF] Ishida | A high-power RF switch IC using AlGaN/GaN HFETs with single-stage configuration[END_REF]. Another AlGaN/GaN HEMT SPDT switch [START_REF] Kaper | Monolithic AlGaN/GaN HEMT SPDT switch[END_REF] has been realized and the measurements showed values of -0.87, -0.96, -1 dB insertion loss and -46, -42 and -41 dB isolation at 0.9, 1.8 and 2.1

GHz respectively. The switch also shows linear performance for power levels up to 1

Watt in the insertion mode and more than 2 Watts in the isolation mode. Due to its good characteristics, GaN technology could be extended to high frequency switch applications without the need for purely resonant operation. This leads to some advantages such as switch size and ease of design [START_REF] Mishra | AlGaN/GaN HEMTs-an overview of device operation and applications[END_REF]. As a result, many SPDT switches have already been successfully demonstrated around X-band, Ku-band or higher frequency. One of these works is [START_REF] Ma | High power AlGaN/GaN Ku-band MMIC SPDT switch and design consideration[END_REF] where an AlGaN/GaN HEMT Ku-band SPDT switch has been reported with an insertion loss of -1.4 dB, isolation of -35 dB, and 36 dBm output power at 18 GHz. Clearly, this work demonstrates the high power capabilities of GaN technology at high frequency for switch device.

Improving the isolation of GaN switch circuits over large frequency bands was a reason leading to the use of resonance inductors in parallel with the HEMTs.

Based on this technique, [START_REF] Janssen | X-Band GaN SPDT MMIC with over 25 Watt Linear Power Handling[END_REF] presents an X-Band GaN SPDT MMIC switch with over 25 Watt linear power handling. The achieved isolation, for this work, is around -35 dB in X-band, while the insertion loss was high and was about -3.5 dB. This insertion loss is high due to the on-resistor value of the device. Continuing with the same concept, a PhD student, Tyler Neil Ross, from the Department of Electronics (DOE) at Carleton University, developed high-power broadband GaN HEMT SPST/SPDT switches based on resonance inductors and shunt-stacked transistors [START_REF] Hettak | Highpower broadband GaN HEMT SPST/SPDT switches based on resonance inductors and shunt-stacked transistors[END_REF]. For the SPST an insertion loss less than -3 dB and isolation better than -13 dB from DC to 20 GHz were achieved, while for SPDT an insertion loss better than -2 dB, an isolation higher than -20 dB and a power handling at 1 dB compression (P1dB) greater than 10 W were reported. Going higher in frequency, in [START_REF] Zheng | Ka-Band High Power GaN SPDT Switch MMIC[END_REF] a Ka-Band high power GaN SPDT switch MMIC was demonstrated with unprecedented 49 dBm at P1dB.

Measured insertion loss is better than -1.3 dB over 27-31 GHz, and the minimum insertion loss of -0.8 dB has been achieved at higher frequencies. Measured isolation is greater than -25 dB over 27-31 GHz, with the highest isolation of -30 dB at higher frequencies.

Clearly, this literature review shows that GaN-based switches can achieve impressive high power handling capability beyond that of other wide band semiconductor technologies. In addition, excellent performances in terms of isolation and insertion loss can be achieved.

NRC GaN Switch Modeling

The AlGaN/GaN HEMTs switches investigated in this thesis work were fabricated by the National Research Council of Canada (NRC). In this process, there is considerable doubt that the provided models are accurate for switching applications because only transistors in common source configuration for power amplifier purposes are provided. In this section, a brief review of GaN modeling will be discussed. Then, a small-signal transistor model useful for analyzing and designing GaN switching circuits using this foundry will be presented.

Proposed GaN Single Switch Model

The modeling part is crucial in HEMT circuit designs. A good model certainly relies on a circuit simulation tool that provides the capability to determine circuit performance with high accuracy against prototype measurements. Also, it allows to carry out a good verification of all the different test structures, leading to prediction of the transistor's overall physical phenomena. However, to cover all the major effects influencing directly the HEMT's characteristics and performance, it is necessary to present a good model that is as general as possible.

For GaN HEMT technology, there has been interest in modeling transistors as basic building blocks. Actually, most of these efforts have focused on empirical [START_REF] Angelov | A new empirical nonlinear model for HEMT and MESFET devices[END_REF]- [START_REF] Angelov | An empirical table-based FET model[END_REF] and semi-physical formulations. GaAs and GaN technology are commonly referred to as wide bandgap compound semiconductors, which lead to the fact that the extraction of GaN HEMT parameters mostly follows the same procedures as those for GaAs HEMTs, with a few exceptions. GaN HEMTs have much higher conduction bands than the corresponding GaAs HEMTs. As a result, GaN models are based on GaAs models, and are then extended or modified to better fit into the form of GaN device's properties.

Various types of models have been employed for HEMTs, most of them are semi-empirical. Therefore, models are today based only on formulas which, also, seem to work for HEMT devices. For example, the hyperbolic tangent function is employed to approximate drain current. For GaN HEMTs, the most important effects which should be considered in a nonlinear modeling are:

 Self-heating: Transistor's ability to predict the dynamic changes of its temperature during operation due to power dissipation [START_REF] Curtice | Nonlinear modeling of compound semiconductor HEMTs, state of the art[END_REF].

 Nonlinear charge modeling, Qgs, Qds and Qgd, has been shown to be critical to accurately simulating bias dependence on all three terminal voltages (Gate, Drain and Source) [START_REF] Curtice | Nonlinear modeling of compound semiconductor HEMTs, state of the art[END_REF].

 Surface traps effects where gate-lag effects and drain-lag effects are included [START_REF] Binari | Trapping effects and microwave power performance in AlGaN/GaN HEMTs[END_REF]- [START_REF] Jardel | An Electrothermal Model for AlGaN/GaN Power HEMTs Including Trapping Effects to Improve Large-Signal Simulation Results on High VSWR[END_REF].

An accurate model is imperatively related to the measure of all reported effects. A mismeasurement or a lack of one affects the results by improperly extracting the device's parameters [START_REF] Lin | FET model parameter extraction based on optimization with multiplane data-fitting and bidirectional search-a new concept[END_REF]- [START_REF] Van Niekerk | Performance and limitations of decompositionbased parameter extraction procedures for FET small-signal models[END_REF]. An overly simple model can have significant effect at high power, leading to fail the device's performance. Using GaN500v1, some performances advantages have been achieved in terms of high breakdown voltage, low on-resistor and low leakage current at the gate [ [START_REF] Ross | Gallium Nitride Phase Shifters[END_REF]. This leads us to assume that the switch designs using GaN500v2 could have better performances compared to the ones fabricated using GaN500v1. The reported currently available simple linear model can be useful to estimate the power-handling capability of a switch circuit. The model presented can be easily integrated into the full circuit simulation, and the voltages across the transistor's terminals and the current through the device can be determined using a linear ac analysis available in virtually all simulators. As long as the input RF power does not induce a voltage swing across transistor's terminals that reaches the transistor's pinch-off voltage or breakdown voltage, the switch-based circuit should function largely as predicted from a linear S-parameter analysis. This simulation does not lend itself well to a quantitative prediction of the 1 dB compression point or input-referred third-order intercept point (IIP3). However, it does provide a designer with some useful information on the power-handling capability of the circuit, which can be refined once a large-signal model can be extracted, or when measurements are performed. switches. The measurements were limited at 10 GHz for two reasons. First, it was observed that the performances of the measured devices decrease at high frequency in terms of insertion loss and isolation. Second, the realized final structure is planning to be used for beam switching antenna at 2.45 GHz.

GaN500v2 Small Signal Switches Measurements

As can be seen, the isolation is good at low frequencies but quickly drops with increasing frequency. So, it can be said that the larger the device used, the lower the insertion loss is achieved but the poorer the isolation. These obtained results are in good agreement with the reported small signal analysis in [ [START_REF] Ross | Design of X-Band GaN Phase Shifters[END_REF] using GaN500v1 from NRC.

A trade-off between insertion loss and isolation should be made to choose the single transistors that will be used for designing switch structure such as SPST and SPDT. Switch structures such as SPST and SPDT are very popular in semiconductor switches applications, they are used for improving the performances (insertion loss and isolation) over large frequency bands. Based on the investigated standalone switches, SPST and SPDT structures will be designed and measured result will be presented and discussed. represented by a small on-resistor, and FET2 is in the pinch-off region, which can be presented as a small capacitor. Then the RF signal goes from the input port to the output port through the FET1 because input impedance is near 50Ω. In the other hand, when the SPST switch is in the off-state, FET1 behaves as a small capacitor instead when a control voltage at -5 V is applied for VG1, and FET2 can be treated as a small resistor by applying 0V for VG2. Therefore, the input impedance is very high, and the input RF signals are completely reflected back to the input terminal. GHz, an isolation of -20 dB is still achievable while with 4x50 single switch an isolation of -10 dB was achieved at 3 GHz. In addition, the insertion loss value achieved by this structure is better than -3.5 dB up to 3 GHz and then starts degrading. However, the insertion loss of 2x100 single switch was better than -2 dB up to 5 GHz (Figure III.47).

Comparing to the 2x100 standalone device, the isolation is improved while the insertion loss slightly dropped by -1 dB. With this proposed topology, it is still a challenge to meet good performances at higher frequency. Another topology based on SPDT is proposed to improve again the switch performances. When the transistor FET1 as well as the FET4 are in the ON state corresponding to VG1 = 0 V and VG2 = -5 V, FET1 behave as a small resistor which allowing to the RF signal to go from port_1 to port_2. Verse-versa for the second state when the signal is from port_1 to port_2 As was expected with this topology, some improvement in isolation is achieved compared to SPST, while the insertion loss is increased by almost 1 dB. For example, at 3 GHz an isolation of -25 dB and an insertion loss of -3 dB were achieved.

Unfortunately, the SPDT structure performances degrade with frequency. As we hope to go higher in frequency, around 10 GHz, with these realized switch topologies it will be impossible to achieve good performance. So, the SPDT structure which presents good overall performances compared to SPST will be used in a beam switched antenna system at 2.45GHz. At this frequency, an isolation better than -25 dB and an insertion loss of -3 dB are possible.

Due to the unavailability of having large signal equipment and also a RF probe of type GSGSG to measure the outputs of the SPDT either at IETR (University of Rennes 1) or DOE (Carleton University), we could not do the large signal measurement of this structure. However, as will be presented in Chapter IV (Section V), once this SPDT structure was mounted over a PCB by using wire-bonding technique, the large signal measurement have been conducted and results are presented and discussed in Chapter IV. In addition, by using ADS the SPDT small Finally, in this chapter, different sizes single standalone switches and switch structures such as SPST and SPDT were fabricated and presented. As mentioned, only small signal measurement were reported. It was observed that, with single transistor in common gate configuration, the isolation is good at low frequencies but quickly drops with increasing frequency. In addition, it was observed that more the device is large, lower the insertion loss is achieved but poorer the isolation will be. A trade-off between insertion loss and isolation has been made to choose the single transistors that will be used for designing switch structures such as SPST and SPDT which could improve the performances at high frequencies.

Comparing to the standalone device, it was revealed that with SPST structure, the isolation is improved while the insertion loss slightly dropped by -1 dB. With this proposed topology, it is still a challenge to meet good performance at our higher frequency. Another topology based on SPDT was proposed to improve more the performances. As was expected with this topology, some improvement in isolation is achieved compared to SPST, while the insertion loss is increased by almost -1 dB. For example, at 2.45 GHz (frequency of interest) isolation better than -25 dB and an insertion loss of -3 dB were achieved. However, the SPDT structure performances degrade with frequency. Moreover, based on these small signal measurements, a model was introduced. It was observed that this linear model is the same to the one proposed in [ [START_REF] Ross | Design of X-Band GaN Phase Shifters[END_REF] by Tyler Ross using GaN500v1 from NRC foundry. In addition, the fabricated SPDT switch structure will be integrated with antenna system to realize and to demonstrate the ability of using GaN devices for a beam switching antenna array.

Chapter IV

--------------------------------------------------------------------------------------------------------------------- Realized GaN-Based Reconfigurable Circuits I. Introduction
As a result of the emerging need for multi-band and multi-services requirements within a single design structure, the electronically tunable or reconfigurable devices have experienced [START_REF] Boeck | RF front-end technology for reconfigurable mobile systems[END_REF], over the last years, with a significant growth in radio frequency (RF)/Microwave wireless front-end applications such as antennas, power amplifier, phase-shifters, oscillators and filters. Moreover, this has been the modern trends in technological of the new compact-structured RF transceivers [START_REF] Vidojkovic | Adaptive Multi-Standard RF Front-Ends[END_REF].

Also, the use of wireless devices has increased dramatically over the last years. This increased utilization has led to a need for more efficiency RF circuits with more functionalities. Therefore, the continuing demands for highly portable, more reliable and more power efficient, low cost wireless systems require that future wireless devices must be compatible with different communication protocols and capable of operating over a wide frequency range. Furthermore, applications of such systems in personal and satellite communications impose the specifications for circuits miniaturized in size and weight. Thus, as wireless devices become more and more compact, the appeal of reconfigurable circuits is that they can decrease the number of components with identical specifications, decrease the number of input/output pins, reduce the size of the integrated circuit area, and reduce power consumption [START_REF] Eshghabadi | MEMS-based Reconfigurable CMOS LNAs for Wireless Applications[END_REF].

Tunable devices with a high performance are, also, more crucial for mobile devices and are for selective applications. Generally, an adaptive RF circuit must have the capabilities to handle different functionalities such as signal routing, impedance matching agility, adjustable gain amplification, tunable phase shifting [START_REF] Lourandakis | Circuit Agility[END_REF], etc. In contrast, a tunable circuit with embedded tuning elements can carry out a switching function of several parallel circuits with a simple change to one of the elements. Figure III.1 illustrates this example for the case of duplexers in a tri-band system where a single tunable duplexer can replace three fixed duplexers [START_REF] Manuel | Composants innovants reconfigurables par voie optique pour des applications microondes[END_REF]. Comparing the above architectures, a single tunable duplexer offers higher flexibility, more functionalities, lighter weight and denser integration. In addition, such a circuit tuning allows an easier controlling of systems through baseband signal processing, which can add attractive features of smartness of combined multiservices. Therefore, it is obvious that due to the convergence towards the integration of multiple standards and services into a single circuit and maintaining their expected performances at various configuration parameters, wireless systems are becoming more complex and highly smart.

Depending on the specific RF design where reconfigurable components are suitable, the result may be a saving in cost, reduction in size, better performance, and/or increased functionality. In order to implement these electronically reconfigurable components, there are currently various main techniques [START_REF] Lourandakis | Circuit Agility[END_REF] that could be listed as follow: based on semiconductors technology have become more attractive for electronic reconfigurability due to their small size, easiness integrations, possibly to be massproduced, etc. Most MMICs currently in production, such as GaAs, SiC, InP, etc, operate in the microwave and mm-w range with modest breakdown voltage and limited power handling capabilities. But there are increasing applications in the RF/microwave spectrum with higher power requirement.


As already presented in chapter II, GaN technology can be ideal for the design of high power devices for microwave and mm-w applications. In-depth research studies have been conducted to explore the performance of GaN technology, as well as reconfigurable RF/microwave circuits using GaN devices [START_REF] Norris | A fully monolithic 4-18 GHz digital vector modulator[END_REF]. GaN HEMTs semiconductor devices are no longer limited to RF power amplifier applications, having been emerged to other applications such as LNAs, VCOs, mixers, and phase shifters.

For the above mentioned strong points of reconfigurable components, GaN technology can be considered as a real promising candidate semiconductors technology, capable of significantly reducing the number of components with identical specifications, size and cost of devices utilized in transmitter and receiver chains of front-end modules used in RF/microwave systems.

II. Tunable Phase Shifters

Review

Phase shifters are important components that find their use in many RF/microwave systems. For example, they can be used in electronic beam-scanning phased arrays, microwave instrumentation and measurement systems, power divider, and as modulators [START_REF] Norris | A fully monolithic 4-18 GHz digital vector modulator[END_REF] in communication sub-systems.

Phase shifters can be analog or digital. Analog phase shifters provide a continuously variable phase, which can be controlled by a voltage. Electronically controlled analog phase shifters can be realized with varactor diodes that change capacitance with voltage, or nonlinear dielectrics. In the other hand, digital phase shifters provide a discrete set of phase states that are controlled by two-state "phase bits". They are characterized by a certain number of bits. This number defines straightforwardly the achieved phase. More bits there are, smaller phase step will be.

For example, a five bits phase shifter would have a 11.25 °(= 360 °25 𝑏𝑖𝑡𝑠 ) as a phase shift step. This means that each bit allows to get a phase shift successively over 11.25°, 22.5°, 45°, 90° and 180°.

98

There are various design parameters that should be considered such as frequency range, total phase variance (Δϕ), phase resolution, insertion loss, bandwidth, return loss, harmonics level, power handling and design area. This makes some challenges in the design and optimization of integrated phase shifters. In addition, there are five main phase shifter topologies and described as follow [START_REF] Garver | Broad-Band Diode Phase Shifters[END_REF]: must be unchanged and equal to 1 in order to not introduce any additional loss into the phase shifter. Practically, it is impossible to achieve perfect reflection from a load which means that some power will be dissipated in the load and |Γ| will be less than one. Therefore, some loss will always occur between the input and output signals. By inserting varactors in the terminations, a continuous control of the phase shift is achieved in this manner. To design this phase shifter topology, firstly, a 3-dB 90° hybrid coupler has been designed and simulated. The simulated results will be discussed later in this section. Then, the two coupler's outputs ports have been loaded by GaN varactors.


The final phase shifter design presented in this section was designed and were planned to be manufactured by using GaN150v1 process. However, due to factors beyond our control, fabrication of this phase shifter at the CPFC-NRC could not proceed as planned. The design was planned to be submitted for fabrication on 7 th September, 2015 and then four months later samples should be delivered. Since this design was the only application for 2x2mm 2 space in this run fabrication referenced as "1503GK" that cost $70K/run and because it is outside of our funds to purchase the entire run. The number of participants is not enough so the run of manufacturing has been completely cancelled. Due to this unexpected fabrication issue, only simulation results and design steps will be reported here However, this design approach could require several mm 2 of area to be realized and renders their size relatively large even for mm-wave frequencies due to the length of λ/4 lines [START_REF] Frye | A 2-GHz quadrature hybrid implemented in CMOS technology[END_REF]. Thus it would consume considerable wafer space of the used semiconductor technology, and the design size gets much larger. Additionally, the most important issue of the distributed λ/4 hybrid coupler in MMIC applications is the chip area which is directly related to the chip cost. Also the most significant challenge when dealing with a priced semiconductor technology especially GaN HFET technology is the high expense costs for technology access. Hence, there have 109 been several attempts to reduce the size of coupled line coupler, the most important alternative solution is to use lumped elements instead of resonant λ/4 transmission lines [START_REF] Frye | A 2-GHz quadrature hybrid implemented in CMOS technology[END_REF]. However, the bandwidth of the structure hybrid coupler is usually narrower than that of the typical λ/4 structure. Another issue comes up is that this structure becomes inefficient at high frequency (i.e mm-w) because of the inductors become very lossy and also tend to resonate at frequencies very close to the operating frequencies.

In this approach, the quarter-wave lines are approximated at operating frequency with lumped elements as depicted in Figure IV.15. Theatrically, the ABCD matrix of a single segment representing a perfect transmission line characterized by a characteristic impedance of Zr is written as follow:

Equation ( 11)

                 Cos Sin Zr 1 j jZrSin Cos D C B A    
The equivalent of this matrix by using lumped elements can be written as follow:

Equation (12)                                     1 1 ² 1 ²L1C1) - C1(2 j L1 j 1 1 ² 1 1 C1 j 0 1 1 0 L1 j 1 1 C1 j 0 1 D C B A C L C L        
From Equation [START_REF] Micovic | W-band GaN MMIC with 842 mW output power at 88 GHz[END_REF] and Equation ( 12), we can drive the following equations: Equation ( 13)

1 1 ² 1 Cos C L     Tunable Phase Shifters 110 Equation (14) L1 ZrSin    Equation (15) ²L1C1) - C1(2 Sin Zr 1    
The combining between Equation 14and Equation 15gives the following equation: Equation ( 16)

² ) 1 ( - L1 C1 2 Zr 1 C  
So the passive elements, L1 and C1, of the hybrid coupler can be calculated by the following driven equations:

Equation (17)   ZrSin 1  L Equation (18)    cos 1 cos 1 Zr 1 1    C
By using the same previous procedures, the other passives elements, L2 and C2, can be, also, calculated as follow:

Equation (19)   ZpSin 2  L Equation (20)    cos 1 cos 1 Zp 1 2    C
So these previous equations give theoretically first guess values of the lumped elements of hybrid coupler.

Based on this topology, we report here a simulation analysis of a new fully embedded lumped element GaN MMIC 3-dB/90° hybrid coupler at X-band that it will be later used to conduct an analysis of a phase shifter design. A three-dimensional layout view of the coupler, showing embedded metal and vias layers, of the designed

coupler is displayed in Figure IV.20.
The built block of the 3-dB/90° hybrid coupler using lumped spiral inductors and metal-insulator-metal (MIM) capacitors is very small in size (0.85x1mm 2 ). It consists of lumped multi-turn spiral inductors and MIM capacitors from NRC GaN150 PDK components library [START_REF]Gallium Nitride MMIC Foundry Design Manual GaN[END_REF]. The mutually coupled inductors turns are rectangular spiral inductor type and are implemented with reinforced between two metals, metal 2 (2ME) on metal 1 (1ME) by using via connection (VIA2), which results in an inductor having a higher Q-factor and size reducing. The current carrying capability of this inductor is limited by the width of the air-bridges. Higher currents can be sustained by increasing the width of the air-bridges, with a commensurate increase in the gap spacing between the inductor turns. A sample layout for a square inductor is shown in Figure IV.16, while in Figure IV.17 an inductor equivalent circuit is presented which is used to estimate the inductance (LS), series resistance (RS) and first resonance frequency FRES.

The MIM capacitors are formed by 2ME on top of 1ME without a VIA2 connection. A layer of silicon nitride, also used to provide insulation, is used as the dielectric layer. The dielectric constant of the silicon nitride is 6.5 and the thickness is 0.3um [START_REF]Gallium Nitride MMIC Foundry Design Manual GaN[END_REF]. The capacitance density of the MIM structure is 0.19 fF/µm 2 . This compact 3-dB 90° GaN MMIC hybrid coupler based on lumped elements has been successfully designed by using NRC GaN150v1 process. The advantages of size reduction and good performance of the proposed 3-dB 90° hybrid coupler at X-band have been confirmed through the simulated results. Comparing to a reported GaAs hybrid coupler using distrubuted elements [106] with a size of 3.1x2.3mm 2 , this design (size of 0.85x1mm 2 ) has shown a size reduction of 70 % in the circuit area at 10 GHz. These results indicate that the proposed hybrid coupler is very promising for high density and tunable MMIC applications. Moreover, the hybrid coupler is used for designing a highly miniaturized and tunable phase shifter. Where 𝛥𝜙 and αmax are the differential phase shift and maximum insertion loss, respectively.

Based on this phase shifter structure, one varactor diode placed at the two coupler's output ports could enough to get a theoretical phase shift of 180° as explained in [START_REF] Garver | 360° Varactor Linear Phase Modulator[END_REF]- [START_REF] Lucyszyn | Analog reflection topology building blocks for adaptive microwave signal processing applications[END_REF]. However, varactor diodes are not perfect. This mean that there are losses due to their equivalent series resistor also these diode have a limited tuning capacitance range, thus makes it hard to achieve a 180° complete phase shift.

The overall simulated performance and operating frequency range of this phase shifter are compared favorably with those of the reflection phase shifters using GaAs MMIC technology [START_REF] Chen | A Low-Loss Ku-Band Monolithic Analog Phase Shifter[END_REF]- [START_REF] Ellinger | Compact reflective-type phase-shifter MMIC for C-band using a lumped-element coupler[END_REF]- [START_REF] Hayashi | A miniaturized MMIC analog phase shifter using two quarter-wave-length transmission lines[END_REF]. In [START_REF] Chen | A Low-Loss Ku-Band Monolithic Analog Phase Shifter[END_REF] a phase shift of 109° with an insertion loss of 1.8 dB over 16-18 GHz was achieved by only placing one inductor and varactor at each coupler's output port. This inductance increases more the overall impedance (varactor and inductor) tuning range. It is possible to achieve a 360° phase shifter range with 4.2 dB insertion loss in the same frequency range by connecting three phase-shifter chips in series as reported in this same work [START_REF] Chen | A Low-Loss Ku-Band Monolithic Analog Phase Shifter[END_REF]. Using GaAs MMIC technology, another work [START_REF] Ellinger | Compact reflective-type phase-shifter MMIC for C-band using a lumped-element coupler[END_REF] was presented where a phase shift of 210° with a loss of 4.9 dB and a 1-dB input compression point of higher than 5 dBm was measured at 6.2 GHz. Based on the same structure, a miniaturized Ku-band GaAs MMIC phase shifter using two quarter-wave-length transmission lines was reported in [START_REF] Hayashi | A miniaturized MMIC analog phase shifter using two quarter-wave-length transmission lines[END_REF]. A phase shift of more than 180° and an insertion loss of 3.6 dB are obtained at the frequency range from 12 to 14 GHz. The chip size of the experimental MMIC phase shifter is less than 3.0 mm 2 . In this section, a GaN phase shifter design based on reflective topology has been presented for high RF power source application. First, the 3-dB 90° hybrid coupler has been designed and simulated using NRC GaN150 PDK. Then two developed GaN varactors were paced on the two coupler's output ports as loads.

Under different bias conditions varying from -3.5 V to -5.5 V, the return loss of the phase shifter is better than 20 dB across a 1 GHz bandwidth centered at 10 GHz, while good match is achievable over this range of 9.5-10.5 GHz. Worst case of the insertion loss is about -3 dB at 10 GHz and occurs at a bias point of -4.3 V where the equivalent resistor of the used varactors is significantly high. Therefore, a continuously variable phase-shift range of 80° is achieved at 10 GHz. These achieved performances were compared favorably with other works that use GaAs technology.

III. GaN Reconfigurable 3-dB/90° Hybrid Coupler Recently, considerable work in tunable couplers consist of MMIC-based active couplers and varactor diode based passive couplers have been reported in the literature which most of them are operating below X-band frequency. However, most of these reported couplers exhibit, in general, a small frequency tuning range. It is still a challenging task to design tunable couplers with tuning ranges wide enough to cover desired frequency bands. For instance, in [START_REF] Scheeler | GaAs MMIC tunable directional coupler[END_REF] a tunable hydrid coupler is reported using GaAs technology. This presented tunable GaAs MMIC directional coupler design uses tunable capacitances to achieve a tunable coupling coefficient (|S31|) at 2 GHz. Over a bias range of 0 V to -1.5 V the measured S-parameters for the coupled port demonstrate a tuning range of -6.6 dB to -60 dB. The input return loss is better than -12 dB for all bias voltage levels with isolation better than -17 dB.

This design is implemented in a 0.5 µm GaAs process with Schottky varactor diodes, and occupies an area of 2.2× 1.4mm 2 . In [START_REF] Hur | Tunable Broadband MMIC Active Directional Coupler[END_REF] a frequency tunable broadband MMIC active directional coupler is discussed. The coupler provides various coupling gains such as 3 dB, 0 dB, -3 dB, and -6 dB, also a tuned frequency from 5.5 GHz to 7.8 GHz was achieved by applying different bias voltages. Using varactors for tunability purpose, a novel compact dual-band reconfigurable power divider is presented in [START_REF] Gatti | A Novel Compact Dual Band Reconfigurable Power Divider for Smart Antenna Systems[END_REF] for smart antenna applications. Measured results show that, by biasing one varactor and keeping the other one at 0V, good performance are achieved at the center frequency 2.45 GHz, while this frequency can be tuned to 5.25 GHz by only switching the bias state of varactors. Also, any value of power ratio can be selected from 1:0 to 0:1 for both the center frequencies 2.45 GHz and 5.25 GHz, maintaining an average insertion loss better than -1.5 dB and -3 dB in the 2.45 GHz and 5.25 GHz bands, respectively, along with a return loss better than -15 dB in both frequency ranges.

Measured isolation at port 2 is always better than -20 dB in both frequencies. An oscillator is an electronic circuit which can generate a periodic signal by converting DC power to RF power. Two approaches, such as feedback oscillator and negative resistance oscillator [START_REF] Voinigescu | High-Frequency Integrated Circuits[END_REF], are commonly used for the design of oscillators.

In the feedback topology, the oscillator is constituted of an amplifier with a positive feedback network (see Figure IV.37) that are forming a loop. The condition for oscillation is that a part of output signal is combined in phase with the input signal and the amplitude of the loop gain is larger than unity. This topology is used to design two oscillators in this thesis. [133]- [START_REF] Hörberg | Phase-Noise Analysis of an X-Band Ultra-Low Phase-Noise GaN HEMT Based Cavity Oscillator[END_REF], while there are a few reported works for millimeter-wave frequency such as Q-band [START_REF] Lan | A Q-band low phase noise monolithic AlGaN/GaN HEMT VCO[END_REF] and V-band [START_REF] Lan | A V-Band Monolithic AlGaN/GaN VCO[END_REF].

Earlier in 2001, the first report of AlGaN/GaN HEMT-based VCOs operating at 6 GHz is presented in [START_REF] Shealy | Low-phase noise AlGaN/GaN FET-based voltage controlled oscillators (VCOs)[END_REF]. An output power of 27 dBm and phase noise level of -92 dBc/Hz at 100 KHz offset were achieved. This achieved phase noise value was favorably equivalent to the single-sideband noise of a GaAs FET-based oscillator [START_REF] Shealy | A 2 watt Ku-band linear (multi-carrier) transmit module for VSAT applications[END_REF]. It is also clearly shown in [START_REF] Shealy | Low-phase noise AlGaN/GaN FET-based voltage controlled oscillators (VCOs)[END_REF] that GaN oscillators with large tuning range can achieve a phase noise of -123 dBc/Hz at 1MHz, whereas those fixed frequency ones can achieve a phase noise as low as -135 dBc/Hz [START_REF] Liu | Design of Ultra-Low Phase Noise and High Power Integrated Oscillator in GaN-on-SiC HEMT Technology[END_REF] at 1MHz. To the authors' knowledge of [START_REF] Liu | Design of Ultra-Low Phase Noise and High Power Integrated Oscillator in GaN-on-SiC HEMT Technology[END_REF], this GaN-based oscillator achieves the lowest phase noise compared to other GaN HEMT based integrated oscillators. An output power of 21 dBm is achieved at 7.9 GHz when the device is biased at Vgs of -3V and Vds of 28 V.

Another design is presented in [START_REF] Xu | Low phase-noise 5 GHz AlGaN/GaN HEMT oscillator integrated with BaxSr1-xTiO3 thin films[END_REF], which used an in-house GaN HEMT facility. The phase noise of the designed 5 GHz oscillator can be kept more or less as low, -105 dBc/Hz at 100 kHz, as the ones (-112 dBc/Hz at 100 kHz) designed in GaAs [START_REF] Zirath | Balanced Colpitt oscillator MMICs designed for ultra-low phase noise[END_REF] while an output power of more than 20 dBm is achieved. Still over the C-band, differential GaN oscillators are reported in [START_REF] Sanabria | A GaN differential oscillator with improved harmonic performance[END_REF]. An output power level of 23dBm at 4.16GHz is reported, while phase noise is -86 dBc/Hz at offsets of 100 kHz and -115.7 dBc/Hz at offsets of 1 MHz. The second and third harmonic suppression are equal to -45dB and -70dB respectively. High power single-ended Colpitts-type GaN oscillators using field-plated HEMTs are described in [START_REF] Xu | High power GaN oscillators using field-plated HEMT structure[END_REF]. Both output power and operation bias can 131 be significantly increased to 32.8 dBm and Vds = 40V with an output power density of 3.8W/mm at 5GHz. The DC-to-RF-conversion efficiency amounts to 21%, while the measured phase noise at 1MHz offset from the carrier amounts to -132 dBc/Hz. A recent C-band work is reported in [START_REF] Do | A MMIC GaN HEMT Voltage-Controlled-Oscillator with High Tuning Linearity and Low Phase Noise[END_REF] where a MMIC GaN HEMT VCO with high tuning linearity and low phase noise is presented. The designed VCO is based on a balanced Colpitts topology and is tunable between 6.45-7.55 GHz with good tuning linearity, constant output power of 2 dBm at a bias condition of Vd = 6 V and an associated Id of 33 mA, and a good phase noise with little variation over the tuning range. The measured phase noise is -98 dBc/Hz at 100 kHz and -132 dBc/Hz at 1 MHz offset frequency.

Over the X-band frequency, some GaN-based oscillators have been implemented. In [START_REF] Tilak | High power monolithic AlGaN/GaN HEMT oscillator[END_REF] a VCO with an output power of 35 dBm between 8.5 and 9.5GHz is presented. Output power density of 2.1W/mm is reached at Vds = 30V. The technology uses GaN HEMTs with a gate length of 0.15 µm. The phase noise is estimated to be -87 dBc/Hz at 100 kHz offset. Another interesting X-band GaN HEMTs-based oscillator is also designed in an in-house facility and is presented in [START_REF] Cheng | A Low Phase-Noise -Band MMIC VCO Using High-Linearity and Low-Noise Composite-Channel HEMTs[END_REF]. The VCO exhibits a frequency tuning range from 9.11 to 9.55 GHz, an average output power of 3.3 dBm with phase noise of -82 dBc/Hz at 100 kHz offsets and -110 dBc/Hz at 1 MHz. These performances are investigated under a 5 V drain bias instead of utilizing large device with high drain bias to achieve the high output power.

Although the phase noise of this oscillator still needs to be further improved, it opens up a new examination on the design of oscillators in GaN HEMT technology for low voltage, moderate output power as well as low phase noise. A design of an X-band GaN oscillator is discussed in [START_REF] Soubercaze-Pun | Design of a X-band GaN oscillator: from the low frequency noise device characterization and large signal modeling to circuit design[END_REF]. At 9.9 GHz, an output power of 20 dBm is measured, with a phase noise level around -105 dBc/Hz at 100 kHz offset frequency, but with a poor 2nd harmonic found at -10 dBc, while the third harmonic is rejected more than -30 dBc from the carrier. An X-band oscillator is presented in [START_REF] Lai | Low Phase Noise GaN HEMT Oscillators With Excellent Figures of Merit[END_REF] using common-gate balanced Colpitts oscillator topology. A commercial Triquint's 0.25 µm GaN MMIC HEMT process is used. Using the same commercial GaN HEMT process as in [START_REF] Lai | Low Phase Noise GaN HEMT Oscillators With Excellent Figures of Merit[END_REF], a recent X-band ultra-low phase-noise GaN HEMT oscillator is described in [START_REF] Hörberg | Phase-Noise Analysis of an X-Band Ultra-Low Phase-Noise GaN HEMT Based Cavity Oscillator[END_REF] by the same authors of [START_REF] Lai | Low Phase Noise GaN HEMT Oscillators With Excellent Figures of Merit[END_REF]. The designed oscillator is based on a GaN HEMT MMIC reflection amplifier and a rectangular aluminum cavity. It oscillates at 9.9

GHz with an output power of 5 dBm and an excellent phase noise of -145 dBc/Hz at 100 kHz and -160 dBc/Hz at 400 kHz. These are the best reported phase noise values so far by using GaN HEMT devices. Going higher in frequency, a low phase noise power-efficient MMIC GaN-HEMT Oscillator at 15 GHz based on a Quasi-lumped onchip resonator is reported in [START_REF] Hörberg | Low phase noise power-efficient MMIC GaN-HEMT oscillator at 15 GHz based on a quasi-lumped on-chip resonator[END_REF]. An excellent phase noise of -106 dBc/Hz and -133dBc/Hz, respectively, at 100 kHz and 1 MHz offsets and a low output power of -6 dBm are experimentally demonstrated.

A Q-band MMIC oscillator (39GHz) is presented in [START_REF] Hörberg | Low phase noise power-efficient MMIC GaN-HEMT oscillator at 15 GHz based on a quasi-lumped on-chip resonator[END_REF]. An output power of 25 dBm at Vds = 25V (Id=156 mA) is reported with a low-phase noise level of -120 dBc at 1MHz offset from the carrier. The DC-to RF-conversion efficiency is about 12%. Around 53 GHz, the same authors report a new V-band GaN MMIC VCO [START_REF] Lan | A V-Band Monolithic AlGaN/GaN VCO[END_REF] realized on a gate length of 0.2µm AlGaN/GaN HEMT process. The delivered output power is around 11 dBm at 53 GHz with an estimated phase noise of -97 dBc/Hz at 1

MHz offset with bias conditions of Vd = 15 V and Vg = -3V. These reported works [START_REF] Lan | A Q-band low phase noise monolithic AlGaN/GaN HEMT VCO[END_REF]- [START_REF] Lan | A V-Band Monolithic AlGaN/GaN VCO[END_REF] demonstrate the potential of using GaN technology for military and commercial applications that require high frequency, high power, and low phase noise frequency sources.

Design of GaN-based Oscillators

In this perspective, using GaN HEMT technology, two X-band oscillators are presented in this section: a fixed frequency oscillator (FFO) and a tunable frequency oscillator (TFO). The two oscillators are designed based on feedback oscillator topology and are fabricated. The GaN HEMT based FFO is designed to operate around 10 GHz while the GaN HEMT based TFO is tunable between 7.8-8.6

GHz, corresponding to a tuning range close to 10%. The frequency reconfigurability 133 of the designed TFO oscillator is based on the in-house developed GaN varactor detailed in chapter III.

Feedback Oscillator Topology

Oscillator based on feedback topology using amplifier approach is introduced in this section. The GaN PA design used here is presented in next section.

A frequency-selective feedback network printed on Neltec NX9300 substrate is designed to return part of the output signal to the amplifier input. From Figure IV.37, it can quickly derive the expression of the transfer function of the feedback oscillator as:

Equation (22) G G Vi Vo . 1   
This design topology offers some very attractive features, including a straightforward design, a relatively pure sine-wave output, and a very stable frequency.

Power Amplifier Design

As reviewed in Chapter II, many GaN-based PAs have been reported.. In this thesis, a single frequency power amplifier operating at X-band is designed and realized, using a GaN discrete bare die transistor from Cree. This PA will be used to conduct the complete analysis of the reconfigurable frequency oscillator using GaN HEMT technology at X-band. This topology of oscillator is mainly based on a PA that may be biased in class A, B, C or AB since a gain more than 1 dB is reached, this allowing to satisfy the first oscillation condition. For this PA, the class AB operation was chosen. and the device's breakdown voltage is about 125 V. As reported in datasheet [139], the active device is unconditionally stable above 12 GHz and potentially unstable otherwise. Therefore, a series gate resistor of a small value around 0.175 Ω is required by Cree to get device stable at 10 GHz. Since, S-parameter response of the device is function of the bias points, stability factors must be simulated at the operating bias conditions. This step in designing the PA is to determine with the small signal model whether the FET transistor is unconditionally stable or potentially unstable. This can be easily analyzed using the stability factor (K-factor) and stability measure delta (Δ). Δ > 0 and K > 1 are necessary and sufficient conditions for stability. The equations for K-factor and Δ are defined as follow: Equation ( 23

PA Simulated Results

Since this PA design will used as a feed-back oscillator, large signal simulation will not be conducted. So Small signal simulation results of the PA are During the measurement, Vgs was fixed at -2.5V and then we start increasing slightly the Vds to get the wanted Ids current. Once the Vds gets higher than 19 V, it was observed that the Vgs starts getting changed alone randomly. The Vds starts to control the Vgs value and we lost completely the control over the Vgs. We could not very understand this weird behaviour. It is known that when there is a drop value in gain this means that the PA is not stable. Maybe a little bit more complicated stability circuits than a simple series resistor recommended by CREE is required to make sure that the device is stable.

The transistor model of the CGHV1J025D provided by CREE Inc. only has one input port and one output port, despite the fact that the transistor itself is constituted from twelve cascaded single transistors, so CGHV1J025D has twelve input ports and output ports. Unfortunately a distributed transistor model that can allow to check if all the single devices work well was not available, also it was impossible to check if the mounded device was damaged or not. Due to our limited funds, we could not offer another fabrication prototype. In addition, to redo another fabrication will be considered as very time consuming.

Since this PA is planned to be used as a feed-back power oscillator, it will be very difficult to achieve our objective and make the oscillator works well. Nevertheless, the study of the two oscillators has been done and presented in the following section. the oscillator or generated by the active and passive devices in the oscillator, is responsible for starting-up the oscillation regime. As the power supply is turned on, noises will be shown up and then oscillation will grow in amplitude. Although, this noise is characterized by a broadband spectrum but only noise at the oscillation frequency will be amplified. The gain of the amplifier and feedback loop must be equal or greater than one at the desired oscillation frequency, while the phase shift through the loop must be zero or some integer multiple of 2π. These are known as

Barkhausen Criterion [START_REF] Caillet | Etude et Conception d'un Système Radar Courte Portée et d'Antennes Reconfigurables en Bande Millimétrique[END_REF] of oscillations and are required to get the oscillating regime process ongoing.

Usually the designed loop gain magnitude is slightly larger than unity at the desired frequency of oscillation because if exact unity is designed, a slight reduction in gain would result in oscillations that decays to zero. So the oscillation conditions in open loop can be summarized as: To ensure that the design meets the necessary conditions for oscillation, a small signal gain analysis has to be done before connecting the feedback network to the amplifier's input. This analysis is mostly called open-loop analysis where gain and phase have to satisfy Barkhausen Criterion of oscillations. In addition, when dealing with oscillator designs, closed-loop analysis is also required to explore the oscillator's performances such as frequency and power of oscillation as well as phase noise. This analysis is conducted once the feedback loop is connected to the amplifier's input terminal. Oscillator's performances at both open-loop and closedloop are discussed here for FFO as well as for TFO.

The goals of the two discussed oscillators designs is firstly to fulfill the indepth progress of GaN oscillator design, and to achieve an high oscillator output power at 10 GHz with low phase noise. Also a frequency reconfigurability over X-band is another goal of this conducted study. The configuration of both the FFO and TFO designs is shown, respectively, in Figure IV.47 and Figure IV.54. The circuits was simulated using ADS. Small signal and harmonic simulations were performed.

Phase Noise in Oscillator

Phase noise is one of the most important figures of merit of a signal generating device and it is often the limiting performance metric. The basic concept of phase noise involves frequency stability, which is defined as the degree to which an oscillating source produces the same frequency throughout a specific period of time. Phase noise is generally specified in dBc/Hz at a given offset frequency for a particular carrier. Therefore, phase noise can be found by measuring the ratio of the power spectral density (1-Hz bandwidth) at a given offset frequency to the total power at the carrier frequency. In the literature, it is common practice to refer to the values of phase noise as "close-in" for the frequency offset range of 1 Hz to 100 Hz. That is, "close in phase noise" refers to the phase noise close to the carrier, less than 1 kHz away. Similarly, "far out" phase noise commonly refers to values 1 MHz or more from the carrier. That is, "far out phase noise" refers to the phase noise far from the carrier, more than 100 kHz away. The mid-range region, 1 kHz to 100 kHz offset, especially for signal generator phase noise, is sometimes referred to as the pedestal region.

It is well known that depending on the application, the offset frequency is selected. Here in this designed oscillator, phase noise is simulated at three offset frequencies, such as 10 KHz, 100 KHz and 1 MHz. As mentioned, the idea behind the design of this oscillation is to verity the performance of using the developed GaN varactors, so these offset frequencies were selected only because most of the reported GaN oscillator designs found in the literature also use these three selected offset frequencies, and we assumed that for a good comparison, the noise phase should be simulated also at the same offset frequencies. As can be seen there are a lot of oscillation frequencies below 10 GHz, this means that the feedback network do not filter these frequencies. As already reported, the PA that is used here for this oscillator did not work as intended from simulation, so it is obvious that the FFO does also not work as expected. At the operating frequency of 10 GHz, it is observed that a peak of a power of -10 dBm is measured while the highest power peak of 26 dBm was observed at very low frequency around 500 MHz as reported in Figure IV.49. These results are recorded at bias points of Vgs of 18 V and Vgs of -2.5 V where an Ids current of 260 mA was achieved. The same problems observed for the PA with the biasing are also observed during the FFO measurements. It was noted that the Vds starts controlling the Vgs.

Due to the fact that the design did not work as intended from simulations, measurements of the power levels of the second and third harmonics as well as of the phase noise are not reported. In Figure IV.56 the output power versus bias voltage VG of the TFO is illustrated. It can be seen that the power is about 20.5 dBm and almost stable over all bias levels with a slight variation over range [START_REF] Ross | Gallium nitride hybrid microwave circuits for low-noise applications[END_REF] where VG is between -5 V and -3.7 V. This drop in power variation can be due to the variation in equivalents series resistor of varactor as presented in previous Chapter III. Compared to the output power of the FFT which is about 34.7 dBm, the output power of TFO is decreased to 20.5. This can be due the RF losses of varactors used for tuning purpose.

Measurement Results

For the measurements part, the same remarks as for the PA and the FFO have been observed for the TFO design.

In this section, two GaN oscillator designs based on feedback topology have been presented for high RF power source. First, the power amplifier and optimized feedback loop were designed and fabricated using a Cree's discreet active FET device and NELTIC9300 substrate for the PCB design. The amplifiers has an open-loop gain of 6 dB, good return loss at input and output better than -12.5 dB, and phase of 0° at 10.12 GHz.

Secondly, the feedback loop was closed to design the first oscillator (FFO).

Phase noise about -110 dBc/Hz @ 100 kHz was simulated at the bias Vds of 40 V and Ids of 240 mA while the output power obtained for the FFO is 34.7 dBm at center frequency of 10.12 GHz. The second and third harmonics were very low and typically better than 40 dBc. These simulations results are favorably compared to the best reported performances of high power GaN oscillator [START_REF] Tilak | High power monolithic AlGaN/GaN HEMT oscillator[END_REF]- [START_REF] Hörberg | Phase-Noise Analysis of an X-Band Ultra-Low Phase-Noise GaN HEMT Based Cavity Oscillator[END_REF]. In [START_REF] Tilak | High power monolithic AlGaN/GaN HEMT oscillator[END_REF] Using a GaN HEMTs with a gate length of 0.15 µm process, an output power of 35 dBm between 8.5 and 9.5GHz was achieved at Vds = 30V while the phase noise was estimated to be -87 dBc/Hz at 100 kHz offset.

Finally, a tunable frequency GaN oscillator (TFO) is designed by loaded symmetrically the feedback loop. For tuning purpose, GaN varactor were used. The oscillation frequency values change significantly and a tuning bandwidth of 0.8 GHz, corresponding to a tuning range about 10% was achieved. The output power is about 20.5 dBm and almost stable over all bias levels with a slight variation over range [START_REF] Ross | Gallium nitride hybrid microwave circuits for low-noise applications[END_REF] while a good phase noise value of -112 dBc/Hz is met at 100 KHz.. However, the output power was not impressive compared to FFO design, being only 20.4 dBm and 20.9 dBm at best. This drop of power can be explained as there is a RF power consumption of varactors due to the presented high equivalent series resistor. These two reported oscillators prove clearly that GaN HEMT technology, with its high breakdown voltage, is a good promising choice for the high RF power sources. Unfortunately the measurements did not work as expected from simulation, this may be due to a difficulty to bias the active device and/or do not have the same ground at drain and gate.

V. Frequency Reconfigurable Active Antennas 1. Review

Active integrated antennas (AIAs) researches have shown unprecedented development along with growing interest for wireless systems applications [START_REF] Sufyar | A frequency agility technique on a miniature omnidirectional antenna[END_REF]. The designs of AIAs are towards the designs that are more and more small, have the capability of operating at multi-frequencies [START_REF] Dhar | Power amplifier based integrated and miniaturized active antenna[END_REF] and of handling significant high power levels, which has been the greatest challenge. There are a plenty of techniques [START_REF] Sufyar | A frequency agility technique on a miniature omnidirectional antenna[END_REF] in which antennas can be frequency tuned such as using switches, varying substrate properties [START_REF] Varadan | Design and development of electronically tunable microstrip antennas[END_REF] and using PIN diodes. Using varactors diodes [START_REF] Whatley | Reconfigurable RF impedance tuner for match control in broadband wireless devices[END_REF] integrated with printed antennas can be another way to tune the antenna operating frequency. This technique based on varactors could be found in various types of reconfigurable active integrated antennas systems. However, active device such as varactor diodes presents some non-linear behaviors with high power RF signals that influence the performances of antenna.

As a result of advanced GaN based HEMT semiconductor devices such as good linearity, the designs of AIAs with high power handling capability and high frequency operation are now achievable. GaN-based HEMT semiconductor devices are no longer limited to radio frequency integrated circuit (RFIC) [START_REF] Hamdoun | Characterization of GaNbased HEMTs as varactor diode devices[END_REF], having now been more attractive to AIAs systems [START_REF]An Update on New Tunable Technologies[END_REF].

Furthermore, reconfigurable AIAs, with high power/frequency, and excellent efficiency, are in great demand. In this chapter, firstly, a novel frequency 154 tunable annular slot antenna using GaN HEMT varactor is presented for the first time. However, using GaN varactor requires a high voltage DC biasing. In addition, GaN varactor presents a rather high insertion loss caused by its series equivalent resistance resulting on reducing the radiation efficiency of antenna. A second frequency tunable aperture coupled slot antenna is then investigated to improve the gain.

Thirdly, a radiation pattern reconfigurable antenna is studied using different oriented printed antenna arrays and integrated with GaN HEMT switches to choose the beam direction of the complete antenna system.

Reconfigurable Annular Slot Antenna using GaN Varactor 2.1 Antenna Design

The used planar antenna is a simple element annular slot coplanar antenna. The structure of the realized antenna is illustrated in Figure IV.58. This antenna was already presented in [START_REF] Martin | Optically transparent frequency-agile antenna for X-band applications[END_REF]. The whole size of antenna is about 50×50x0.7 mm 3 and is printed onto glass substrate with Ԑr of 5.7 and tanδ of 0.006 at 2 GHz.

CST Studio simulator was used as an EM simulator to design and simulate the antenna. First, the antenna itself was simulated and results are presented and discussed. Then the developed GaN varactor model was added to load the antenna. This allows to electronically adjust the antenna frequency by varying the varactor bias voltage applied at the gate. As will be shown, measurement results of the frequency tuning range showed good agreement with the simulations.

The frequency tunability of this antenna has been investigated firstly by loading a GaAs varactor diode bought from the market [START_REF] Martin | Optically transparent frequency-agile antenna for X-band applications[END_REF], after that the developed GaN varactor diode (W200L0.5) using GaN500v2 process is used instead. Since, the used GaN varactor provides various capacitive load to the antenna, the resonant

Simulated Return Loss and Radiation Pattern

The antenna itself without the diode was firstly simulated and good matching was obtained at resonant frequency around 11 GHz as shown in The E-plane and H-plane are normalized to the maximum gain of the antenna.

Because the antenna structure is a coplanar simple element, it radiates in two opposite directions. In E-plane, the maximum beam direction occurs symmetrically around 30° and -30° where a max gain was observed, this can be due to the diffraction As can be seen, the minimum and the maximum values of the simulated return loss are, respectively, -10 dB and -6 dB over the bandwidth from 7.53GHz to 7.85GHz corresponding to a tunability up to 4% narrow bandwidth inside. It is clearly demonstrated that the S11 is not good as for the first simulation with only the antenna itself; here the overall impedance presented by the varactor is significantly impacting the matching of the antenna. As said before this antenna was designed and optimized to be integrated with a GaAs varactor diode. Because the antenna was already fabricated and prototypes were available in the laboratory (IETR), we kept the same design and only integrate the model of GaN varactor and the simulations were relunched. This is why we could not re-optimize the antenna design to achieve a good matching once the GaN varactor is integrated. However, the idea here was to validate the developed GaN varactor model and demonstrate the use of GaN varactor for reconfigurable AIA. This can be seen from the simulated S11 results where the resonant frequency of the antenna is electronically controlled by the DC voltage applied over the varactor model.

The structure of this used antenna is discussed in [START_REF] Martin | Optically transparent frequency-agile antenna for X-band applications[END_REF] where the design procedures, fabrication steps and details are presented. Maximum return loss occurred when the GaN varactor presents a high REq at some bias DC as plotted in chapter III. In this section, we provide experimental results of the proposed active integrated antenna. In order to verify the tunability of the antenna experimentally, the GaN varactor was integrated with the antenna using the wire banding technique and then the whole design was measured.

Firstly, the measured return loss was measured and it can be seen from Freq [GHz] 166

The developed varactor diode was integrated with the antenna design, and the whole active antenna system was simulated. Figure IV.72 presents the variation of the return loss of the antenna versus frequency for different values of the varactor diode bias voltage. As can be seen, the minimum and the maximum values of the simulated input return loss are, respectively, -46 dB and -11.4 dB over a narrow bandwidth from 9.34GHz to 9.5GHz. Also it can be observed that, once the varactor model was added, the antenna achieves a good matching for all the DC bias voltage better than the simulated matching for the first proposed antenna (annular slot antenna). However, the tunability of the achieved bandwidth is two times narrower.

It is about 1.7% of bandwidth from 9.34 GHz to 9.5 GHz. In this section, an electronically beam steering antenna using the developed SPDT switch structure is proposed for applications that operate at 2.45 GHz frequency. Again, the idea here is to validate the developed GaN SPDT switch structure fabricated in NRC GaN500v2 process, for this a global system with four rectangular 4x1 patch array antenna is designed to achieve a beam steering over the H-plane for four Φ possibilities such as, 0°, 90°, 180° and 270°. With the use of this structure, the global array antenna can steer its beam in H-plane by a single control DC voltage (Vc_1, Vc_2, Vc_3 and Vc_4) applied at the different single switches constituting the SPDT structure. In E-plane the beam is stable for all the SPDT switch structures possibilities. As discussed in the section about SPDT switch structure, the RF performances of this switch structure degrade versus frequency, so at this frequency of 2.45 GHz, an isolation better than -25 dB and an insertion loss around -3 dB have been predicted from the measurements of the single switches. In addition, the antenna system was designed on Neltec substrate 9300 having Ԑr of 3.5 and a thickness t of 1.6 mm. Due to its high thickness, this substrate is very rigid and make the vertical system easier to be mounted. Therefore a cooling plate was cut from a piece of aluminium (Al), and is mounted on the backside (ground plane) of each 4x1 patch arrays antenna and of the feeding system design to transfers the heat generated by GaN switch structure and provides an infrastructure to mount the whole antenna system, the supply circuits and the RF connectors. The 3D model of the designed antenna system is illustrated in dimensions with a length of 32 mm and a width of 38 mm in order to achieve higher gain that can be still higher to 0 once the RF signal goes through all the losses due to the SPDT (around -6 dB, wire-bonding and the substrate). In this design, the patch elements are connected using microstrip line with a length around half-wavelength (λg/2) in series feed and have been designed to operate around 2.45 GHz with input impedance of 50 Ω. The length and the width of the each connecting microstrip line are, respectively, 37.7 mm and 1 mm.

The SPDT switch structure are connected to the feeding system illustrated in The EM simulations allowing to get the return loss and the radiation pattern were performed by using HFSS. Table IV.5 summarizes the four cases or the four beam steering possibilities corresponding to the three SPDT switches states. For a better understanding of beam switching, the antenna radiation pattern is tilted towards the positive z-axis having maximum at Φ=90°, θ=0°, here switches are configured as mentioned in Table IV.5 for case 1, is tilled towards the positive x-axis when switches are configured as mentioned in case 2 having maximum at Φ= 0°, θ=90°, is tilled towards the negative z-axis when switches are configured as mentioned in case 3 having maximum at Φ= 90°, θ=-180° and for the fourth case it is tilled towards the negative x-axis when switches are configured as mentioned in case 4 having maximum at Φ= -180°, θ=-90°.

The overall gain of one 4x1 patch arrays antenna is found to be above 10.9 IV.5 . Furthermore, there will be also other losses due to the substrate and the wires-bonding connecting the SPDT to the transmission lines. The overall losses is measured for every case and will be presented later in this section. V curve, we conclude that the equivalent series capacitance is dependent on the geometry of fingers. This means that the larger is the device, the higher are the capacitance values. Moreover, it was observed that the extracted capacitance values from varactors with GaN500v2 are at least twice as large as those extracted with the same device size fabricated with GaN500v1 process. The tuning range of the varactor diode is limited, and it was found that over a range of 1 V, the CEq changes rapidly.

In addition, an equivalent simple physical circuit model was proposed and simulated, yielding good agreement between the simulated and the measured results. The obtained Cmax/Cmin ratios and figures of merit compare favorably with recently reported GaN varactors.

On the other hand, different size single standalone switches and switch structures such as single pole single throw (SPST) and single pole double throw (SPDT) were fabricated using NRC GaN500v2 and were presented. Here only small signal measurement were presented and discussed. It was observed that, with a single transistor in common gate configuration, the isolation is good at low frequencies but quickly drops with increasing frequency. In addition, it was observed that the more the device is large, the lower the insertion loss achieved but the poorer the isolation will be. A trade-off between insertion loss and isolation has been made

to choose the single transistors that will be used for designing switch structures such as SPST and SPDT which could improve the performances at high frequencies.

Comparing to the standalone device, it was revealed that with SPST structure, the isolation is improved while the insertion loss slightly dropped by -1 dB. With this proposed topology, it is still a challenge to meet good performance at our higher frequency. Another topology based on SPDT was proposed to improve more the performances. As was expected with this topology, some improvement in isolation is achieved compared to SPST, while the insertion loss is increased by almost -1 dB. For example, at 2.45 GHz (frequency of interest) isolation better than -25 dB and an insertion loss of -3 dB were achieved. However, the SPDT structure performances degrade with frequency. Moreover, based on these small signal measurements, a model was introduced. In addition, the fabricated SPDT switch structure will be integrated with antenna system to demonstrate the ability of using GaN devices for a beam switching antenna array.

Finally, in this thesis, the use of GaN on highly efficient reconfigurable designs for broadband RF/microwave applications operating below 10 GHz was demonstrated. The varactors and switches used for tunable purpose were firstly fabricated, characterized and then modeled. The proposed models were used to achieve the reconfigurability of RF/microwave circuits. Tunable phase shifter, reconfigurable 3-dB 90° hybrid coupler, tunable frequency oscillator, beam switching antenna array and matching reconfigurable patch antenna were developed along with the development of GaN modeling varactors and switches, and were presented as applications results of this thesis research.

Using GaN150v0 process, two reconfigurable RF designs have been designed and reported. Firstly, a GaN phase shifter design based on reflective topology has been described, and only simulation results were presented. First, the 3-dB 90° hybrid coupler has been designed and simulated using NRC GaN150 PDK.

Then two developed GaN varactors were placed on the two coupler's output ports as loads. Under different bias conditions varying from -3.5 V to -5.5 V, the return loss of the phase shifter is better than 20 dB across a 1 GHz bandwidth centered at 10 GHz, while good match is achievable over this range of 9.5-10.5 GHz. Worst case of the insertion loss is about -3 dB at 10 GHz and occurs at a bias point of -4.3 V where the equivalent resistor of the used varactors is significantly high. Therefore, a continuously variable phase-shift range of 80° is achieved at 10 GHz. Secondly, a varactor tunable broadband quadrature hybrid coupler has been investigated and simulated from X-band to Ku-band frequencies applications, and only simulated results were reported. First, a static coupler has been designed to meet the specifications of a typical quadrature hybrid coupler (3-dB insertion loss, good return and isolation loss, and 90° phase difference between coupled and through ports) at 10

GHz. Then, a tunable broadband coupler has been designed by inserting varactor diodes as shunt capacitance elements in order to control the operating frequency. This design occupies an area of 1.35×0.9 mm 2 and good simulation results have been demonstrated. This novel varactor reconfigurable hybrid coupler is proposed to overcome the need of high power tunable RF/microwave circuits which are required for multi-standard and multiservice applications. By changing the bias voltage of the varactor to control the resonance condition of parallel LC resonator, this proposed hybrid coupler can provide the desired operating frequency from 10 GHz to 16 GHz.

Over this bandwidth, the overall return loss is better than -13.5 dB over a tuning range 1 V while the isolation is below than -14 dB. In addition, the simulated magnitude of S21 parameter is close to -3.5 dB and is getting poorer as varactor capacitance values decreases. A worst value of -5 dB is achieved at the minimum capacitance value. While at port 3, the S31 parameter behaves as S21 but in a reverse way. As a result, the amplitude imbalance is varying from -0.2 dB at 10 GHz and -2 dB at 16 GHz. A minimum well-balanced quadrature phase of 86° is obtained between the coupled port 2 and the through port 3. This phase is increasing and getting closer to 92 ° as varactor capacitor values are decreasing.

Using GaN500v2 process, a tunable frequency GaN oscillator was designed by loading symmetrically the feedback loop, and was presented. For tuning purpose, the previous developed GaN varactors were used. The oscillation frequency values change significantly and a minimum oscillation frequency of 7.8 GHz was achieved when CEq is at its maximum for VG above -3.5 V, and a maximum oscillation frequency of 8.6 GHz was simulated when CEq is at its minimum for VG bellow -4.5 V. While, when VG varies from -5 V to -3. a slot layer and a simple element patch layer and between them an air-gap layer of 1mm. As presented, the minimum and the maximum values of the simulated input return loss are, respectively, -48 dB and -13 dB over a narrow bandwidth from 9.34GHz to 9.5GHz. This tunability is achieved by integrating a developed varactor with the antenna design. The last reported reconfigurable antenna design is an electronically beam steering antenna using the developed SPDT switch structure is proposed for applications that operate at 2.45 GHz frequency. The idea here is to verify the developed GaN SPDT switch structure fabricated in NRC GaN500v2 process, for this a global system with four patch arrays antenna with beam steering over the H plane for four Φ possibilities such as, 0°, 90°, 180° and 270° is designed.

  Figure II.1: Most used Semiconductor Technologies and their Frequency/Power limits [6] .................. 9

1 : 9 Figure II. 2 :

 192 Figure II.1: Most used Semiconductor Technologies and their Frequency/Power limits [6] .................. 9 Figure II.2: Current due to trapping effect and thermal effect [18] .......................................................

Figure II. 3 :

 3 Figure II.3: A sketch of an AlGaN/ GaN HEMT structure with trapped electrons [19] .......................

Figure II. 4 :

 4 Figure II.4: Cross-Section of HEMTs (Not Drawn to Scale) ...................................................................

Figure II. 5 :

 5 Figure II.5: Cross-sectional view of HFET, showing process sequence (not to scale) [31] ....................

Figure II. 6 :

 6 Figure II.6: Photo of a T/R-Module front-end with GaN MMIC chips [43] ............................................

Figure II. 7 :

 7 Figure II.7: Measured output power versus frequency of the GaN MMIC DA at .................................

Figure II. 8 :

 8 Figure II.8: Measured power performance of the GaN MMIC HPA at 11 GHz ....................................

Figure II. 9 :

 9 Figure II.9: Measured gain and noise figure versus frequency of the GaN MMIC ...............................

Figure

  Figure II.10: (Left) A schematic view and (Right) a photograph of the Fabricated T/R module using HTCC technology [44] ...............................................................................................................................

Figure II. 11 :

 11 Figure II.11: Measured Pout of fabricated T/R module as a function of frequency, biased at 30 V [44] ....................................................................................................................................................................

Figure II. 12 :

 12 Figure II.12: Measured S-parameters of GaN LNA MMIC (Vds = 15 V) [44] ......................................

Figure II. 13 :

 13 Figure II.13: Photograph of the fabricated amplifier [45] .......................................................................

Figure II. 14 :

 14 Figure II.14: Measured PAE versus output power [45] ..........................................................................

Figure II. 15 :

 15 Figure II.15: Measured gain versus output power [45] ...........................................................................

Figure II. 16 :

 16 Figure II.16: The concept of impedance matching conditions of the band and class reconfigurable power amplifier [46] ..............................................................................................................................................

Figure II. 17 :

 17 Figure II.17: Photograph of the fabricated class-reconfigurable power amplifier [46] .........................

Figure II. 18 :

 18 Figure II.18: Simulated gain verses input power at different matching states [46] .............................

Figure II. 19 :

 19 Figure II.19: Simulated output power verses input power at different matching states [46] ..............

Figure II. 20 :

 20 Figure II.20: Simulated power added efficiency verses input power at different matching states [46] ....................................................................................................................................................................

Figure II. 21 :

 21 Figure II.21: (Left) Top and cross sectional view and (Right) Picture of the packaged PA [47] ..........

Figure II. 22 :

 22 Figure II.22: Acetone configuration: measurements at 2.3 GHz, simulations at 2. 4 GHz [47] ...........

Figure II. 23 :

 23 Figure II.23: Air configuration: measurements at 5.7 GHz, simulations at 5.8 GHz [47] ....................

Figure III. 1 :

 1 Figure III.1: Work Plan .............................................................................................................................

Figure III. 2 :

 2 Figure III.2: C-V characteristics of devices with different electrode length (spacing = 1 um, width = um) [53] ......................................................................................................................................................

Figure III. 3 : 40 Figure III. 6 : 40 Figure III. 7 : 41 Figure III. 8 : 46 Figure III. 15 : 51 Figure III. 22 : 52 Figure III. 23 : 53 Figure III. 24 : 53 Figure III. 25 : 55 Figure III. 26 : 61 Figure III. 32 : 62 Figure III. 33 : 63 Figure III. 34 : 64 Figure III. 35 : 76 Figure III. 44 : 83 Figure III. 48 : 84 Figure III. 49 : 84 Figure III. 50 : 90 Figure III. 57 : 95 Figure IV. 2 : 100 Figure IV. 5 : 101 Figure IV. 6 : 101 Figure IV. 7 : 101 Figure IV. 8 : 102 Figure IV. 9 : 103 Figure IV. 10 : 107 Figure IV. 15 : 107 Figure IV. 16 :

 340640741846155122522353245325552661326233633464357644834884498450905795210051016101710181029103101071510716 Figure III.3: Extracted capacitance (black line) and resistance (white line) of the MSM varactor over the entire bias range at 1.1 GHz. The inserted schematic shows the first-order equivalent circuit of the MSM varactor [54] .....................................................................................................................................

17

 17 Figure IV.17: An Equivalent Circuit to Estimate LS, RS and Fres [105] ............................................ 113

113 Figure IV. 18 : 113 Figure IV. 19 :

 1131811319 Figure IV.17: An Equivalent Circuit to Estimate LS, RS and Fres [105] ............................................ 113 Figure IV.18: Layout of A MIM Capacitor [105] .................................................................................... 113 Figure IV.19: Layout of the designed hybrid coupler ............................................................................ 114 Figure IV.20: 3-D Layout view of the designed hybrid coupler ............................................................ 115 Figure IV.21: Insertion Loss versus frequency ...................................................................................... 116 Figure IV.22: Return Loss versus frequency ......................................................................................... 116 Figure IV.23: Isolation versus frequency ............................................................................................... 116

Figure IV. 25 :

 25 Figure IV.25: Layout of the designed Phase Shifter .............................................................................

Figure IV. 26

 26 Figure IV.26: 3-D Layout view of the designed Phase Shifter .............................................................

Figure IV. 27 :

 27 Figure IV.27: Return Loss at several DC Bias from -3.5 V to -5.5 V ...................................................

Figure IV. 28 :

 28 Figure IV.28: Insertion Loss of the Phase Shifter at several DC Bias from -3.5 V to -5.5 V ..............

Figure IV. 29 :

 29 Figure IV.29: Phase Shift versus DC Bias at 10 GHz ...........................................................................

Figure IV. 30 :

 30 Figure IV.30: Layout of the designed hybrid coupler ............................................................................

Figure IV. 31

 31 Figure IV.31: 3-D Layout view of the designed hybrid coupler ............................................................

Figure IV. 32 :

 32 Figure IV.32: Return Loss at several DC Bias from -3.7 V to -4.8 V ...................................................

Figure IV. 33 :

 33 Figure IV.33: Isolation at several DC Bias from -3.7 V to -4.8 V .........................................................

Figure IV. 34 :

 34 Figure IV.34: Level of S21 at several DC Bias from -3.7 V to -4.8 V ...................................................

Figure IV. 35 :

 35 Figure IV.35: Level of S31 at several DC Bias from -3.7 V to -4.8 V ...................................................

Figure IV. 36 :

 36 Figure IV.36: Phase Difference at several DC Bias from -3.7 V to -4.8 V ............................................

Figure IV. 37 :

 37 Figure IV.37: Block diagram of a feedback oscillator ............................................................................

Figure IV. 38 :

 38 Figure IV.38: Photo of the CGHV1J025D GaN Transistor Die ............................................................

Figure IV. 39 :

 39 Figure IV.39: 3D Bond wire model. ........................................................................................................

Figure IV. 40 :

 40 Figure IV.40: Simulated I-V Curves of the transistor ...........................................................................

Figure IV. 41

 41 Figure IV.41: K-factor and Delta ............................................................................................................

Figure IV. 42 :

 42 Figure IV.42: A typical distributed bias network with radial stub ......................................................

Figure IV. 43 :

 43 Figure IV.43: Fabricated PA under Test ................................................................................................

Figure IV. 44 :

 44 Figure IV.44: Simulated Small Signal S-Parameters ...........................................................................

Figure IV. 45 :

 45 Figure IV.45: Measured Small Signal S-Parameters ............................................................................

Figure IV. 46 :

 46 Figure IV.46: Ids of 260 mA at Vds of 18.6 V and Vgs of -2.5 V ...........................................................

Figure IV. 47 :

 47 Figure IV.47: Fixed Frequency Oscillator Fabricated Prototype .........................................................

Figure IV. 48 :

 48 Figure IV.48: Small Signal Simulation Results (Open-Loop) ...............................................................

Figure IV. 49 :

 49 Figure IV.49: Harmonics Power Simulation of the FFO .......................................................................

Figure IV. 50 :

 50 Figure IV.50: Pure Voltage Sinusoid at the Output of the FFO ...........................................................

Figure IV. 51 :

 51 Figure IV.51: Pure Current Sinusoid at the Output of the FFO ..........................................................

Figure IV. 52 :

 52 Figure IV.52: Simulated Phase Noise of the FFO .................................................................................

Figure IV. 53 :

 53 Figure IV.53: Harmonics Power Measurement of the FFO ..................................................................

Figure IV. 54 :

 54 Figure IV.54: Tunable Frequency Oscillator fabricated Prototype ......................................................

Figure IV. 55 :

 55 Figure IV.55: Oscillation frequency versus gate voltage ......................................................................

Figure IV. 56 :

 56 Figure IV.56: Output power versus gate voltage at fundamental frequency ......................................

Figure IV. 57 :

 57 Figure IV.57: Phase Noise versus gate voltage at 10 KHz, 100 KHz, and 1 MHz ..............................

Figure IV. 58 :

 58 Figure IV.58: Printed antenna prototype with GaN varactor load. .....................................................

Figure IV. 60 :

 60 Figure IV.60 : Simulated antenna gain versus frequency ....................................................................

Figure IV. 61 :

 61 Figure IV.61 : Simulated normalized radiation pattern of the antenna at 11 GHz ............................

Figure IV. 62 :

 62 Figure IV.62: Simulated return loss of the loaded antenna for different reverse-bias voltages ........

Figure IV. 63 :

 63 Figure IV.63: Measured return loss of the loaded antenna for different reverse-bias voltages .........

Figure IV. 64 :

 64 Figure IV.64 : Measured loaded antenna gain versus frequency for various DC bias ........................

Figure IV. 65 :

 65 Figure IV.65 : Measured radiation pattern in H-plane of the loaded antenna under various DC bias ..................................................................................................................................................................

Figure IV. 66 :

 66 Figure IV.66 : Measured radiation pattern in E-plane of the loaded antenna under various DC bias ..................................................................................................................................................................

Figure IV. 67 :

 67 Figure IV.67 : (left) patch antenna (right) slot feeding loaded with GaN varactor .............................

Figure IV. 68 :

 68 Figure IV.68 : Stuck-up whole antenna system structure ....................................................................

Figure IV. 69 :

 69 Figure IV.69 : Simulated return loss of the antenna alone ..................................................................

Figure IV. 70 :

 70 Figure IV.70 : Simulated antenna gain versus frequency ....................................................................

Figure IV. 71 :

 71 Figure IV.71 : Simulated normalized radiation pattern of the antenna at 11.35 GHz .......................

Figure IV. 72 :

 72 Figure IV.72: Simulated reflection coefficient (S11) of the loaded antenna for different varactor reverse-bias voltages ...............................................................................................................................

Figure IV. 73 :

 73 Figure IV.73: Measured return loss (S11) of the loaded antenna at no DC bias applied ...................

Figure IV. 74 :

 74 Figure IV.74 : Measured loaded antenna gain versus frequency at no bias DC applied ....................

Figure IV. 75 :

 75 Figure IV.75 : Measured normalized radiation pattern of the loaded antenna at 10.6 GHZ with no bias DC applied ................................................................................................................................................

Figure IV. 76 :

 76 Figure IV.76: 3D model of designed antenna system. ...........................................................................

Figure IV. 77 :

 77 Figure IV.77: Fabricated prototype of the antenna ...............................................................................

Figure IV. 78 :

 78 Figure IV.78: The feeding system including the three SPDT switch structures .................................

Figure IV. 79 :

 79 Figure IV.79: Zoom on SPDT_1 switch ..................................................................................................

Figure IV. 80 :

 80 Figure IV.80: Zoom on the wires bonding connecting the SPDT_1 to The feeding system ................

Figure IV. 81 :

 81 Figure IV.81: Front view of the fabricated antenna ..............................................................................

Figure IV. 82 :

 82 Figure IV.82: Simulated return loss of the antenna .............................................................................

Figure IV. 83 :

 83 Figure IV.83: Simulated antenna gain versus frequency .....................................................................

Figure IV. 84 :

 84 Figure IV.84: Simulated normalized H-plane at 2.43 GHz for four cases mentioned in Table IV.4 ..

Figure IV. 85 :

 85 Figure IV.85: Simulated normalized E-plane at 2.43 GHz ...................................................................

Figure IV. 86 :

 86 Figure IV.86: Measured return loss of the antenna ..............................................................................

Figure IV. 87 :

 87 Figure IV.87: Measured antenna gain versus frequency ......................................................................

Figure IV. 88 :

 88 Figure IV.88: Measured normalized H plane at 2.43 GHz for four cases mentioned in Table IV.4 .

Figure IV. 89 :

 89 Figure IV.89: Measured normalized E plane at 2.43 GHz ....................................................................

Figure IV. 91 :

 91 Figure IV.91: Measured S-parameters for case_2 at Pin of 0 dBm ......................................................

Figure IV. 92 :

 92 Figure IV.92: Measured S-parameters for case_3 at Pin of 0 dBm ......................................................

Figure IV. 93 :

 93 Figure IV.93: Measured S-parameters for case_4 at Pin of 0 dBm ......................................................

Figure IV. 94 :

 94 Figure IV.94: Measured insertion loss versus input RF power for case_1 at 2.43 GHz .....................

2 :

 2 Parameters values of the equivalent circuit ........................................................................ 49 Table III.3: Comparison of capacitance performances between GaN500 and GaN150 ........................ 58 Table III.4 : Presented GaN500v2 Devices Information ......................................................................... 58 Table III.5: Parameters values of the equivalent circuit ........................................................................ 59 Table III.6: Model Circuit Parameters Values ........................................................................................ 67 Table III.7: Model Circuit Parameters Values ........................................................................................ 85 Table III.8: Performances of the SPDT at the Marker m4 ..................................................................... 91 Table IV.1: Optimized Values of Inductors L1 and L2 .......................................................................... Table IV.2: Optimized Values of Capacitors C1 and C2 ....................................................................... Table IV.3: Dimensions of Stubs............................................................................................................. Table IV.4: Predicted different impedances of the SPDT ports. ........................................................... Table IV.5: Switching configuration to achieve radiation pattern reconfigurability. ......................... effect transistor HEMT High Electron Mobility Transistors 2DEG two-dimensional electron gas HBT Heterojunction bipolar transistor HEMT High electron mobility transistor HVPE Hydride vapour phase epitaxy LNA Low-noise amplifier PA Power Amplifier MBE Molecular beam epitaxy MEMS Micro-electromechanical systems MIM Metal-insulator-metal MMIC Monolithic microwave integrated circuit MOCVD Metal-organic chemical vapour deposition NRC National Research Council Canada PCB Printed circuit board RF Radiofrequency SPDT Single-pole-double-throw SPST Single-pole-single-throw AIA Active Integrated Antennas هلل (Alhamdulillah). Seldom anything is achieved without help, support, guidance, encouragement, love and care of other people. I am fortunate to have such people around me who have made my journey during my doctoral studies fruitful, rewarding and enjoyable. Firstly, I am particularly indebted to my supervisors Pr. Mohammed HIMDI from University of Rennes 1 and Pr. Langis ROY from Carleton University, and my co-supervisor, Dr. Olivier LAFOND from University of Rennes 1, for the unconditional support, valuable guidance and endless patience they generously provided during my thesis research. I must also note the financial support of my advisors. I am also grateful to Pr. Rony AMAYA and Pr. Garry TARR for their useful advice when I start dealing with GaN HEMT devices. I am also grateful to Dr. Ibrahim HAROUN from NRC for giving me the opportunity to get the possibility of an internship in NRC and to work side by side with the NRC GaN group team. I have gained valuable insight into the GaN technology. I graciously thank Pr. Steve McGARRY, Pr. Mustapha YAGOUB, Pr. Nathalie ROLLAND and Dr. Xvier CASTEL for accepting to be a part of my PhD thesis committee members. I wish to express my sincere thanks to all staff of Laboratory IETR and the department of electronics at Carleton University, for providing me with all the necessary facilities for the research. Special thanks go to Blazenka POWER, Anna LEE and Mikhail NAGUI from Carleton University, and also Noelle LE BER from University of Rennes 1 for their previous helps.

a

  brief comparison, Figure II.1 recapitulates the most used semiconductor technologies and their limits in terms of power handling capability and operating frequency.

Figure II. 1 :

 1 Figure II.1: Most used Semiconductor Technologies and their Frequency/Power limits[START_REF] Callet | Caractérisation et Modélisation de Transistors HEMT AlGaN/GaN et InAlN/GaN pour l'Amplification de puissance en Radio-Fréquences[END_REF]

  collapse and rolling off as shown in Figure II.2. This existence, of dispersion effects observed in GaN devices, acts to restrict the drain current and voltage excursions, thereby limiting the initial expectations of device's performances in term of high power and high frequency.

Figure II. 2 :

 2 Figure II.2: Current due to trapping effect and thermal effect[START_REF] Rüdiger | Gallium Nitride Electronics[END_REF]

Figure II. 3 :

 3 Figure II.3:A sketch of an AlGaN/ GaN HEMT structure with trapped electrons[START_REF] Md | Mechanism and Suppression of Current Collapse in AlGaN/GaN High Electron Mobility Transistors[END_REF]

13 themselves

 13 out of the channel, causing significant roll off in the high voltage and high current region, which is observed in DC current variations, as shown in Figure II.2. The simplified vertical semiconductor structure of a GaN/AlGaN HEMTs is illustrated Figure II.4.

Figure II. 4 :

 4 Figure II.4: Cross-Section of HEMTs (Not Drawn to Scale)

  Figure II.5 are reproduced with permission of the NRC.

Figure II. 5 :

 5 Figure II.5: Cross-sectional view of HFET, showing process sequence (not to scale)[START_REF]Gallium Nitride MMIC Foundry Design Manual GaN500v2[END_REF]

Figure II. 6

 6 figure of the low noise amplifier (LNA) is in the range of 1.5 dB as shown in Figure II.9.

Figure II. 6 :Figure II. 8 :

 68 Figure II.6: Photo of a T/R-Module front-end with GaN MMIC chips[START_REF] Schuh | GaN MMIC based T/R-Module Front-End for X-Band Applications[END_REF]

  low-noise amplifier (LNA) with a gain of 15.9 dB over 3.2-20.4 GHz. Figure II.10 shows the fabricated T/R module, while the Figure II.11 and Figure II.12 illustrate, respectively, the measured Pout of fabricated T/R a function of frequency and the measured S-parameters of GaN LNA MMIC.

Figure II. 10 :

 10 Figure II.10: (Left) A schematic view and (Right) a photograph of the FabricatedT/R module using HTCC technology[START_REF] Masuda | C-Ku band GaN MMIC T/R frontend module using multilayer ceramics technology[END_REF]

Figure II. 13 :

 13 Figure II.13: Photograph of the fabricated amplifier[START_REF] Gustafsson | A GaN MMIC Modified Doherty PA With Large Bandwidth and Reconfigurable Efficiency[END_REF]

  Figure II.16 and Figure II.17 show, respectively, the concept of impedance matching conditions and a topology of the photograph of the proposed reconfigurable PA. The proposed PA can be reconfigured to operate in class A or class B to achieve, respectively, an extreme high linearity and high efficiency where high linearity is not strictly required. As a result, the proposed reconfigurable PA design can be reconfigured for multiple bands and for switching between operating classes. This reconfigurability is clearly seen in Figure II.18, Figure II.19 and Figure II.20 where gain, output power and PAE versus the swept input power, respectively, are plotted.

Figure II. 16 : 29 Figure II. 19 :

 162919 Figure II.16: The concept of impedance matching conditions of the band and class reconfigurable power amplifier[START_REF] Jantarachote | Frequency band and operating class reconfigurable GaN HEMT power amplifier for sustainable wireless communications and energy applications[END_REF]

Figure II. 21 : 4 GHz [ 47 ]Figure II. 23 :

 2144723 Figure II.21: (Left) Top and cross sectional view and (Right) Picture of the packaged PA[START_REF] Chlieh | Microfluidically reconfigurable GaN power amplifier on multilayer organic substrate for s-band and c-band applications[END_REF]

  GaN varactor devices and GaN switch devices, as outlined in the "Work Plan" chart below (see FigureIII.1). The design of these GaN components was realized using the Canadian National Research Council (NRC) GaN HEMTs monolithic microwave integrated circuit (MMIC) processes. Three NRC GaN HEMTs processes, such as, GaN150v0 with a gate length of 0.15 um, GaN500v1 with gate length of 0.5 um and GaN500v2 which is the developed version of GaN500v1, have been used to manufacture the varactor and the switch devices.An accurate large signal model requires extensive laboratory equipment to extract all the effects related to high power GaN devices, such as thermal and trapping effects typically present in GaN devices. Carleton University is not as well equipped with these laboratory facilities as are the NRC or IETR. Due to our incapability to have large signal measurement facilities, only small signal measurement will be presented and discussed. However, for varactors fabricated 35 using GaN500v2, we finally succeed to do large signal measurements. These measurements have been done at XLIM at University of Limoges, France.

Figure III. 1 :

 1 Figure III.1: Work Plan

  voltage varactors diodes. Up to now, the progress on developing GaN varactors have been limited and only a few works have been already reported. Based on GaN HEMT technology, some varactors have previously been reported, mostly employing a Metal-Semiconductor-Metal (MSM) junction[START_REF] Marso | MSM diodes based on an AlGaN/GaN HEMT layer structure for varactor and photodiode application[END_REF]-[START_REF] Chu | Q-factor characterization of RF GaN-based metal-semiconductor-metal planar interdigitated varactor[END_REF]-[START_REF] Chang | Fast surge suppression by inserting an AlGaN/GaN-based varactor[END_REF]-[START_REF] Tien | Capacitance Swing and Capacitance Ratio of GaN-Based Metal-Semiconductor-Metal Two-Dimensional Electron Gas Varactor with Different Dielectric Films[END_REF] or Schottky barrier contact[START_REF] Jin | A novel GaN-based high frequency varactor diode[END_REF]-[START_REF] Lu | InGaN/GaN Schottky Diodes With Enhanced Voltage Handling Capability for Varactor Applications[END_REF]-[START_REF] Lu | Critical design considerations for GaN-based microwave power varactors[END_REF].Later 2002, one of the first investigated GaN varactor work was reported in[START_REF] Marso | MSM diodes based on an AlGaN/GaN HEMT layer structure for varactor and photodiode application[END_REF]. This work investigates the properties of MSM diodes that are based on an AlCaN/GaN HEMT layer structure (MSM-2DEG), which shown that devices made in this material system are realizable with tuning capacitance range. As can be seen from Figure III.2, the capacitance is constant between 2.7 and -2.7V and within a very small voltage range of about 0.3 V the capacitance starts getting changed. The reported measured CMAX/CMIN ratios vary from 5 to 100 and were dependent on electrode geometry. This investigation shown clearly that these ratio values were exceeding those ratios for varactor diodes in GaAs. However, a high series resistance varies from 40 Ω to 70 Ω was extracted and these values were dependent on the electrode spacing from 0.5 um to 2 um. This high series resistance of the device, which was announced as a challenge for varactors diodes, can affect directly the figure of merit, quality factor (Q-factor), of the device. In addition, this high value of series resistors is due to the unavoidable ohmic contact resistance. Researchers have begun to investigate the possibilities in order to reduce this specific contact resistance.

Figure III. 2 :

 2 Figure III.2: C-V characteristics of devices with different electrode length (spacing = 1 um, width = 50 um)[START_REF] Marso | MSM diodes based on an AlGaN/GaN HEMT layer structure for varactor and photodiode application[END_REF]

Figure III. 3 : 39 Figure

 339 Figure III.3: Extracted capacitance (black line) and resistance (white line) of the MSM varactor over the entire bias range at 1.1 GHz. The inserted schematic showsthe first-order equivalent circuit of the MSM varactor[START_REF] Chu | Q-factor characterization of RF GaN-based metal-semiconductor-metal planar interdigitated varactor[END_REF]

Figure

  Figure III.5: A Schematic diagram of the AlGaN/GaN-based varactor[START_REF] Chang | Fast surge suppression by inserting an AlGaN/GaN-based varactor[END_REF]

Figure III. 7 :

 7 Figure III.7: Schematic diagram of the AlGaN/GaN MSM-2DEG varactor of the samples prepared in this study[START_REF] Tien | Capacitance Swing and Capacitance Ratio of GaN-Based Metal-Semiconductor-Metal Two-Dimensional Electron Gas Varactor with Different Dielectric Films[END_REF]

  expected from such GaN-based Schottky barrier varactor diode. The structure used in this work consists of a 2 μm bottom ohmic (n + ) contact layer and a 300 nm lightly doped (n -) GaN layer which is grown on top of the ohmic layer. The ohmic contacts (Ti/Al/Ti/Au) were deposited once the mesa mask was isolated by Ar etching and then Pt was deposited which forms the Schottky contact metal. The schematic of the GaN Schottky varactor structure is shown in Figure III.9. It was shown that epilayer designs can help in controlling such characteristics. As shown in Figure III.10, the maximum capacitance at 0 V bias is 40.476 fF and the minimum capacitance is about 26.4fF when a bias of less than -7 V is applied. This reported varactor had a breakdown voltage of 20 V and a cut-off frequency of 360 GHz.

Figure III. 9 :

 9 Figure III.9: Schematic Cross-sectional view of the GaN Schottky Varactor[START_REF] Jin | A novel GaN-based high frequency varactor diode[END_REF]

Figure III. 11

 11 Figure III.11 schematically shows the structure of a GaN varactor. Furthermore, the depletion width W varies in the n-region under several external bias voltage, this resulting to a change in equivalent C as well as in its associated series resistors R.

  Figure III.12 shows the C-V curves over the range of 0-100 V for the InGaN-containing samples and for a control sample.

Figure III. 11 :

 11 Figure III.11: Schematic diagram of a GaN-based varactor and its layer structure[START_REF] Lu | InGaN/GaN Schottky Diodes With Enhanced Voltage Handling Capability for Varactor Applications[END_REF]

  Figure III.13 shows the simulated C-V curves for single and two anti-parallel structures, while in Figure III.14, Q-factor versus the reverse external bias is depicted.

Figure III. 13 :

 13 Figure III.13: Capacitance-voltage curves for (a) single diode structure and (b) twoanti-parallel diode structure[START_REF] Lu | Critical design considerations for GaN-based microwave power varactors[END_REF]

Figure III. 15

 15 Figure III.15 illustrates the layout design for one varactor device. To compare between the two processes (GaN500v1 and GaN150v0), two samples of each process with different fingers width are studied as detailed in Table III.1:

Figure III. 15 :

 15 Figure III.15: Schematic cross-section of HEMT, showing contacts and epitaxial structures, and physical equivalent circuit origin at pinch-off (VG = -3.5 V)

Figure III. 16 :

 16 Figure III.16: W40L500v1 Device in Microwave Probe Test

Figure III. 17 :

 17 Figure III.17: Varactor equivalent circuit model

Figure III. 18 : 53 Figure III. 23 :

 185323 Figure III.18: Measured (lines) and Simulated (Dot) S11-parameter for G W40L0.5

Figure III. 25 :

 25 Figure III.25: Exctacted and Simulated REq versus VG at 2.5 GHz for GaN500v1

  Figure III.27: Extracted Q factor versus Frequency at VG = -3.5 V for GaN500v1

  As can be 59 seen in Figure III.29, Figure III.30 and Figure III.31, a good match between the experimental results and the simulation model results is generally achieved. For W100L0.5 and W200L0.5, agreement is nearly perfect, while for W50L0.5 there are some issues with this device at high frequencies. Here some parasitic inductances are not captured in the simple proposed equivalent model. As detailed previously, there are three main ranges that describe the change in the parameters of the varactor model. Range (1) and range (3) corresponding to the device's CMAX and CMIN, respectively, while, range (2) corresponding to the tuning capacitance range. Table

Figure III. 29 : 5 Figure 61 Figure III. 31 : 5 4. 1 . 1

 2956131511 Figure III.29: Measured (lines) and Simulated (Dot) S11-parameter for W50L0.5

Figure III. 32 :

 32 Figure III.32: CEq versus frequency for the W50L0.5 device at VG =-3V

Figure III. 33 :

 33 Figure III.33: Extracted and Simulated CEq versus VG for W50L0.5

Figure III. 34 : 5 Figure III. 35 :

 34535 Figure III.34: Extracted and Simulated CEq versus VG for W100L0.5

Figure III. 36 :

 36 Figure III.36: Extracted REq versus VG for GaN500v2 Devices

frequency of 5

 5 GHz. The measurement method consists in performing periodic CW time-domain large-signal load-pull RF measurements using Large-Signal Network Analyzer (LSNA). The LSNA can measure the harmonics of the high voltages and high currents at the DUT planes from 1GHz up to 18 GHz with a dedicated test bench. Thanks to on-wafer absolute and relative calibrations with absolute phase reference of LSNA, the time-domain RF voltages and currents (at fundamental and harmonic frequency [f0, 2f0 and 3f0]) and periodic CW power performance are obtained at the DUT planes. The choice of this frequency of 5 GHz was made for the reason that the equivalent capacitance versus frequency (as shown in Figure III.32) do not change up

Where A is the

  finger area. The parameters B, C, F, G, H, I and K were adjusted to properly fit the CEq equation to the measured characteristics of an arbitrary varactor size. According to Matlab simulations, the values of these empirical coefficients can be fixed for all devices, respectively, to 67, 44.4, 4.1, 1.5, 0.0565, 0.4 and 2.8. Equation (8) is dependent only on three main parameters, which are: finger geometry (number, width and length of fingers), bias voltage (VG) and RF power (Pin). Due to their small size and to avoid breaking down the devices, we did not use any driver power amplifier to apply larger RF signals. It was noticed that above 5 dBm, the value of CEq starts being affected by the power level.

Figure III. 37 ,

 37 Figure III.37, Figure III.38 and Figure III.39 show the measured and the modeled equivalent capacitance values versus VG for various RF power levels (Pin),respectively, for W50L0.5, W100L0.5 and W200L0.5. It can be seen that all CEq-VG curves exhibit the same general behavior. Under small power levels up to 5 dBm, this behavior can be separated into three regions which are range (1) for VG above -3.4V, range (2) for -4.5V≤ VG ≤-3.4V and range (3) for VG < -4.5 V. However, more the power level increases more range (2) is extended. For range (1) and range[START_REF] Johannsen | Modular antenna array concept for millimeter-wave beam-steering applications[END_REF] corresponding to the device's maximum (CMAX) and minimum (CMIN) values, the capacitance values do not change significantly versus VG. In range (2), the capacitance values change appreciably and rapidly over a small DC bias range from -4.5 V to -3.4 V. This bias range extends as well as the power is increased. However, the CMAX and the CMIN values, respectively, decrease and increase as well while the input power increases. This means that the CMAX/CMIN ratio drops significantly, resulting in device saturation and non-linearity at high power levels.

Figure III. 37 : 5 Pin 71 Figure III. 38 :

 3757138 Figure III.37: Extracted (left) and Simulated (right) CEq versus VG for various power levels [-10, 18] dBm for device W50L0.5

Figure III. 40 :Figure III. 41 : 5 Figure III. 42 : 5 Psrc

 40415425 Figure III.40: Extracted REq versus VG for various power levels [-10, 18] dBm for device W50L0.5 (left), for W100L0.5 (middle) and for W200L0.5 (right)

  GaN500v2 and GaN150v0 have lower series resistor value than those value reported with GaN500v1 devices. Two analytic equations describing the behavior of equivalent capacitance and series resistor versus bias voltage were proposed. These equations are only bias voltage and device geometry dependency.Large signal analyses have been done only for devices manufactured withGaN500v2 process. The influence of RF power has been taken into account and an analytic expression was proposed to approximately describe equivalent capacitance versus bias voltage for various power levels. This equation accurately describes the dependency of junction capacitance upon applied reverse bias as well as upon RF power for all characterized devices. This modeling work has led to one published journal paper[START_REF] Hamdoun | Characterisation and analytical modelling of GaN HEMT-based varactor diodes[END_REF] and one other submitted to IEEE Transactions on Device and Materials Reliability in August 75 2016 [151]. A conference paper has been also presented in European Microwave Conferences in Paris in 2015 [150]. III. GaN-based Switch Devices Modeling 1. Review of GaN HEMT for Power Switching DevicesApplicationsAs previously mentioned in the introduction section, GaN-based switch components is a part of the proposed thesis research. The main aim of this part is to conduct a complete modeling GaN-based switch study using a GaN HEMT process from NRC. Our investigation was planned to consider both small and large signal analysis. Here in this section, different sizes single standalone switches and switch structures such as single pole single throw (SPST) and single pole double throw (SPDT) were fabricated using NRC GaN500v2 and will be presented. Due to our incapability to have large signal measurement facilities, only small signal measurement will be presented and discussed. An accurate large signal model requires extensive laboratory equipment to extract all the effects related to high power GaN devices, such as thermal and trapping effects typically present in GaN devices. Carleton University is not as well equipped with these laboratory facilities as are the NRC or IETR.

  resistor and capacitor in the path of the RF signal, as shown in Figure III.43. However, in practice a more complex model is used to model the GaN HEMTs-based switch. Subsection d explains in detail the modeling of the GaN switch.

Figure III. 43 :

 43 Figure III.43: Simple Model for a HEMT in the switch configuration (a) ON-state and (b) OFF-state mode

Figure III. 44 : 81 Figure

 4481 Figure III.44: Linear model used to model the switching GaN HEMT transistors (4x40) [[90]

3. 1

 1 Different Sizes Standalone Signal Switch Devices Here the conventional measurements setup of a transistor in common gate configuration is described. The simplified schematic presented in Figure III.46 (left) is used as a test bench to characterize the standalone switches while the Figure III.46 (right) illustrates the manufactured design for one switch device. As the ability to control the current through the switch should be independent of the current flowing, then it is recommended to put a strong gate resistance, Rbias, for isolation between the control and RF signals. The value of this resistor was chose to be 10 KΩ. The onwafer RF measurements were performed with an Agilent VNA Network Analyzer over a frequency range of 0.1-10 GHz. The control voltage at the gate, VG, was switched between 0 V and -5 V to control the switch, respectively, for ON and OFF states.

Figure III. 46 :

 46 Figure III.46: Schematic (left) of Common Gate Configuration standalone switch 2x200_GaN500v2 and Device in Microwave probes measurement setup (right) to Evaluate RF behavior

Figure III. 47 :Figure III. 48 :Figure III. 50 :

 474850 Figure III.47: Measured insertion loss of different transistor size in common gate, biased with VG=0V (ON state)

Figure III. 51 :

 51 Figure III.51: Linear model used to model the switching GaN HEMT transistors.

Figure III. 52 :FreqFigure III. 53 :

 5253 Figure III.52: Measured and Modeled Insertion Loss and Return Loss (S11) of switch 2x25 biased with VG=0V (ON state)

FreqFigure

  Figure III.54: Schematic (left) and Layout (right) of the Proposed SPST Switch Structure

89 Figure

 89 Figure III.55: Measured SPST performances

Figure III. 56

 56 Figure III.56 illustrates the SPDT topology. If the Port_1 to Port_2 path is ON and the Port_1 to Port_3 path is OFF, then FET1 and FET4 are biased ON, while FET2 and FET3 are biased OFF.

FreqFigure

  Figure III.56: Schematic (left) and Photograph (right) of the Proposed SPDT Switch Structure

  be built based on the small signal model of a single standalone switch. Here as shown in Figure III.57, only small signal measurements of this structure are reported.

Figure III. 57 :

 57 Figure III.57: Measured and Modeled Return Loss, Isolation, and Insertion Loss of SPDT Switch Structure

Figure IV. 1 :

 1 Figure IV.1: (left) Tri-band system with three fixed duplexers and (right) with asingle tunable duplexer[START_REF] Manuel | Composants innovants reconfigurables par voie optique pour des applications microondes[END_REF]



  Semiconductor varactor diodes or switches  Micro-electromechanical systems MEMS During the last years, Monolithic Microwave Integrated Circuits (MMICs)

99 Figure IV. 2 : 100 Using

 992100 Figure IV.2: Simplified RF front-end for a typical beamforming system in a non-GaN technology [[30]

Figure IV. 4 :

 4 Figure IV.4: Photograph of a novel GaN HEMT based high power high-pass/lowpass phase shifter[START_REF] Hettak | A new type of GaN HEMT based high power high-pass/low-pass phase shifter at X band[END_REF]

Figure IV. 8 : 103 Figure IV. 9 :

 81039 Figure IV.8: Microphotograph of the 0°/22.5° GaN switched-filter phase shifter [100]

Figure IV. 11 105 Figure

 11105 Figure IV.11: T-Phase shifter Circuit Configuration and microscope image [101]

Figure IV. 14 :

 14 Figure IV.14: Phase shifter based on 3-dB/90• branch-line hybrid coupler using distributed elements

2. 1

 1 Quadrature 3-dB 90° Hybrid Coupler Quadrature 3-dB 90° hybrid couplers are usually realized in some type of microstrip or stripline technology using distributed quarter-wavelength (λ/4) transmission lines (see Figure IV.14). They have been extensively progressed because of their good bandwidth of operation and of its straightforward design and reliability.

Figure IV. 19

 19 Figure IV.19 illustrates the capacitor layout.

Figure IV. 16 : 114 For

 16114 Figure IV.16: Layout of a Spiral Inductor Generated from NRC GaN150 PDK [105]

Figure IV. 19 :

 19 Figure IV.19: Layout of the designed hybrid coupler

Figure IV. 21 :

 21 Figure IV.21: Insertion Loss versus frequency

Figure IV. 25 :

 25 Figure IV.25: Layout of the designed Phase Shifter

Figure IV. 28

 28 Figure IV.[START_REF] Hashimoto | Improvement of crystal quality in ammonothermal growth of bulk GaN[END_REF] shows the insertion loss of the phase shifter in different bias level from -3.5 V to -5.5 V. At 10 GHz worst case of the insertion loss is about -3 dB and occurs at a bias point of -4.3 V. This value could be due to the higher series resistor value, 9 Ω, of the used varactor (i.e W100L0.15) at this bias voltage as reported in Chapter III (see FigureIII.26). The amplitude imbalance between the two extreme DC bias states (i.e -3.5 V and -5.5 V) is about 1.07 dB at 10 GHz, and is therefore better than 1.2 dB over the extended band from 9.5 to 10.5 GHz.

Figure IV. 29

 29 Figure IV.29 presents the phase shift obtained from the circuit versus DC bias at 10 GHz. Therefore, a continuously variable phase-shift range of 80° is achieved as shown. This phase shift range versus DC bias exhibits the same general behavior as for equivalent capacitor of varactor versus DC bias as discussed in Chapter III in Figure III.24, which can be separated into three regions: range (1) for VG above -3.5 V corresponding to the minimum phase shift value (-80°), range (3) for VG < -5 V corresponding to the maximum phase shift value (0°) and range (2) for -5 V ≤ VG ≤ -3.5 V where the phase shift values change significantly and rapidly.

Figure IV. 27 :

 27 Figure IV.27: Return Loss at several DC Bias from -3.5 V to -5.5 V

  previous presented 10 GHz GaN hybrid coupler, a frequency reconfigurable 3-dB 90° hybrid coupler is designed. In order to achieve frequency reconfigurability, varactors diodes are used instead of the four fixed shunt lumped capacitors C1 of the previous static frequency hybrid coupler (see FigureIV.19). In this configuration, an electronically frequency tuned is achieved by changing the bias voltage applied across the varactors. This means that changing the resonance frequency of branch-line (LC parallel resonator) hybrid of the coupler, we can provide the different operating frequency bands. Additionally, the tuned frequency of the coupler is defined as the frequency at which high isolation and low return loss are achieved, while realizing phase shift of 90° between the isolated and the coupled ports. The novel tunable GaN hybrid coupler employs the developed GaN varactors elements to provide the reconfigurability from 10 GHz to 16 GHz.[START_REF] Sanabria | A GaN differential oscillator with improved harmonic performance[END_REF]

124 2 .

 2 Design of Reconfigurable 3-dB/90° CouplerIn this context, here we report a continuous broadband frequency tunable GaN hybrid coupler. The achieved tuning range is from 10 GHz to 16 GHz corresponding to a bandwidth of 46%. As far as the authors know, this proposed frequency reconfigurable hybrid coupler is the first reported work that uses GaN HEMTs technology. Figure IV.30 and Figure IV.31 show the tunable layout design version of the coupler, where the shunt MIM capacitors are replaced by the developed GaN varactors model. The varactors provide the function of frequency tunability by adjusting their voltage level. This varactor device, W200L0.15, is a two fingers and 200 um gate length and is fabricated using GaN150 process.

Figure IV. 30 :

 30 Figure IV.30: Layout of the designed hybrid coupler

Figure IV. 32 : 128 Figure

 32128 Figure IV.32: Return Loss at several DC Bias from -3.7 V to -4.8 V

Figure IV. 37 :

 37 Figure IV.37: Block diagram of a feedback oscillator

2. 2 . 1

 21 PA Design Procedure a) Cree's CGHV1J025D GaN Descrete Transistor Die 134 Cree's CGHV1J025D GaN discrete transistor die is chosen as the active device to design the PA as well as the fixed frequency and the tunable frequency oscillators. The manufacturer provides a nonlinear model for the device, so DC and small signal analysis can be easy done using one of commercially available electromagnetic simulation tool. Again for this design, ADS Agilent was used as EM and schematic simulator. From the datasheet [139] provided by Cree, the total gate length of the CGHV1J025D is 5 mm and specified power density of the device is around 5 W/mm. Therefore, it can provide approximately 25 W output power with 10 dB power gain in 9-11 GHz frequency band. However, these values are measured using on-wafer measurements so they do not include any circuit loss. A photo of the transistor is shown in Figure IV.38.

Figure IV. 38 : 3 

 383 Figure IV.38: Photo of the CGHV1J025D GaN Transistor Die

Figure IV. 39 :

 39 Figure IV.39: 3D Bond wire model.

Figure IV. 40

 40 Figure IV.40 shows the simulated I-V curve of the device. The FET transistor is biased at Vgs=-2.5V for class AB operation with a bias current which is approximately 10.2 % of maximum Ids. The high Vds allowed by Cree is about 40V

) 𝐾 = 1 +

 1 |𝛥| 2 -|𝑆(11)| 2 -|𝑆(22)| 2 2|𝑆(12)𝑆(21)| Equation (24) 𝛥 = 𝑆(11)𝑆(22) -𝑆(12)𝑆(21)However, there is an inverse proportionality between k-factor and maximum available gain (MAG) so that k-factor must be barely higher than 1 in the frequency band of interest but as large as possible out of band to have high gain and to prevent out of band oscillations. Stability analysis must be done from very low 137 frequencies to very high frequencies to observe potentially unstable regions. The simulated K-factor and Δ results are shown in Figure IV.41. So a series resistor of 1.45 Ω [139] was required for this design to get device stabilized. Therefore, this stability resistor is thought as a part of the input matching circuit.

Figure IV. 40 :

 40 Figure IV.40: Simulated I-V Curves of the transistor

Figure IV. 42 :

 42 Figure IV.42: A typical distributed bias network with radial stub

Figure IV. 43 :

 43 Figure IV.43: Fabricated PA under Test

  given in Figure IV.[START_REF] Masuda | C-Ku band GaN MMIC T/R frontend module using multilayer ceramics technology[END_REF]. It is showing that gain of the amplifier is around 6 dB which satisfy the first oscillation condition (> 1 dB) at 10.1 GHz. Moreover, input and output return loss value of the PA is better than -12 dB this clearly shows that a good matching is achieved. In addition, S12 is lower than -12 dB.

Figure IV. 44 :

 44 Figure IV.44: Simulated Small Signal S-Parameters

Figure IV. 45

 45 Figure IV.45 represents the small signal measurement results of the fabricated PA. It is observed that the PA does not have any gain, a value of -5.3 dB was observed for S21 at 9.6 GHz, even good match is achieved at the input (S11 = -16.4 dB)as well as the output (S22 = -11.6dB) as can be seen. It seems that this PA does not works

Figure IV. 45 :

 45 Figure IV.45: Measured Small Signal S-Parameters

  kind of oscillator topology, it is preferred to proceed firstly by get the matching done (S11 << 1), then secondly the gain and the phase can be optimized to satisfy the oscillation conditions.

3. 3

 3 Figure IV.47 illustrates the final realized FFO prototype. As can be seen in Figure IV.48, the open-loop analysis has been done and simulated results aredisplayed. An open-loop gain around 6 dB with a good matching at the input is achieved at 10.12 GHz. While the signal phase is around 0° at this frequency. These values satisfy the required oscillation conditions.

Figure IV. 52

 52 Figure IV.52 shows the results of phase noise at 10 KHz, 100 KHz and 1 MHz offset where values of -103 dBc/Hz, -110 dBc/Hz, and -143 dBc/Hz were simulated, respectively.

Figure IV. 47 :

 47 Figure IV.47: Fixed Frequency Oscillator Fabricated Prototype

Figure IV. 53 :

 53 Figure IV.53: Harmonics Power Measurement of the FFO

Figure IV. 54 :

 54 Figure IV.54: Tunable Frequency Oscillator fabricated Prototype

  Simulated phase noise versus bias voltage across varactors at three offset frequencies for the TFO is shown in Figure IV.57. For all bias points, phase noise values of -92 dBc/Hz, -112 dBc/Hz and -132 dBc/Hz are met at 10 KHz, 100 KHz and 1 MHz, respectively. Also, due to the variation of REQ, a slight variation of phase noise is noticed from -5 V to -3.7 V.

Figure IV. 55 :

 55 Figure IV.55: Oscillation frequency versus gate voltage

  frequency of the antenna can be tuned electrically by implementing the GaN varactor between radiation element and ground as it is shown in Figure IV.58. The varactor junction capacitance varies as a function of bias voltage as discussed in chapter III. These different capacitive loadings correspond to various electrical lengths of the antenna, and thus different resonant frequencies.

Figure IV. 58 :

 58 Figure IV.58: Printed antenna prototype with GaN varactor load.

Figure IV. 59 .

 59 Figure IV.59. At this frequency, the simulated gain was about 4.6 dB as displayed in Figure IV.60. The radiation pattern of the antenna is simulated in the E-plane and H-plane for co-polarization (copol) at 11 GHz and results are plotted in Figure IV.61.

  of the limited size of the ground conductor and also some perturbation caused by the SMA connector used for the antenna excitation. The presented results of the antenna will be later experimentally verified and rediscussed.

Figure IV. 59 :

 59 Figure IV.59: Simulated return loss of the antenna alone

Figure IV. 62 :

 62 Figure IV.62: Simulated return loss of the loaded antenna for different reverse-bias voltages

Figure IV. 63 ,

 63 Figure IV.63, the S11 varies from -7.13 dB to -6.9 dB in the band from 7.53 GHz to 7.83 GHz which corresponds to the same simulated tunable bandwidth. Therefore, as can be seen from both Figure IV.62 and Figure IV.63, the simulation results match quite well with the actual prototype measurement even though the minimum return loss has been decreased slightly from the simulation results that can be probably

Figure IV. 63 :

 63 Figure IV.63: Measured return loss of the loaded antenna for different reverse-bias voltages

Figure IV. 69 :

 69 Figure IV.69 : Simulated return loss of the antenna alone

Figure IV. 72 :

 72 Figure IV.72: Simulated reflection coefficient (S11) of the loaded antenna for different varactor reverse-bias voltages

Figure IV. 73 :

 73 Figure IV.73: Measured return loss (S11) of the loaded antenna at no DC bias applied

Figure IV. 76 .

 76 The whole concept size is 14x14x29cm 3 . As shown in Figure IV.76, the four sides rectangular 4x1 patch array antennas with series feed are placed perpendicular over the feeding system as illustrated in Figure IV.77. Here 4 elements are used and each elements has the same

  Figure IV.79 while in Figure IV.80 a zoomed picture of the wires bonding is displayed to show the connected pads of the SPDT switch. Moreover, to facilitate the wire bonding connection between the feeding system design and the SPDT circuit, a gold layer over the copper metal was added as seen in Figure IV.78.

Figure IV. 76 :

 76 Figure IV.76: 3D model of designed antenna system.

Figure IV. 77 :

 77 Figure IV.77: Fabricated prototype of the antenna

Figure IV. 81 :

 81 Figure IV.81: Front view of the fabricated antenna

  this frequency, a return loss value better than -30 dB is obtained in simulation for one 4x1 patch arrays antenna. Because the return loss of the four 4x1 patch arrays antenna are similar, here we report only return loss for one which corresponding to one case.The radiation pattern in H-plane corresponding to the four cases possibilities is shown in Figure IV.84. A shift of 90° between the four cases in the radiation pattern is clearly achieved. The 3-dB beam width where the four beam configuration cases intersect for each case is found to be approximately 85°.

dB at the operating frequency of 2 .

 2 43 as depicted in Figure IV.83. It is important to keep in mind that in the EM simulation only short and open were used instead of the model of the SPDT, this means that almost -3dB of losses of one SPDT switch structure has to be considered. So this will results on the maximum antenna gain value that would decrease with almost -6 dB because the RF signal go through two SPDT switch structures as explained in Table

Figure IV. 82 : 4 Figure IV. 85 : 3

 824853 Figure IV.82: Simulated return loss of the antenna

Figure IV. 86 : 4 Figure IV. 89 :

 86489 Figure IV.86: Measured return loss of the antenna

Figure IV. 90 :

 90 Figure IV.90: Measured S-parameters for case_1 at Pin of 0 dBm

Figure IV. 94 :

 94 Figure IV.94: Measured insertion loss versus input RF power for case_1 at 2.43 GHz

  7 V, the oscillation frequency values change significantly and a tuning bandwidth of 0.8 GHz is obtained, corresponding to a tuning range about 10%. The output power is about 20.5 dBm and almost stable over all bias levels with a slight variation over range[START_REF] Ross | Gallium nitride hybrid microwave circuits for low-noise applications[END_REF] when VG varies while a good phase noise value of -112 dBc/Hz is met at 100 KHz. Also three reconfigurable antennas were designed and presented. The first design is a simple element square loop coplanar antenna printed onto glass substrate. The frequency tunability has been achieved by loading the previous developed GaN varactor diode. The antenna was electronically tuned by varying the varactor bias voltage applied at the gate, VG. As reposted, the minimum and the maximum values of the measured input return loss are, respectively, -6.9 dB and -7.1 dB over the bandwidth from 7.53GHz to 7.85GHz corresponding to a tunability up to 4% narrow bandwidth inside. This frequency tunability is achieved by using a developed GaN varactor. Another frequency reconfigurability antenna was described and presented. The used structure is based on aperture coupled microstrip patch antenna feed technique. The antenna structure with overall dimensions L of 50 mm and W of 50 mm consists of two dielectric layers:

Table II .

 II 1: Comparison between properties of the most commonly used semiconductor technologies [1] .................................................................................................................................................................... 10 Table II.2: Parameters used in the selection of a grow substrate for GaN devices: lattice constant (α), thermal conductivity at 300K (K300), thermal expansion coefficient (αL), wafer diameter and approximate cost [30] ................................................................................................................................ 16 Table III.1: Presented GaN500v1 and GaN150v0 Devices Information ................................................ 47

Table III .

 III

  Last but certainly not least, my future wife Imane EL ABBOUDY, whose love, understanding, and patience gave me the confidence and strength I needed to Les travaux de thèse s'inscrivent dans une stratégie de développement des dispositifs GaN nécessaires à la conception de circuits RF reconfigurables. Les travaux sont divisés en deux parties principales. La première partie de ce travail de thèse est consacrée au développement et à la caractérisation des circuits actifs réalisés à base de la technologie GaN. Pour cela une investigation approfondie autour de diodes Varicap et de commutateurs a été menée et présentée. Les composants actifs ont été réalisés en utilisant le processus de fabrication GaN HEMTs offert par le Conseil National de Recherches du Canada (CNRC). Pour ce faire, trois processus Single Pole Double Throw) ont été fabriqués, en utilisant le processus de fabrication GaN500v2. En analysant les mesures en petit signal des commutateurs simples, on a pu observer que l'isolement est bon aux basses fréquences, mais se dégrade rapidement avec l'augmentation de celle-ci. En outre, on a pu conclure que plus le dispositif est grand, plus les pertes d'insertion s'améliorent, alors que l'isolation se dégrade. Donc un compromis entre perte d'insertion et isolation a été fait pour choisir les transistors individuels qui ont par la suite été utilisés pour la conception des structures de commutateurs plus complexes (SPST et SPDT). De plus, en se basant sur ces mesures petit signal, un modèle circuit a été introduit.

	Résumé en français Integrated Circuits (MMICs), based on semiconductor technology, are one of the keys
	to meeting the requirements of this technology mixing. Additionally, most MMICs
	Cette thèse a pour objectif de démontrer la faisabilité de l'utilisation de la currently in production, such as gallium arsenide (GaAs), silicon carbide (SiC), silicon
	technologie Nitrure de Gallium (GaN) dans les systèmes RF / micro-ondes (Si), indium phosphide (InP), etc, operate in the microwave and millimeter-wave
	reconfigurables. La technologie GaN doit permettre de supporter des puissances (mm-w) range with modest breakdown voltages and limited power handling
	élevées avec un rendement important. Le GaN qui présente une large bande interdite capabilities. But there are increasing applications in the RF/microwave spectrum
	possède des propriétés physiques lui permettant d'être largement utilisé pour la where higher power/frequency is required.
	conception de circuits actifs à forte puissance. En effet, ses qualités intrinsèques,
	achieve success.
		Abdelaziz HAMDOUN
		December, 2016
	O Allah, benefit me by that which You have taught me, and teach	َا م ِي ن ْ م ِّ َل ع وَ ِي ن َ ْت م َّ ل عَ	َا م ِ ب ِي ن عْ َ ف ْ ان َّ ُم ه َّ الل
	me that which will benefit me, and increase me in knowledge.	ًا م ْ ل عِ ِي ن دْ ِ ز وَ ِي ُن ع َ ف ْ َن ي
			2

champs de claquage élevé, forte conductivité thermique, forte densité d'électrons, et bonne vitesse de saturation, lui confèrent de très bonnes performances aux hautes fréquences que ce soit pour les applications de puissance ou faible bruit. La technologie GaN est donc un candidat très prometteur pour la réalisation de fonctions hyperfréquence haute puissance/haute fréquence, telles que les amplificateurs de puissance, les mélangeurs, les oscillateurs contrôlés en tension (VCO) ou encore les amplificateurs faible bruit (LNA). Cependant, malgré les caractéristiques intéressantes des dispositifs GaN, il reste encore des défis de fabrication à relever avant que cette technologie ne soit largement adoptée à l'avenir. Ces défis sont liés à deux aspects, le premier technologique et le second commercial. Technologiquement, les dispositifs HEMTs basés sur le GaN sont affectés par deux phénomènes importants, à savoir l'effet de piégeage et l'effet thermique (ou auto-échauffement des dispositifs) qui peuvent directement causer la dispersion du courant. Commercialement, un inconvénient de la technologie GaN concerne le coût de fabrication des composants. La réduction du coût de production est un facteur important pour pouvoir largement commercialiser les dispositifs GaN. De plus, le peu de disponibilité de matière GaN conduit à épitaxier le GaN sur des substrats tels que Silicium (Si), Carbure de Silicium (SiC) et le Saphir, ce qui peut dégrader les caractéristiques des composants GaN. Ces aspects sont abordés dans le chapitre II de ce manuscrit. GaN HEMT : le GaN150v0 avec une longueur de grille de 0,15 um, le GaN500v1 avec longueur de grille de 0,5 um et le GaN500v2, ont été investigués pour fabriquer des diodes varicap avec des tailles différentes. Ces diodes ont ensuite été caractérisées en DC, en petits et grands signaux RF jusqu'à 20 GHz. Ici, les diodes varicap sont réalisées simplement en reliant le drain à la source, conduisant à réaliser une jonction Schottky qui peut être considérée comme « Planar Schottky Varactor Diode (PSVD) ». La tension VG de polarisation est alors appliquée entre le drain/source et la grille. En outre, ce type de diode ne nécessite aucun changement particulier des couches épitaxiales du procédé de fabrication HEMTs. Ensuite, les diodes varicap ont été modélisées en petit et grand signal par des équations analytiques. Ces expressions ont été introduites pour décrire la variation de la capacité équivalente (CEq) et la résistance série (REq) en fonction la tension DC appliquée sur le composant. Ces équations peuvent être utilisées comme un modèle général pour représenter le comportement non-linéaire des diodes varicap basées sur la technologie GaN. Pour le modèle petit signal, les équations analytiques développées ne sont dépendantes que de la tension de polarisation appliquée sur le dispositif et de la géométrie de la diode, tandis que pour le modèle grand signal, l'influence de la puissance RF a aussi été prise en compte pour décrire le comportement de REq et CEq. Pour étendre l'étude autour des composants GaN actifs utilisés dans les systèmes RF reconfigurables, plusieurs commutateurs avec différentes tailles configurés en grille commune ainsi que des structures de commutateurs tels que le SPST (Single Pole Single Throw) et le SPDT (La deuxième partie de ces recherches de thèse aborde les aspects de l'intégration de ces dispositif actifs GaN (diodes varactor et commutateurs) et de la conception de circuits reconfigurables proposés, tels qu'un déphaseur reconfigurable, un coupleur (3dB, 90°) reconfigurable, un oscillateur accordable en fréquence, ainsi qu'un réseau d'antennes à commutation de faisceau et une antenne unitaire accordable en fréquence. Initialement, un nouveau RUN technologique devait nous permettre de concevoir de nouveaux designs complètement intégrés en technologie GaN notamment le déphaseur et le coupleur (3dB, 90°) reconfigurable en fréquence mais malheureusement ce RUN technologique n'ayant pas été assuré, il a donc été décidé d'utiliser les diodes varactors et les commutateurs préalablement fabriqués et testés sous pointe. Ceci a engendré des soucis puisque ces puces étaient de très petites dimensions et donc difficilement découpables puis intégrables dans les circuits imprimés tels que l'oscillateur ou les antennes. Toutefois, dans un souci de mener jusqu'au bout cette étude et de réaliser les circuits et antennes reconfigurables, de gros efforts ont été consentis afin de récupérer des diodes et commutateurs sur les wafers de base. Il a donc pu être possible de réaliser l'oscillateur et les deux types d'antennes reconfigurables. Si l'oscillateur contrôlé en tension et reconfigurable grâce à une diode Varicap préalablement fabriquée et caractérisée n'a pu aboutir à des résultats expérimentaux concluants, les prototypes d'antenne élémentaire reconfigurable en fréquence ou en diagramme à l'aide d'un réseau d'antennes commutable ont permis de démontrer la faisabilité de tels concepts en utilisant des composants actifs en technologie GaN.

Introduction I. Research Motivations

Nowadays, people are living in a society in which processing, flow and exchange of information are vital for their existence. Connectivity and mobility are two major issues in such society, which are related to the flow and exchange of information. On one hand, portable devices with internet provide connectivity which allows communication as well as access to information. On the other hand, wireless technologies bring mobility. People can move and still be able to communicate while having access to various kinds of information. This is the reason why radio frequency (RF)/microwave systems have evolved to integrated formats. This approach is called system-on-package (SoP), where many technologies are mixed in one package allowing increased simultaneous functionalities. Furthermore, the recent trend in RF/microwave technology is toward circuits that are highly miniaturized integrated and reliable, with low power consumption, and reduced cost. Monolithic Microwave

  of the Literature of Gallium

	wireless base stations, communication and military radar systems, where its focus is
	shifting to further advantages such as high efficiency as well as the reduction of
	worldwide energy consumption [4].
	Historically, development of semiconductor technologies based on
	Monolithic Microwave Integrated Circuit (MMIC) has always advanced to more and
	more performance improvements in terms of more high power handling capabilities,
	more gain, high efficiency, high linearity, decreasing noise figure [2]-
	Nitride Technology
	I. Gallium Nitride Technology
	1. Overview
	Since gallium nitride (GaN) is used as a semiconductor technology to
	design the circuits for this thesis, it is necessary to review the characteristics of this
	technology and make a brief comparison with commonly used semiconductor
	technologies. GaN technology has been and still getting more and more attention for
	RF/microwave systems that require high power handling capabilities and low noise
	figure. This sets the field of GaN electronics in motion, and today the technology is
	improving the performance of several applications including, mobile communication,
	8

Table II

 II

	.1: Comparison between properties of the most commonly used
		semiconductor technologies [1]
	Semiconductor (Typical Materials)	Si	GaAs	InP	SiC	GaN
	Characteristic	Unit
	Bandgap	eV	1.1	1.42	1.35	3.25	3.49
	Electronic Mobility at 300 ºK	cm 2 /V.s	1500	8500	5400	700	1000-2000
	Saturated Electronic Velocity	x10 7 cm/s	1	1.3	1	2	2.5
	Critical Breakdown Field	MV/cm	0.3	0.4	0.5	3	3.3
	Thermal Conductivity	w/cm.k	1.5	0.5	0.7	4.5	>1.5
	Charge channel Density	x10 13 /cm 2	0.3	0.2			1
	Resistivity	Ω.cm	1.5	1x10 7		1x10 5
	Relative Dielectric Constant	Ԑr	11.5	12.8	12.5	10	9
	Loss tangent (at 10 GHz)	Tg	0.015	0.0004		0.003

Table II

 II

	.2: Parameters used in the selection of a substrate for GaN devices: lattice development and months later its release was announced. Recently, the GaN500
	constant (α), thermal conductivity at 300K (K300), thermal expansion coefficient process has been stopped to be commercialized in meanwhile the GaN150 process has
	(αL), wafer diameter and approximate cost [30] just been released. Some varactors have been fabricated using the initial version
	Material (non-commercial) of this process when it was still under development. The access α (ºA) k300 (W/K.m) αL 5.59(ppm/K) Diameter (inches) Cost
	GaN SiC availability to the NRC GaN process is possible only through a company called CMC 3.189 130 5.59 ≤ 4 High 3.08 490 4.2 ≤ 4 Medium
	Sapphire Microsystems which is an intermediate between NRC and costumers. 2.747 41 7.5 ≤ 6	Low
	Si	5.4301	148	3.59	≤ 12	Low
	AiN Diamond	3.112 3.75 The proposed foundry [31], combines high power density, excellent thermal 285 4.2 ≤ 2 High 2000 1.18 ≤ 4 High
	In the process used for designing the circuits for this thesis, a SiC substrate conductivity, and high gain performance. The HEMT devices are fabricated on an
	is used. SiC substrate provides a good compromise between cost, excellent thermal aluminum gallium nitride/gallium nitride (AlGaN/GaN) hetero-structure grown on
	properties which makes this device optimal to operate at high power. semi-insulating 4H-SiC substrates. Semi-insulating SiC is one of the most attractive
	substrate materials for electronic applications, because of the favorable combination
	II. NRC GaN500 MMIC HEMT Process Features of lattice mismatch, isolation, and thermal conductivity [[18]. In addition, AlGaN/GaN
	HEMT structures are formed by using ammonia molecular beam epitaxy (MBE) in a
	and Limitations SVTA nitride MBE system and are consisted of a 3 nm thick aluminum nitride (AlN)
	nucleation layer, a 1.1 μm thick carbon-doped GaN insulating buffer, followed by a
	The great interest raised by AlGaN/GaN HEMT devices in the 0.2 μm thick undoped GaN layer, and then 1.5 nm AlN spacer plus 20 nm AlGaN
	international semiconductor scientific community in general, for high barrier
	power/frequency applications, has resulted in that an important number of industries
	and academics labs research (i.e United Monolithic Semiconductors (UMS),
	Fraunhofer IAF, Ferdinand Braun Institute (FBH), TriQuint, CREE, National
	Research Council Canada (NRC), etc) in the world are nowadays fully involved to
	demonstrate new microwave HEMTs devices in this new GaN technology. The
	presented varactor and switch circuits in this thesis were manufactured by the
	Canadian Photonics Fabrication Center (CPFC) at the National Research Council
	Canada (NRC).
		NRC offers a foundry services for AlGaN/GaN HEMT-based MMIC process
	for both university and industry customers. At the time of this thesis research, the
	technology features include a metal gate of 0.5 μm in the GaN500 process which was
	commercialized. Also, a metal gate of 0.15 μm in the GaN150 process was still under

  Table III.1: Table III.1: Presented GaN500v1 and GaN150v0 Devices Information

	Device	Number of	Fingers	Fingers	Process
	Terminology	Fingers	Width	Length	Terminology
	W40L0.5	2	40 μm	0.50 μm	GaN500v1
	W100L0.5	2	100 μm	0.50 μm	GaN500v1
	W50L0.15	2	50 μm	0.15 μm	GaN150
	W100L0.15	2	100 μm	0.15 μm	GaN150

  .3. Table III.3: Comparison of capacitance performances between GaN500 and GaN150

		GaN500v1	GaN150v0
	Capacitance	2.7fF/um	1.5fF/um
	CMAX/CMIN	2.4	3.7
	Tuning Voltage	-4.5 V to -3.5 V	-4.5 V to -3.5 V

Table III

 III

	.4 : Presented GaN500v2 Devices Information
	Device	Number of	Fingers	Fingers	Process
	Terminology	Fingers	Width	Length	Terminology
	W50L0.5	2	50 μm	0.50 μm	GaN500v2
	W100L0.5	2	100 μm	0.50 μm	GaN500v2
	W200L0.5	2	200 μm	0.50 μm	GaN500v2

Table III .

 III 5: Parameters values of the equivalent circuit

	Device	VG (V)	Lpara (pH)	REq (Ω)	CEq (fF)
	W50L0.5	-3	5	7.9	466
		-4	5	11	375
		-6	5	9.5	279
	W100L0.5	-3	16	5.3	870
		-4	16	7.35	637
		-6	16	6.5	501
	W200L0.5	-3	100	4.7	1810
		-4	100	5.85	1403
		-6	100	5.2	1016

  Table III.6. Table III.6: Model Circuit Parameters Values

		20
									W50L0.5
		18							W100L0.5
		16							W200L0.5
		14
		12
		10
		8 REq [Ohms]
		6
		4
		2
		-6 0	-5.5	-5	-4.5	-4	-3.5	-3	-2.5	-2
						VG [V]
	Device	VG (V)	GaN500v1 Process			GaN500v2 Process
		Lpara (pH)		REq (Ω)	CEq (fF)	Lpara (pH)	REq (Ω)	CEq (fF)
	W50L0.5	-3	25		17	205		5	7.7
		-4	25		50	190.7		5	9.95
		-6	25		15.5	82.7		5	9.2
	W40L0.5	-3	30		6.9	533		16	5.2
		-4	30		13.3	439		16	6.9
		-6	30		6.16	220		16	6.2
	W200L0.5	-3							100	4.7	1810
		-4							100	5.6	1403
		-6							100	5.2	1016

Table III .

 III

	IV. Summary
	In this chapter, GaN HEMT-based varactor devices have been fully
	characterized and modeled for the first time, showing limited usefulness for high
	power applications due to their rapidly changing C-V characteristic. These GaN-
	based varactors were designed in a standard GaN HEMT process, and were
	fabricated with two different gate lengths, 0.5 µm (GaN500v1 and v2) and 0.15 µm
	(GaN150v0). The varactor diode is realized by connecting together the drain and
	[dB]
	Freq [GHz]
	Performances Secondly, a general large signal model for GaN varactor diodes Values
	manufactured with gate lengths of 0.5 µm (GaN500v2) has been presented. This
	Modeled Insertion Loss Measured Insertion Loss model is based on an analytic expression which describes the Ceq-VG curve and has -2.723 dB -3.164 dB
	Modeled Isolation Measured Isolation shown to maintain a high level of accuracy, well into the large power range up to 18 -28.158 dB -25.937 dB
	Modeled Return Loss dBm. In addition, the proposed circuit model in small signal analysis is always valid -11.847 dB
	Measured Return Loss	-14.697 dB
	for large signal by considering the new analytical equation of CEq versus the input
	power (Pin.) With reverse bias and RF power level included in the model, the junction
	capacitance ratio (CMAX/CMIN) of the realized varactors was shown to be decreased

8: Performances of the SPDT at the Marker m4 source terminations thus resulting in a single Schottky junction which can be considered as Planar Schottky Varactor Diodes (PSVDs). First of all, a small signal analysis has been conducted and by analyzing the extracted CEq-VG curves, mathematical expressions have been proposed to capture the nonlinear behavior that can be used to represent a variety of different GaN based varactor sizes. These equations maintain the simplicity of equivalent circuit modeling while offering good agreement with measurements. In addition, an equivalent simple circuit model was proposed and simulated, yielding good agreement between the simulated and the measured results. The obtained Cmax/Cmin ratios and figures of merit compare favorably with recently reported GaN varactors. from 1.73 to 1.25. This ratio can be much further decreased, if the RF power level is increased further. From the presented results and given GaN's high breakdown voltage, these varactors would be very useful in various applications in receive mode, where power limiters could be completely excluded as protection circuitry from jammer signals.

  Table IV.1 and Table IV.2. Table IV.1: Optimized Values of Inductors L1 and L2

		Inductor L1	Inductor L2
	Value (nH)	6.7	8.45
	Ns (turns)	10	10
	L1 (µm)	120	155
	L2 (µm)	125	125
	L3 (µm)	110	135
	Ln (µm)	20	20
	W (µm)	15	15
	S (µm)	5	5
	Wb (µm)	10	10
	Ab	90	90
	Be (µm)	30	30
	Table IV.2: Optimized Values of Capacitors C1 and C2
		Capacitor C1	Capacitor C2
	Value (fF)	369	136
	W (µm)	43.86	26.63
	L (µm)	43.86	26.63
	Wc (µm)	15	10

Table IV . 3 :

 IV3 Dimensions of Stubs

			Input Matching Network	Output Matching Network
		Length	3.8 mm	2.24 mm
	Stub 1	Width	2 mm	2 mm
		Length	3.66 mm	3.75 mm
	Stub 2	Width	2 mm	2 mm

Table IV .

 IV 5: Switching configuration to achieve radiation pattern reconfigurability. As highlighted in one 4x1 patch arrays antenna return loss which is shown in Figure IV.82, an impedance bandwidth of approximately 40 MHz (2.45 GHz to 2.41 GHz) is achieved, with a resonance frequency sensing at approximately 2.43 GHz. At

				38 mm
					32 mm
				37.7 mm
		SPDT_1		SPDT_2		SPDT_3		Φ
								(deg)
		RF_Output_1 RF_Output_2 RF_Output_1 RF_Output_2 RF_Output_1 RF_Output_2
	Case_1	ON	OFF	OFF	OFF	OFF	ON	0°
	Case _2	OFF	ON	OFF	OFF	OFF	ON	90°
	Case _3	OFF	OFF	ON	OFF	ON	OFF	180°
	Case _4	OFF	OFF	OFF	ON	ON	OFF	270°

Acknowledgements ix

Acknowledgements

Praise is to God. الحمد

 -------------------------------------------------------------------------------------------------------------------Some contributions to the field of microwave integrated and reconfigurable circuit designs have been made from this thesis research. As mentioned in Chapter III, one journal paper [START_REF] Hamdoun | Characterisation and analytical modelling of GaN HEMT-based varactor diodes[END_REF] has been published and one other has been submitted [START_REF] Hamdoun | Large Signal Analytic Modeling of HEMT-based Varactor Diodes[END_REF]. A conference paper has also been presented at the European Microwave Conference in Paris in 2015 [START_REF] Hamdoun | Characterization of GaNbased HEMTs as varactor diode devices[END_REF]. Furthermore, a part of the work of the beam steering patch antenna arrays using GaN SPDT switch that was presented in Chapter IV section 4 will be presented at the European Conference on Antennas and Propagation in Paris in 2017 [START_REF] Hamdoun | Switched Beam Patch Array Antenna Using SPDT GaN HEMT Switches[END_REF] and also based on the antenna work one journal paper is expecting to be submitted for publication. Despite all the obstacles that we faced during the manufacturing of the GaN design (such as, runs cancelled, chips dicing …), we succeed to overcome some of them and develop GaN varactor and switch devices which were later integrated with PCB designs to achieve reconfigurability.

There are several areas related to this thesis which are ripe for study. First, if the required equipment became available, large-signal GaN device modeling, particularly for varactor switch devices is an area which should be developed.

Improved performance in large signal operation and going higher in frequency of these devices can be expected when improved commercial NRC GaN150 become fully available.

Initially, a NRC GaN150 run was planned and highly miniaturized phase shifter and reconfigurable hybrid coupler designs were planned to be fabricated and tested. Unfortunately these designs have not been manufactured because the run has been cancelled by NRC. So future work could be include both phase shifter and reconfigurable hybrid coupler using the advanced developed version of NRC GaN150 Summary 192 process. Also it was expected to develop a frequency tunable antenna feeder such as annular slot antenna using varactor diode for reconfigurability. This feeder can be used to tune the frequency of an antenna that is fabricated by using a low cost substrate.