Keywords: radar, amplificateur à faible bruit, mélangeurs, millimétrique, radiofréquence, 4. Performances of implemented RF-Blixer.......

Conception de récepteurs aux fréquences millimétriques pour des applications radars automobiles embarqués

Depuis plusieurs années, l'industrie automobile multiplient les systèmes d'aide à la conduite dans l'objectif d'améliorer l'autonomie des véhicules. Pour cela, chaque véhicule intègre un panel de capteurs de plus en plus étoffé. Le radar 80GHz embarqué présente alors des avantages multiples : immunité aux conditions extérieures, mesure simultanée de la distance et la vitesse… Ainsi, l'industrie du semiconducteur se trouve face à deux défis : réduire le coût des puces et améliorer leurs performances. Cela passe par l'utilisation de technologie CMOS avancées et, par l'amélioration des principaux facteurs de mérite du radar comme la puissance de sortie de son transmetteur, le bruit de phase de son oscillateur local et le facteur de bruit de son récepteur. L'objectif de cette thèse est de proposer des architectures et topologies susceptibles d'améliorer significativement les performances d'une récepteur radar. Dans ce cadre, le récepteur radar est soumis à deux contraintes qui peuvent être contradictoires. Placé derrière le pare-chocs, le module radar est affecté par la réflexion du signal transmis. Face au risque d'apparition de non linéarités, donc de fausses cibles, de fortes contraintes sont exigées en termes de point de compression. De l'autre côté, pour réduire le facteur de bruit globale du récepteur, il y a besoin de maximiser le gain et minimiser le facteur de bruit de sa partie millimétrique. Pour répondre à cela, cette thèse propose plusieurs pistes. Le premier axe est centré sur les techniques de conception de circuits millimétriques. D'un côté, il s'agit de réduire la consommation des circuits au travers d'étages complémentaires et l'opération en moyenne inversion. De l'autre côté, il s'agit d'implémenter des techniques d'annulation de bruit. Ces techniques permettent simultanément de travailler sur de larges bandes passantes et d'améliorer les performances en bruit. Cela a amené au développement d'une topologie dite « complémentaire à couplage transversale capacitif ». Le second axe de ce travail se concentre sur les techniques de mélange. Les mélanges actifs présentent un bon compromis entre gain, linéarité et facteur de bruit mais également un bruit en 1/f élevé, malgré l'implémentation de techniques pour le réduire. Ces topologies peuvent adresser des applications de RADAR courte portée mais pas longue portée. Les mélangeurs passifs présentent de meilleures performances en bruit mais sont plus contraignants en termes de linéarités. Pour cela, cette thèse propose d'implémenter un mélangeur en courant à 80GHz pour améliorer simultanément les performances en bruit et linéarité, capable d'adresser des applications RADAR longue portée.

Abstract

Design of millimeter-Wave Receivers for Embedded Automotive Radars

a. A utom ated vehicles trends

The next decades will present many challenges for automotive industries. First, safety is a major concern as 1 million people died each year in road crash due to human error [START_REF]Autonomous cars: a big opportunity for European industry[END_REF].

Besides, traffic congestion becomes an issue for many cities and will cost more The Society of Automotive Engineers proposes a classification of automation levels for vehicles from 0 to 5 [START_REF]SAE International Releases Updated Visual Chart for Its "Levels of Driving Automation" Standard for Self-Driving Vehicles[END_REF]. A quick description is proposed:

• Level 0 -N o autom ation: Human driver performs all the dynamic driving tasks, safety features only provide warning or momentary assistance such as Automatic Emergency Breaking, Blind Spot or Lane Departure warning,

• Level 1 -D river assistance: Human still drive and monitor the environment but features proposes assistance for basic driving nodes like cruise control or lane centering,

• Level 2 -P artial autom ation: The car can drive in some driving modes, but human driver must constantly monitor the environment and be ready to take control,

• Level 3 -C onditional autom ation: Automated system can drive under limited conditions (ex. Traffic jam chauffer) but human must drive when feature requests,

• Level 4 -H igh autom ation: The automated driving system performs all driving task in many driving circumstances,

• Level 5 -Full autom ation: Automation system can drive anywhere in all condition using only its own systems.

b. Sensors in autom ated vehicles

To reach to SAE level 5, autonomous cars manufacturers require a continuous and ubiquitous monitoring of the car environment. [START_REF]EN 301 091[END_REF], [START_REF]EN 302 264[END_REF] and [START_REF]ETSI EN 302 858[END_REF] For instance, a 0.1° accuracy is needed for spatial resolution. Due to these constraints, radar transmitter is supposed to generate a signal with a high effective isotropic radiated power (EIRP) whereas the receiver is expected to achieve a high sensitivity. This metric denotes the ratio between signal power and noise power and it's a key parameter to define RF performances of a radar system.

Standards

R adar applications in autom ated vehicles

II. FM C W M odulation B asics

The radar industry develops many modulations: pulsed radar, continuous wave, FSK modulation… Frequency Modulated Continuous Wave (FMCW) modulation is still dominant in ADAS. This technique is based on "chirps", a continuous signal whose frequency varies linearly with time, which gives to FMCW some advantages:

-Easy generation of the transmitted signal, -Simultaneous measurement of target distance and velocity with a good resolution, -Simple baseband for signal processing at few MHz, -Low peak power constraint compared to pulsed radar. Considering the characteristics of the various radars exposed in the Table 3, the Figure 15 proposes the maximum archivable range regarding the fbeat frequency. In the range of ten microseconds, and for distant target, fbeat increase up to 40MHz. These trends make more challenging the analog signal processing in baseband sections and lead to faster RF bandwidth. and this section focus on its main specifications: gain, noise and linearity. In the following sections, the terms downconverter and RX-RFFE will be used as equivalent.

D istance m easurem ent

V elocity m easurem ent

C onclusion

N oise in autom otive radar [17]

As shown in Figure 5 The Radar Cross Section σ, expressed in m² or dBsm [dB(m²)], represents the target signature and depends on the target size, its physical properties and its shape. In Table 6, some common targets radar cross sections are proposed: The Figure 21

Linearity in A utom otive R adar [20]

For an ideal Front-End, the output signal is a linear image of the input signal.

Unfortunately, the downconverter introduces some distortions which corrupt the original signal.

These non-linearities are further discussed in this section.

a.

In-band linearity (IC P 1) The presence of the in-band blocker, located at fbump as illustrated in Figure 29, is an important issue for downconverter linearity. First, the LNA should not be placed in saturation mode. Besides, if one or two tones are present at the input, important intermodulation products will be created as illustrated in Figure 30. These spurs can be considered as targets by the MCU. The ghost targets are obviously a major issue in automated vehicles. To handle this major issue, the 1dB compression point specification is revised to -5dBm.

IF 2 -IF 1 2IF 1 -IF 2 2IF 2 -IF 1 IF 1 IF 2 2IF 1 2IF 2 IF 2 + IF 1 3IF 1 IF 2 + 2 IF 1 IF 1 + IF 2 3IF 2 2f RF1 -f RF2 f RF2 2f RF2 -f RF1

IV . C onsiderations on receiver R FFE for autom otive radar

As discussed in previous section, the RX specifications are challenging and as consequences, the receiver is a critical block for automotive radar. This section focusses on the selection of the technology for the next generation radar, which will have an impact on RX-RFFE possible architecture.

R F C M OS opportunities

Automotive radars were originally designed in III-V GaAs technologies. As shown in On the other hand, as expressed in 3, radar modules are about to be massively deployed in modern cars, so the price of radar IC become critical.

Equation 12 -Cascaded IIP3 formula 1 𝐼𝐼𝑃3 𝑅𝑋 = 1 𝐼𝐼𝑃3 1 + 𝐺 1 𝐼𝐼𝑃3 2 + 𝐺 1 𝐺 2 𝐼𝐼𝑃3 3 + ⋯ + 𝐺 1 … 𝐺 𝑛-1 𝐼𝐼𝑃3 𝑛
Frequency down-converters based on active and passive mixers are further investigated in this work.

b. R F signal processing

Figure 35 -IQ downconverter RF f LO (0°/180°) f LO (90°/270°) BB ADC ADC < < < < < < < < < < IF I IF Q Digital recombination IF f LO -f LO f LO -f LO -f IM -f RF f RF f IM f IF -f IF f IF -f IF f IF -f IF f IF -f IF BB 90°
A

C onclusion

The

b. N oise cancellation principle

Matching Main stage (A main )

Auxiliary stage (-A aux )

+ - Σ R S v S v in v o+ v o- v n,in v n,+ v n,-

Signal

Noise However, the technique of noise canceling significantly relaxes the trade-off between input matching, noise and gain performances. Equation 14 -Amplifier load in common-gate path (i=1) and common source path (i=2)

C om m on-gate based noise cancelling

𝑌 𝐿𝑖 = 1/(𝑅 𝐿𝑖 + 1 𝑗𝜔𝐶 𝐿𝑖 ⁄ )
The small signal equivalent circuit of the noise cancelling LNA is shown in Figure 39(b).

The admittances at nodes

𝑌 𝑑𝑠1 = 1 𝑍 𝑑𝑠1 ⁄ = 𝑔 𝑑𝑠1 + 𝑗𝜔𝐶 𝑑𝑠1 ≈ 1 𝑟 𝑑𝑠1 ⁄ M 2 Out- V bCS RF Z L1 Z L2 Out+ M 1 V bCG L 1 (a) Out- RF Z gd1 Z ds2 Out+ g m1 •v A Z gd2 Z gs Z ds1 -g m2 •v A (b) A A B B C C

Auxiliary path

Main path

𝑌 𝑑𝑔1 = 1 𝑍 𝑑𝑔1 ⁄ = 1 𝑅 𝐿1 + 𝑗𝜔 • (𝐶 𝑑𝑔1 + 𝐶 𝐿1 ) ≈ 1 𝑅 𝐿1 ⁄ 𝑌 𝑑𝑠2 = 1 𝑍 𝑑𝑠2 ⁄ = 1 𝑅 𝐿2 + 𝑔 𝑑𝑠2 + 𝑗𝜔(𝐶 𝑑𝑠2 + 𝐶 𝐿2 ) ≈ 𝑅 𝐿2 • 𝑟 𝑑𝑠2 𝑅 𝐿2 + 𝑟 𝑑𝑠2 𝑌 𝑑𝑔2 = 1 𝑍 𝑑𝑔2 ⁄ = 𝑗𝜔𝐶 𝑑𝑔2 ≈ 0 𝑌 𝑔𝑠 = 1 𝑍 𝑔𝑠 ⁄ = 𝑗𝜔(𝐶 𝑔𝑠1 + 𝐶 𝑔𝑠2 ) - 𝑗 𝜔𝐿 1 ≈ 0 a.

Im pedance m atching

Equation 16 -Common gate impedance

𝑍 𝑖𝑛,𝐶𝐺 = 𝑍 𝑑𝑠1 + 𝑍 𝑑𝑔1 1 + 𝑔 𝑚,𝐶𝐺 𝑍 𝑑𝑠1 = 𝑟 𝑑𝑠1 + 𝑅 𝐿1 1 + 𝑔 𝑚,𝐶𝐺 𝑟 𝑑𝑠1 ≈ 1 𝑔 𝑚,𝐶𝐺

The Equation 16 proposes an expression of the common-gate input impedance (Zin,CG).

Considering a high drain source impedance regarding the load and neglecting parasitic capacitances, the input impedance can be approximated to the transconductance inverse. Then a wideband real impedance can be achieved as long as reactive elements are negligible.

Z L1 Z L2 L 1 A B C 𝑣 𝐵 𝑣 𝐶 𝑣 𝐴 𝐴 𝑣,𝐶𝐺 𝐴 𝑣,𝐶𝑆
Equation 18 -Gain over the main path

𝐴 𝑣,𝐶𝐺 = 𝑣 𝐵 𝑣 𝐴 = 1 + 𝑔 𝑚1 𝑍 𝑑𝑠1 1 + 𝑍 𝑑𝑠1 𝑍 𝑑𝑔1 ⁄ = 1 + 𝑔 𝑚1 𝑟 𝑑𝑠1 1 + 𝑟 𝑑𝑠1 𝑅 𝐿1 ⁄ = 𝑔 𝑚,𝐶𝐺 • 𝑅 𝐶𝐺 = 𝑅 𝐶𝐺 𝑅 𝑆
For the common-gate path, the voltage gain (Av,CG) is described by Equation 18.

Considering a drain-source resistor significantly higher than the load resistor, the main path

gain
𝑣 𝑛,𝐶 = -𝐴 𝑣,𝐶𝑆 • 𝑣 𝑛,𝐴 = -𝑔 𝑚,𝐶𝑆 • 𝑅 𝐶𝑆 𝑅 𝑖𝑛,𝐶𝐺 𝑅 𝑖𝑛,𝐶𝐺 + 𝑅 𝑆 • 𝑖 𝑛 • 𝑅 𝑆 𝑣 𝑛,𝐶 = -𝑔 𝑚,𝐶𝐺 • 𝑅 𝐶𝐺 𝑅 𝑖𝑛,𝐶𝐺 𝑅 𝑖𝑛,𝐶𝐺 + 𝑅 𝑆 • 𝑖 𝑛 • 𝑅 𝑆 = 𝑣 𝑛,𝐵

d. D istortion cancellation

B i CG (DC) i CG R S v S R CG B A v CG,A v S R S i CG (v gs ,v ds ) v CG,B (a) (b) (c) C i CG (v gs ,v ds )

R CS nonlinearties, the source signal (vs) creates a nonlinear current in common gate path (iCG) which is converted into voltage at the input A (vCG,A). This voltage is decomposed in a Taylor

series as detailed in Equation 20.

𝑣 𝐶𝐺,𝐵 = 𝑖 𝑐𝑔 • 𝑅 𝐶𝐺 = 𝑣 𝑠 -𝑣 𝐶𝐺,𝐴 𝑅 𝑆 • 𝑅 𝐶𝐺 = ((1 -𝑎 1 ) • 𝑣 𝑆 -𝑣 𝑁𝐿 ) • 𝑅 𝐶𝐺 𝑅 𝑆 𝑣 𝐶𝑆,𝐶 = -𝑣 𝐶𝐺,𝐴 • 𝑅 𝐶𝐺 𝑅 𝑆 = -(𝑎 1 • 𝑣 𝑠 + 𝑣 𝑁𝐿 ) • 𝑅 𝐶𝐺 𝑅 𝑆

N oise cancelling applied to m ixer

Z CS LO + B C A Z 1 Z CS Z 1 Z CG V B V B 𝑣 𝑛,𝐶𝐺 𝐶𝐺 𝑣 𝑛,𝐶𝐺 𝐶𝑆 𝑖 𝑛,𝐶𝐺

b. A ttractive properties of B lixer

C onclusion

The Table 10 shows a comparison between the proposed mixer and some works of the state of the art. The proposed design presents competitive RF performances with an interesting trade-off between the die area and the power consumption. First, the bandwidth is significantly higher than most of the others low power downconverters. Some references, [START_REF] Kargaran | A Sub-1-V, 350-μW, 6.5-dB Integrated NF Low-IF Receiver Front-End for IoT in 28-nm CMOS[END_REF], [START_REF] Ramella | A SAW-Less 2.4-GHz Receiver Front-End With 2.4-mA Battery Current for SoC Coexistence[END_REF] and [START_REF] Lin | A 2.4 GHz ZigBee Receiver Exploiting an RF-to-BB-Current-Reuse Blixer + Hybrid Filter Topology in 65 nm CMOS[END_REF],

include a baseband analog chain with a large gain feature to compensate for the mixer losses.

Besides

III. 77G H z N oise C ancelling M ixer

In

Flicker noise v n ∑ 2 • 𝐼 𝑚𝑎𝑥 𝑆 𝐿𝑂 𝛿(𝑡 -𝑘 𝑇 𝐿𝑂 2 ) ∞ =-∞ ∑ 𝑉 𝑛 (𝑓 -2𝑘𝜔 𝐿𝑂 ) 𝜋 • 𝐴 𝐿𝑂 ∞ =-∞

Sampled

M 2 IF- V bCS LO - Z L Z L IF+ LO + LO + M 1 V bCG 1.2V 1.2V V bCB V bCB L s1 L s2
RF Y mixer RF M 2 IF- V bCS LO - Z L Z L IF+ LO + LO + M 1 V bCG 1.2V 1.2V V bCB V bCB L s1 L s2
𝐺 𝑚𝐶𝐺 = 𝑔 𝑚1 + 𝑌 𝑑𝑠1 1 + 𝑌 𝑑𝑠1 /𝑌 𝑑𝑔1 ≈ 𝑔 𝑚1 + 𝑔 𝑑𝑠1 1 + 𝑔 𝑑𝑠1 𝑌 𝑠𝑤𝑖1 ; 𝐺 𝑚𝐶𝑆 = (1 -𝑔 𝑚2 𝑍 𝑑𝑔2 ) • (𝑌 𝑠𝑤𝑖2 + 𝑌 𝑑𝑠2 ) 1 + 𝑍 𝑑𝑔2 • (𝑌 𝑠𝑤𝑖2 + 𝑌 𝑑𝑠2 )
To study the impact of this imbalance on the downconverter functionality, an analysis of the common-mode is performed. Post layout simulations shows that this imbalance does not affect the mixer performance as the common mode signal is fifty times lower than the differential IF signal.

c. N oise

M 2 IF- RF LO - Z L Z L IF+ LO + LO + M 1 A B C i n i n | M2 v n | A v n | IF+ v n | IF-
In practice, the thermal noise of M1 cannot be fully canceled [START_REF] Belostotski | No Noise is Good Noise[END_REF] in the millimeter-wave domain due to the phase imbalance induced by the different parasitics attached to each path.

The Table 11 proposes an overall summary of the mixer noise sources at an IF of 10MHz. Postlayout simulations show a thermal noise contribution of the common-gate transistor which is 2.5 times lower than the common source.

M easurem ent and sim ulations

This section studies the performance of the proposed mixer for automotive applications accounting for the input matching, the conversion gain, the linearity and the noise figure.

Measurements have been done in typical case but discussions regarding PVT variations are

proposed based on post-layout simulations before and after a calibration procedure. Post-layout simulations show a minor impact of supply voltage variation on the radar performances, less than 5% of variation compared to nominal conditions. For a sake of clarity, its impact is not reported on the presented figures.

(

1) Input m atching

The mixer exhibits a wideband input matching in post layout simulations, Figure 74(a The Table 12 reports the performance of some millimeter-wave mixers of the state of the art with their FOM defined in [START_REF] Park | Design and Analysis of Low Flicker-Noise CMOS Mixers for Direct-Conversion Receivers[END_REF]. The two receivers presented in [START_REF] Yang | A low noise, high gain, highly linear mixer for 77 GHz automotive radar applications in SiGe:C bipolar technology[END_REF] and [START_REF] Murakami | A 9 dB Noise Figure Fully Integrated 79 GHz Automotive Radar Receiver in 40 nm CMOS Technology[END_REF] exhibit a good RF performances but a large power consumption due to the bipolar technology for the first one and the architecture, with a LNA and a passive mixer, for the second one. The two active CMOS mixers in [START_REF] Shi | A 60-GHz broadband Gilbert-cell down conversion mixer in a 65-nm CMOS[END_REF] and [START_REF] Liu | A 62-90 GHz High Linearity and Low Noise CMOS Mixer Using Transformer-Coupling Cascode Topology[END_REF] presents an interesting trade-off between the overall performance and the power consumption. However, the NF is only evaluated in the thermal noise region at high IF, which is not relevant for the targeted application. The paper [START_REF] Kim | A 77 GHz Low LO Power Mixer With a Split Self-Driven Switching Cell in 65 nm CMOS Technology[END_REF] proposes a 77GHz active mixer with some measurement at an IF of 1MHz but suffers from a high NF and a poor gain. The circuit proposed in [START_REF] Ciocoveanu | Modified Gilbert-Cell Mixer With an LO Waveform Shaper and Switched Gate-Biasing for 1/f Noise Reduction in 28nm CMOS[END_REF]

IV . C onclusion on noise cancelling m ixer

Regarding the RX-RFFE specifications developed in Table 9

I. Introduction to conventional R F architecture

𝐹 𝑅𝑋 = 𝐹 𝐿𝑁𝐴 + 𝐹 𝑚𝑖𝑥𝑒𝑟 -1 𝐺 𝐿𝑁𝐴 + 𝑒 𝑉𝐺𝐴0 2 4𝑘𝑇 • 𝑅 𝑆 • 𝐺 𝐿𝑁𝐴 • 𝐺 𝑚𝑖𝑥𝑒𝑟 + 𝑒 𝑉𝐺𝐴1 2 4𝑘𝑇 • 𝑅 𝑆 • ∏ 𝐺 𝑉𝐺𝐴0 𝐿𝑁𝐴 + 𝑒 𝑉𝐺𝐴2 2 4𝑘𝑇 • 𝑅 𝑆 • ∏ 𝐺 𝑉𝐺𝐴1 𝐿𝑁𝐴
Considering the noise contributions and the gain of each stage, Figure 81

C onclusion on LN A Specifications

Hence the design of the LNA is led by a tradeoff between the linearity and the NF performance. To achieve a high ICP1 at system level, the gain of the LNA should be reduced, according Equation 43, which degrades the overall noise figure according Equation 42.

II. Single-balanced noise-cancelling LN A

In In this section, the conditions to design a common-gate-based noise-cancelling amplifier are first reminded. Finally, the architecture is discussed throughout post-layout simulations.

N oise-cancelling LN A m odelling

M 2

Out-

V bCS Out+ M 1 V bCG L 1 (a) Out- Z gd1 Z ds2 Out+ g m1 •v A Z gd2 Z gs Z ds1 -g m2 •v A (b) A A B B C C Common source path Common gate path L CG L CS R s v rf R s v rf

This section will discuss about the efficiency of this noise-cancelation technique in mm-waves

context regarding the input matching, the gain and the noise. To perform this analysis, the model presented in Figure 82(b) is considered, where M1 stands for the common-gate transistor and M2 for the common-source one. For sake of readibility, some preliminary equations are defined in Equation 46.

a) Input M atching

As detailed in Chapter 2.III.2, the input matching is dominated by the common-gate transistor. Nevertheless, in millimeter-wave, the parasitics as well as the common-source stage cannot be neglected. For this analysis, two paths are first considered, and an overall input impedance is then calculated. In Equation 16, the input impedance of the main path is expressed. We note if we neglect the capacitive contributions and output conductance in Equation 47, we find the conditions defined in Equation 17.

b) C onversion gain

As discussed in the introduction, the gain balance between the two paths is important to perform the cancellation of the common-gate noise and to ensure the integrity of the signal.

Equation 50 presents the conversion gain for each path. The gain balance, in terms of phase and amplitude, is tuned by adjusting the output loads Zdg1 and Zds2 through LCG and LCS respectively.

Equation 50 -Definition of conversion gain in each path

𝐴 𝑣𝐶𝑆 = 𝑣 𝑑𝑠2 𝑣 𝑔𝑠2 = 1 -𝑔 𝑚2 𝑍 𝑑𝑔2 1 + 𝑍 𝑑𝑔2 𝑍 𝑑𝑠2 ⁄ ; 𝐴 𝑣𝐶𝐺 = 𝑣 𝑑𝑔1 𝑣 𝑠𝑔1 = 1 + 𝑔 𝑚1 𝑍 𝑑𝑠1 1 + 𝑍 𝑑𝑠1 𝑍 𝑑𝑔1 ⁄ c) N oise figure
As introduced in Chapter II, the noise contribution of the main path, i.e. the commongate stage, can be cancelled, at least significantly reduced. To achieve the cancelation of the thermal noise contribution of common-gate transistor MCS, the common-gate and commonsource paths must be balanced. In millimeter-Wave, the balance between both paths is difficult to ensure due to the impact of parasitics. As consequences, the contribution of the CG thermal noise is not completely cancelled but still significantly reduced compared to the CS noise contribution. ( (3) N oise figure

P ost-layout sim ulations

2) C onversion gain and linearity

Single B alanced LN A and dow n converter perform ances

Introduced in [START_REF] Blaakmeer | Wideband Balun-LNA With Simultaneous Output Balancing, Noise-Canceling and Distortion-Canceling[END_REF], the common-gate based noise-cancelling architecture significantly improves the RF performances, summarized in Table 17, with a limited DC power consumption.

In comparison to the literature, referenced in

Power consumption (mW) / VDD (V) 4.6 / 0.6V < 5 / 0.6V < 5 / 0.6V < 5 / 0.6V

To discuss the performance of the LNA at system level, the Table 17

i n+ V n+,in+ Out + Out - R S /2 R S /2 v in+ v in- + + (a) (b) 2.
𝐼 𝑅𝐹 = 2 • 𝐼 𝑅𝐹,𝑝 • 𝑁 𝑅𝐹,𝑝 𝑁 𝑅𝐹 = 2 • 𝐼 𝑅𝐹,𝑛 • 𝑁 𝑅𝐹,𝑛 𝑁 𝑅𝐹 M n+ v out- v out+ M n- M p+ M p- r f r f v RF,n v RF,p + + A B C D v RF,p+ v RF,p- v RF,p v RF i RF i RF,p i RF
M n- g m,n •(v A -v B ) A B r ds,n r ds,n 2•R S /(2•n²) 2•R S /(2•n²) -v S /(2•n) +v S /(2•n) D C M p+ g m,p •(v D -v C ) M p- g m,p •(v C -v D ) r ds,p r ds,p 2•R S /(2•n²) 2•R S /(2•n²) -v S /(2•n) +v S /(2•n) Z L+ i out+ Z L- i out- v C v D v RF v A v B n:1 n:1 N RF,p N RF,n N RF 1:1 Out+ Out- i B i A i C i D M n+ g m,n •(v B -v A )
This section focuses on the modelling of the gain and the input matching of the C4 transconductance amplifier. For this analysis, a small signal equivalent model is proposed in

𝑍 𝑖𝑛,𝐴𝐵 = 𝑣 𝐴 -𝑣 𝐵 𝑖 𝐴 -𝑖 𝐵 = 𝑣 𝐴,𝑂𝑢𝑡+ -𝑣 𝐵,𝑂𝑢𝑡- 𝑖 𝐴 -𝑖 𝐵 + 𝑍 𝐿± = 𝑟 𝑑𝑠 + 𝑍 𝐿± 1 + 2𝑔 𝑚 • 𝑟 𝑑𝑠 ≈ 1 2 • 𝑔 𝑚
Considering the expression of the input impedance of the amplifier core and the transformer output impedance, a matching condition is then expressed in Equation 54. To achieve this matching, the transconductance of the transistor must be adjusted to the transformer coil ratio. However, the current consumption increases accordingly.

Equation 54 -Matching condition on gm expression

𝑍 𝑖𝑛,𝐴𝐵 = 𝑅 𝑆 𝑛 2 = 1 2 • 𝑔 𝑚 ⇔ 𝑔 𝑚 = 𝑛 2 2𝑅 𝑆 (2)

c. N oise analysis of C 4 LN A

Z L i out+ Z L i out- 𝑖 𝑛 𝑔 ,𝑝+ - | | - | | v C v D v A v B N RF,p N RF,n 1:1 𝑖 𝑛 𝑔 ,𝑝- 𝑖 𝑛 𝑔 ,𝑛- 𝑖 𝑛 𝑔 ,𝑛+ C D A B M n- g m,n •(v A -v B ) M p+ g m,p •(v D -v C ) M p- g m,p •(v C -v D ) M n+ g m,n •(v B -v A )
level, M7, M8 and AP exploited to implement the 3-coil transformer, together with the small foot print of the device at mm-Waves, contribute to achieve a low value of the winding sheet resistance. In this simplified analysis, two main sources of noise are considered. First, the noise contribution of amplification transistors, detailed for the Mn+ transistor. Secondly, the noise of the source modelled by noise sources and resistors seen through the input transformer. Equation 62 -Noise factor expression and To discuss the performance of the LNA at system level, the Table 20 Fortunately, the sum can be reduced, with a good approximation, to the two first terms, a0 and a1.

𝐹 = 1 + 4𝑖 𝑜𝑢𝑡 | 𝑛+ ̅̅̅̅̅̅̅̅̅ 2 𝑖 𝑜𝑢𝑡 | 𝑅 𝑆 ̅̅̅̅̅̅̅̅̅ 2 = 1 + 4 4 𝑖 𝑛 | 𝑛+ ̅̅̅̅̅̅̅̅ 2 ( 𝑛 𝑅 𝑆 ) 2 • 𝑣 𝑆 | 𝑅 𝑆 ̅̅̅̅̅̅̅ 2 = 1 + 4𝑘𝑇 • 𝛾 • 𝑔 𝑚 • 𝑅 𝑆 4 • 4𝑘𝑇 • 𝑛 2 = 1 + 𝛾 2

d. C 4 am plifier design trade -off

II. V oltage m ode dow nconverter

In this section, the architecture of a downconverter featuring a low noise amplifier based on a C4 amplifier and a fully differential passive mixer is first presented. The post-layout simulations of this front-end are then presented and discussed. Finally, a conclusion is proposed regarding the automotive radar application. ( The performances of the RX-RFFE are summarized in Table 23. This front end achieves (2) C onversion gain (3) N oise Figure

D em odulation chain

v out ± R l R l R f vdd v b,src v b,src v b,casc v b,casc v b,amp v bb+ v bb- C hpf C hpf R hpf R hpf v b,amp

P ost layout sim ulations

2) C onversion gain

P ost-layout sim ulations

1 . 3 . 1 . 2 . 1 . 1 . 2 . 3 . 1 . 2 . 1 . 2 . 3 . 1 . 2 . 3 . 2 . 3 . 1 . 2 . 3 . 1 . 2 . 3 . 1 . 3 .Chapter 5 .Finally, a quick description of the different automotive radars is proposed in section 3 . 1 .

 13121123121231232312312313531 Since few years, automotive industry implements more and more advanced driven assistance systems to improve safety and autonomy of vehicles. To achieve high autonomy level, each vehicle may include more and more sensors. The embedded 80GHz radar presents many advantages for car manufacturers: great immunity to outdoor conditions, real-time measurement of distance and velocity… To face such requirements, semiconductor industry has to simultaneously reduce cost and improve radar performances. It requires first to use advanced CMOS technologies (40nm, 28nm and smaller nodes). However, the main figures of merit should be improved such as the output power of the transmitter, the phase noise of the local oscillator and the noise figure of the receiver. The objective of this thesis is to propose new architectures and topologies able to improve the radar receiver performances. In an automotive context, the radar receiver face two main issues. Mounted behind the bumper, the radar module is impacted by the reflection of the transmitted signal. It may cause non linearities in the demodulation chain thus creating false targets. For this reason, the level of linearity is high, especially regarding the compression point and the limited front end (FE) gain. The improvement of the RADAR sensitivity requires both to maximize the FE gain and to minimize the noise figure (NF) of the receiver front-end. To face such contradictory specifications about the FE gain, this thesis works on more advanced design techniques and topologies. In this thesis, a specific focus is proposed on two design techniques. First, the circuits are designed to improve their efficiency (gm/ID) by operating in the moderate inversion region which contributes to reduce the power consumption. Secondly, the common-gate based noise cancelling technique is explored at 80GHz. This topology achieves simultaneously a wideband behavior and a low noise figure. It leads to develop a topology called "complementary capacitor cross-coupled" which achieves high RF performance at a reduced power consumption with respect to the operating frequency. Besides this thesis studies various mixing approach to determine the best suited for radar applications. First, considering its interesting performance trade-off in BICMOS technology implementations, active mixers are explored. Whereas the level of 1/f can allow for short range operations, it is too large for long range RADAR. To address this issue, passive mixers are then studied. They achieve better noise performances but severely impact the linearity performances. To overcome this drawback, an 80GHz current mode mixer featuring noise and linearity improvement is developed. K eyw ords: radar, low noise amplifier, mixer, millimeter-Wave, radiofrequency Acknowledgment This work would not have been possible without the support of many people. I would first like to thank my research advisor, Pr. Thierry Taris, for his strong support, availability and constant help during these three years. I would like to thank my industrial co-advisor, Dr. Gilles Montoriol for his ability to share his wide experience. I express all my gratitude to Mr. Xavier Hours and Mr. Christophe Landez for giving me the opportunity to work in their design teams and study this interesting field of automotive radar. I am thankful for the professional and personal support I received from many of my colleagues of NXP Toulouse design & validation teams which provides me a great technical support: Thierry, Cristian, Jean-Paul, Fabien, Jean, Birama, Stéphane, Jean-Stéphane, Pierre, Sébastien, Léo and Salime; and from NXP Eindhoven design team, especially: Kostas, Anton and Marteen. I would also like to thank all my colleagues at the IMS Laboratory, especially Magali de Matos, all the professors and associate professors of IMS Design teams -and all my PhD colleagues. I also thank the thesis committee members for accepting to participate as a jury and evaluate my research activities. First, I would like to thank Pr. Jean-Baptiste Begueret from Université de Bordeaux who acts as president of the jury. Second, I would express all my gratitude to Pr. Sylvain Bourdel from Université Grenoble Alpes and Pr. Domenico Zito from Aarhus University for reviewing my manuscript and providing interesting comments. Furthermore, I also thank the thesis examiner Christophe Landez and Xavier Hours from NXP Semiconductors. Finally, let me express all my gratitude to my parents, Ludovic and all my friends who have given me a constant support, encouragements and help during these three years. This work would not have been possible without them. Contents Chapter 1. 77GHz Embedded Automotive Radar .............................................................. I. Radars in automated vehicles ..................................................................................... Introduction to automated vehicles ........................................................................ a. Automated vehicles trends .................................................................................. b. Sensors in automated vehicles ............................................................................. c. Sensor data fusion ............................................................................................... 2. Automotive radar overview .................................................................................... a. Radar system ...................................................................................................... b. Expectations of automotive radar ....................................................................... c. Spectral occupation ............................................................................................. d. Automotive Radar Standards .............................................................................. Radar applications in automated vehicles ............................................................... a. Short Range Radar (SRR) .................................................................................. b. Medium Range Radar (MRR) ............................................................................. c. Long Range Radar (LRR) ................................................................................... II. FMCW Modulation Basics ..................................................................................... Distance measurement ............................................................................................ Velocity measurement ............................................................................................ 3. Conclusion .............................................................................................................. III. 77 GHz FMCW Radar IC ....................................................................................... Noise in automotive radar [17] ................................................................................ a. Maximum range regarding the demodulator NF .................................................. b. Noise distribution on RF demodulator chain ....................................................... c. Noise figure de-sensitization ................................................................................ c. Conclusion on noise specifications ....................................................................... Linearity in Automotive Radar [20] ........................................................................ a. In-band linearity (ICP1) ..................................................................................... b. Intermodulation (IIP3) ........................................................................................ c. Bumper impact on linearity (ICP1) .................................................................... Conclusion .............................................................................................................. IV. Considerations on receiver RFFE for automotive radar .......................................... 1. RF CMOS opportunities ......................................................................................... 2. mmW Front-End architecture ................................................................................ a. RFFE Basics ....................................................................................................... b. RF signal processing............................................................................................ Conclusion .............................................................................................................. V. Thesis outline ......................................................................................................... Chapter 2. 77-GHz Noise-cancelling active downconverter ................................................ I. Noise cancelling principle ........................................................................................... Why noise cancelling is interesting? ........................................................................ a. Limitation of noise matched structures ............................................................... b. Noise cancellation principle ................................................................................. Common-gate based noise cancelling ...................................................................... a. Impedance matching ........................................................................................... b. Gain balance ....................................................................................................... c. Noise cancellation ............................................................................................... d. Distortion cancellation ........................................................................................ e. Conclusion .......................................................................................................... II. RF Noise Cancelling Blixer ..................................................................................... Noise cancelling applied to mixer ............................................................................ a. Noise cancellation in a mixer ............................................................................... b. Attractive properties of Blixer ............................................................................ Noise figure improvement ....................................................................................... Low power improvement ........................................................................................ a. Auxiliary path improvement ............................................................................... b. Design in moderate inversion .............................................................................. c. Impedance matching ........................................................................................... I. Introduction to conventional RF architecture ......................................................... LNA specification: Noise Figure .............................................................................. LNA specification: Linearity ................................................................................... Conclusion on LNA Specifications .......................................................................... II. Single-balanced noise-cancelling LNA ..................................................................... 1. Noise-cancelling LNA modelling.............................................................................. a) Input Matching ................................................................................................... b) Conversion gain .................................................................................................. c) Noise figure ......................................................................................................... Post-layout simulations .......................................................................................... Single Balanced LNA and down converter performances ........................................ III. Complementary capacitor-cross-coupled amplifier................................................... Introduction to differential noise-cancelling ............................................................ a. Fully differential noise-cancelling LNA ................................................................ b. Capacitor-cross-coupled LNA .............................................................................. Principle of C4 LNA ............................................................................................... a. Modelling of the three-coil transformer ............................................................... b. Modelling of C4 LNA .......................................................................................... c. Noise analysis of C4 LNA ................................................................................. d. C4 amplifier design trade-off ............................................................................. Implemented mmWave C4 LNA ........................................................................... 4. C4 LNA Conclusion .............................................................................................. IV. Conclusion ............................................................................................................ Chapter 4. 77-GHz Downconverter ................................................................................. I. Introduction to CMOS passive mixer modelling ....................................................... II. Voltage mode downconverter ................................................................................ Demodulation chain .............................................................................................. a. Baseband analog signal processing .................................................................... b. Millimeter-Wave Voltage-mode Front-End ....................................................... Post layout simulations ........................................................................................ Conclusion ............................................................................................................ III. Current mode downconverter ............................................................................... Demodulation chain .............................................................................................. a. Baseband analog signal processing .................................................................... b. Millimeter-Wave Current mode front-end ......................................................... 2. Post-layout simulations ........................................................................................ Conclusion ............................................................................................................ IV. Conclusion ............................................................................................................ Conclusion ..................................................................................................... Bibliography ...................................................................................................................., automotive industry moves to safer and autonomous cars. In this way, car manufacturers integrate more and more advanced driver assistance systems to monitor car vicinity. In this chapter, some generalities of radar markets are first proposed, specific focus is then proposed on modern modulations for automotive radar. The third section details the constraints and specifications of a 77GHz radar receiver. Finally, some specific points are opened in the last section. I. R adars in autom ated vehicles Since few years, Automotive industry is hugely working on safer and more autonomous vehicles. The main trend of vehicle automation is presented in section 1 while the constraints on automotive radar are detailed in section 2. Introduction to autom ated vehicles This section proposes a brief overview of automated vehicle and draws the context where automotive radars take place. First, a top-level outline of automation levels is first proposed. To reach vehicles to the highest level, different sensors are proposed with their own advantages and drawbacks as presented in section b. Finally, section c details the different architectures of sensor data fusion.

than 4 billion dollars to the western economies between 2013 and 2030 [ 2 ]

 2 . According European Commission, transportation represents 25% of all European CO2 emissions [3] while environmental issues lead to change our mobility and reduced greenhouse gas emissions. To face those challenges, Automated Vehicles (AV) are a promising solution [4]. A more efficient driving should reduce both the risk of accident and the traffic congestion. Besides, safer vehicles allow car manufacturers to reduce the physical protections and, finally, their weight which is highly correlated to CO2 emissions. Finally, AV open up to new mobility paradigms.

Figure 1 -

 1 Figure 1 -Levels of Driving Automation defined by SAE J3016

  Lidar technology uses a pulsed laser to illuminate the target and to measure the reflected pulses to evaluate the distance. Achieving high accuracy, Lidar is used to make a digital map of the car environment. Nevertheless, lidar is not yet industrialized with concurrent technologies: solid-state lidar, MEMS mirrors, optical phase arrays… and remains expensive, up to more than ten thousand euros. Radar systems use the reflection of Electro-Magnetic (EM) waves on objects to determine its distance, velocity and angle. Thanks to its great immunity to outdoors conditions (luminosity, rain…) and its high detection range, radar is a key element for detecting objects. To replace Lidar solutions for 3D cloud mapping, Imaging Radars (IR) are a promising solution. Implementing a Multiple-Input Multiple-Output architecture, IR provide an angular resolution lower than 1°. Even it's not as performant as lidar as illustrated in Figure 2, IR has a very competitive cost.

Figure 2 -

 2 Figure 2 -Cloud map of car environment with Lidar and high-resolution radar[START_REF] Ramasubramanian | Using a complex-baseband architecture in FMCW radar systems[END_REF]

Figure 3 -

 3 Figure 3 -Data fusion possible topologies[START_REF] Salle | Architecture of an autonomous vehicle[END_REF]

Figure 4 -

 4 Figure 4 -Example of Radar board with a Transceiver (a), Antennas (b), PMIC (c) and MCU (d)

Figure 5 -

 5 Figure5-Atmospheric attenuation in mm-Waves domain[START_REF] Sayed | Millimeter wave tests and instrumentation[END_REF]

Figure 6 -

 6 Figure 6 -Radar applications in automated vehicle

  Shorts range radars are designed to monitor the car vicinity and detect targets within 80 meters. Thanks to the use of numerous SRR, it's possible to create a 360° safety cocoon around the car with several applications such as surround view, pedestrian detection, park assist or side impact detection. To address these various applications, SRR has a high azimuth aperture angle, up to 90°, and need a high spatial resolution to detect simultaneously several targets. However, as autonomous cars are supposed to used more than ten radar modules, the power consumption and the price of SSR modules are critical. b. M edium R ange R adar (M R R ) Many applications such as blind spot detection, rear or forward cross traffic alert or autonomous emergency braking needs medium range radar with a higher detection range up to 160-m but a lower angular aperture. MRR performances represents a trade-off between SRR and LRR ones. c. Long R ange R adar (LR R ) Long range radars need to detect long distant targets up to 300 meters. The applications such as lane change assist, adaptive cruise control need a very narrow field of view with a small horizontal angle, around 5°, with strong constraints on object separation or angular resolution.

Figure 7 -

 7 Figure 7 -FMCW modulation among time in frequency domain (a) and signal for transmitted signal (b)

Figure 8 -

 8 Figure 8 -FMCW modulation for Automotive Radar regarding the (a) RF Frequency and (b) the frequency difference fbeat

Figure 9 -

 9 Figure 9 -Example of Doppler effect

Figure 10 -

 10 Figure 10 -Two targets situation

Figure 11 -Figure 12 -Equation 2 -Equation 3 -

 111223 Figure 11 -Slow chirp FMCW modulation in a case with multiples targets for (a) transmitted and received signals and (b) velocity and range system resolution

Figure 13 -

 13 Figure 13 -2-D image of car vicinity in a situation with multiple targets and the ley parameters of radar performance expecting the angular ones[START_REF] Ramasubramanian | Using a complex-baseband architecture in FMCW radar systems[END_REF]

Figure 14 -

 14 Figure 14 -Improving the range resolution regarding the RF bandwidth;

Figure 15 -

 15 Figure 15 -Maximum range regarding the beat frequency

Figure 16 -

 16 Figure[START_REF] Dudek | The Impact of Phase Noise Parameters on Target Signal Detection in FMCW-radar system simulations for automotive applications[END_REF] -Architecture of automotive radar module[START_REF] Souria | Conception d'interfaces boitiers innovantes pour le radar automobile 77-GHz. Application à la conception otpimisée d'une chaine de réception radar en boitier[END_REF]

Figure 17 -

 17 Figure 17 -Simulation of the influence of phase noise on target signal quality when a distant target approaching to a 75m object considering a LO with: (a) a low phase noise; (b) a high phase noise[START_REF] Dhar | Modeling and analysis of the effects of PLL phase noise on FMCW radar performance[END_REF]

, atmospheric losses are negligible at 77GHz and the losses are mainly due to

  natural dispersion of the EM wave. The purpose here is to determine the maximum range due to the signal treatment. Three phenomena are then considered: the propagation of radar signal considering radar module characteristics, the noise distribution overall the demodulation chain, the de-sensitization of the receiver due to cross-talk between transmission and reception.

Figure 18 -

 18 Figure 18 -Schematic of radar signal attenuation in use case

Figure 19 -Figure 20 -

 1920 Figure 19 -Maximum achievable range of various targets regarding the RF demodulator NF for (a) long range radar and (b) short range radar

Figure 20 .

 20 Figure 20. Conventional downconverters have a noise figure from 10 to 16dB in the worst case.Considering this typical value, the impact of RX-RFFE gain is critical. According to Figure21, GRX-RFFE, should exceed 10dB to significantly moderate the noise contribution of baseband circuit to NFDEMOD. In the following analysis, a gain of 12.5dB will be considered.

Figure 21 - 7 . 7 -Figure 22 -Equation 8 . 8 - 2 (

 217722882 Figure 21 -RF Demodulator noise performance regarding RX-RFFE characteristics

Figure 23 -

 23 Figure 23 -Effect of de-sensitization on (a) Noise figures for various transmitted powers; (b) Maximum achievable range regarding the transmitted power for various NFRX-RFFE

Figure 24 -

 24 Figure 24 -Maximum range regarding the RFFE noise figure for various targets for (a) long range applications; (b) short range applications

Figure 25 -Equation 9 -Figure 26 -

 25926 Figure 25 -1dB compression point definition

Figure 27

 27 Figure 27 -IIP3 definition

Equation 10 -

 10 intercept point (IP3) is a figure-of-merit which evaluate this intermodulation. The IP3 is a hypothetical point at which the power of the third order harmonic rise to the fundamental harmonic power. This parameter can be referred at the input (IIP3) or at the output. The in-band IIP3 specification is expressed in Equation 10 considering the RX chain gain and is estimated at -26dBm. In band IIP3 specification 𝐼𝑛𝑏𝑎𝑛𝑑 𝐼𝐼𝑃3 (𝑑𝐵𝑚) = 16 -𝐺 𝑅𝑋 ≈ -26 𝑑𝐵𝑚 c. B um per im pact on linearity (IC P1) According equations Equation 9 and Equation 10, the intrinsic specifications for inband linearity of a radar RX are not tough. However, these two equations don't consider the environment of radar in the car. As radars antennas are mounted behind bumpers, the transmitted signal may be reflected by the bumper to the radar receiver. Intensive researches investigate the development of invisible bumpers, so far the reflection is limited to -20dB over the entire 76-81GHz band as shown in Figure 28.

Figure 28 -

 28 Figure 28 -Transmission and reflection through a bumper with metallic paint[START_REF] Blöecher | Impact of system parameter selection on radar sensor performance in autoimotive applications[END_REF]

Figure 29 -

 29 Figure 29 -Spectrum at Front-End input

Figure 30 -

 30 Figure 30 -IF spectrum with intermodulations due to bumper

Figure 31 -Figure 32 -Figure 33 -

 313233 Figure 31 -Transition frequencies of CMOS & BiCMOS technologies

Figure 34 -Equation 11 -

 3411 Figure 34 -Radar mmW typical architecture

large majority of recent radars use a frequency down conversion based on in-phase signal processing as illustrated in Figure 34 .

 34 This strategy is mainly motivated by a lower complexity and a reduced cost due to an implementation for baseband analog[START_REF] Ramasubramanian | Using a complex-baseband architecture in FMCW radar systems[END_REF]. As discussed in section 1, the evolution of radar module specifications requires an overall improvement of RF performances. To address this challenge, a better RF signal processing is required. To leverage such challenge a complex (or I-Q) mixer is investigated in radar receivers. This kind of architecture presents in Figure35, is based on a parallel processing, in-phase and inquadrature, in the down conversion operation to cancel out the image contribution fIM of the RF signal fRF. I-Q downconversion improves in theory the Noise Figure by 3dB. Indeed, a real receiver downconverts the noise at image frequency, the image band foldback is cancelled with a I-Q configuration. In practice, the benefit is limited due to signal loss in the I-Q imbalance due to the implementation. Inherent to the image cancellation the I-Q signal processing also mitigates the intermodulation occurring in RFFE and improve "bumper rejection" and interference tolerance. This feature can be exploited for better safety monitoring. Besides, it mitigates the noise degradation due to TX/RX crosstalk.

  overall trend to integrate all the digital blocks with the RF transceiver leads to select highly integrated CMOS technologies for the next generation of radar. New receiver architectures are investigated to address the challenging specifications inherent to radar modules. Various RX implementations (passive or active, voltage and current mixing) will be discussed throughout this manuscript using noise cancelling approach for simultaneously wideband behavior and noise performances. V. Thesis outline Due to the development of automotive full automation, new generations of 77GHz radar require both a high level of digital integration and a low-cost development which leads to the selection of advanced 28nm RF CMOS technology. Bearing in mind the general specifications of power consumption, form factor or PVT robustness, the radar module presented at Figure 36 has four critical specifications: PLL phase noise, PLL linearity, TX output power and the RX noise figure. This work focusses on the design of downconverters which are critical for noise specifications.

Figure 36 -Table 9 -Embedded 3 ,Equation 13 -

 369313 Figure 36 -The RX-RFFE in the radar module

Figure 37 -

 37 Figure 37 -Example of 94-GHz BiCMOS LNA noise and gain circle in Smith chart[START_REF] Afroz | Power-Efficient W-Band (92-98 GHz) Phased-Array Transmit and Receive Elements With Quadrature-Hybrid-Based Passive Phase Interpolator[END_REF]

Figure 38 -

 38 Figure 38 -Noise cancelling principle

Figure 39 -

 39 Figure 39 -Common-gate based noise cancelling LNA in (a) schematic; and (b) small signal model

Equation 15 -

 15 A, B and C are proposed in Equation 15 considering the transistors M1 and M2 and the passive elements. For sake of clarity, the imaginary parts of admittances are neglected. Equations of small signal model impedances

Equation 17 -Figure 40 -

 1740 Figure 40 -Signal in common-gate based amplifier

Equation 19 -Equation 20 -Figure 41 -Equation 21 -

 19204121 Figure 41 -Common-gate noise in a noise cancelling LNA

Figure 42 -

 42 Figure 42 -(a) Schematic of the common-gate path; (b) Small signal equivalent model of the common gate path; (c) Small signal equivalent model of the common-source path

Equation 23 -

 23 Voltage at the input with its decomposition in a linear and unwanted nonlinear term 𝑣 𝐶𝐺,𝐴 = ∑ 𝑎 𝑛 𝑣 𝑠 𝑠 + 𝑣 𝑁𝐿 Considering the expression of the approximated gain of both paths detailed in section b, the output voltages of CG and CS stages are expressed in Equation 24. When the difference of output voltages is performed, the nonlinear terms is then cancelled, and only linear terms remain. It leads to improve the linearity of the amplifier for weak non linearities of commongate path. Still the non-linearities of the CS path remains like was the noise contribution. Equation 24 -Voltages at the output (nodes B & C) due to nonlinear behavior

  , a brief analysis of common-gate based noise cancelling amplifier is proposed. In this topology, a common-gate amplifier acts as a main path while the auxiliary path uses a common-source for cancelation. A wideband real input impedance can be achieved by adjusting common-gate transconductance. Besides, to maintain a balanced operation the signal must experience the same conversion gain in each path. Considering these two design conditions, the channel noise current generated by the common-gate transistor is then cancelled at the output. In literature, this topology presents several advantages. First, a wideband behavior is assured thanks to the common-gate amplifier as expected from a non-resonant matching. Simultaneously, the noise of this path is cancelled by the auxiliary path, and the noise figure can be reduced with specific design of the CS path. Besides, this architecture offers an inherent single-to-differential behavior which could lead in some cases to superior performances with respect to fully differential amplifier solutions with balun. II. R F N oise C ancelling B lixer Until few years, the emergence of Internet of Things (IoT) leads to change paradigms on connectivity and mobility. The development of multiple sensor platforms and the improvement of their connectivity induce major evolutions in application fields such as: smart cities, transportation, infrastructure, healthcare or industry. In order to integrate more and more sensors in wireless local networks, multiples standards are proposed: Bluetooth, ZigBee, Wi-Fi Ha-Low [27]. To address the challenge of interoperability in IoT deployment, low power multistandard radios are mandatory. Substantial efforts are dedicated to reduce the cost and the form factor by minimizing the number of off-chip devices, and by increasing the silicon integration. Besides the constrain on the form factor drastically limits the power supply capability of the devices. As consequences, the power efficiency is of utmost importance in the development of Radio modules. In the transmitter part, this issue is addressed by the class of operation of the power amplifier. In the receiver (Rx) part, all the functions featuring the RF front end are concerned, and among the most critical ones is the frequency down-conversion. Previously, in section I, the principle of noise cancelling applied to amplifier is detailed and literature shows competitive performances. Combining an active mixer with a noise cancelling transconductance amplifier, "Blixer" topology can achieve very high performances over a wide bandwidth [28]. Originally proposed for software-defined wideband radio receivers, mixer-first architectures using noise cancelling principle can perform low noise figure and high linearity as requested by the multiple standards of the RF band. In this section, a Blixer is proposed to address low power IoT market with specific techniques to reduce the power consumption.

Figure 43 -

 43 Figure 43 -Basic topology of blixer

Figure 44 -

 44 Figure 44 -Noise cancellation in mixer

Figure 45 - 2 𝑛

 452 Figure45-Gain over RF band for the noise cancelling LNA, basic Blixer and I/Q-Blixer[START_REF] Blaakmeer | The Blixer, a Wideband balun-LNA-I/Q-Mixer Topology[END_REF]

Figure 46 -Figure 47 -

 4647 Figure 46 -Impact of topological factor (n) on NF (dB) and normalized power (no units)

Figure 48 -

 48 Figure 48 -Different FoM of 28-nm transistor regarding Vgs: (a). fT & gm/ID; (b). gm.fT/ID

Figure 49 -•Figure 50 -Figure 52 .

 495052 Figure 49 -Input matching (S11) between 1-GHz and 6-GHz for various gm,CS values and a source resistance Rs = 50-Ω

Figure 51 -

 51 Figure 51 -Schematic of proposed mixer

Figure 52 -

 52 Figure 52 -Micrograph of the implemented downconverter

Figure 53 -

 53 Figure 53 -Input matching: (a) for measurement in both modes; (b) for post-layout simulations in PVT variations for nominal modes

Figure 54 -

 54 Figure 54 -(a) Conversion gain and IIP3 in measurement for both modes; (b) Conversion gain in post layout simulations in PVT for nominal mode

Figure 55 -

 55 Figure 55 -Single side band NF (a) in measurement for both modes; (b) in post layout simulations with PVT variations for nominal mode

  , the proposed mixer achieves a low noise figure, close to the best performance of the referenced works which are I/Q architectures. The silicon footprint is the smallest with [9]. Some references such as [30] and [33] are designed to achieve very low power but suffer from a dramatically reduced bandwidth. The proposed mixer covers a large bandwidth at a moderate power consumption. This first RF mixer dedicated to IOT applications demonstrates the inherent wideband behavior and active balun operation of a noise canceling architecture. It further allows to experiment some circuit techniques applied to frequency down conversion: the reduction of the NF in a noise cancelation topology by beta factor balancing, the improvement of the current efficiency in a transconductance stage throughout current reuse and transistor biasing. Some of these techniques will be further exploited to implement mm-Wave circuits when it is possible.

1 . 1 𝑓

 11 BiCMOS technologies, RF Front-End usually features active mixers due to their interesting trade-off between gain, noise figure and linearity. As discussed in Chapter I.1.2.2, a -5dBm compression point is critical for rear target measurement while a high conversion gain, typically higher than 10dB, is necessary to reduce the noise contribution of baseband chain. This section explores the architecture of active mixer for 77GHz downconverter in RF-CMOS technologies. As radar may work in diverse environments, the downconverter may have enough robustness to process corners, to operate over a large range of temperature and to support a variation of its supply voltage of +/-5%. According section II, the noise cancelling topology based on common gate presents inherent wideband operation and, for this reason, a better immunity to PVT variations. Some techniques, such as the beta factor and the current-reuse OTA, implemented in the RF demonstrator cannot be exploited at millimeter-Wave due to the large parasitic capacitances at critical nodes. However, the biasing of transistors in moderate inversion is considered when it is possible. The issue of 1/f noise in CMOS technologies is first explored for active mixers and some techniques to reduce its impact are investigated. The second part proposes to model the noise cancelling mixer topology for millimeter-Wave applications. Finally, the implementation of a noise cancelling mixer with flicker noise reduction at 77GHz is proposed with post-layout simulations in PVT corners and measurement results in typical case. Flicker noise in C M OS active m ixers For many years, the flicker noise is identified as one of the main issues of CMOS active mixers [35]. A lot of work has investigated on it in order to reduce its impact on mixer performance. This sub-section first details the generation of 1/f noise in a switching pair throughout direct and indirect mechanisms. Then, it explores the different techniques to reduce it and discusses their interest in the millimeter-Wave domain. Finally, the last part focuses on the techniques exploited in the implemented mixer. a. Flicker noise in C M OS sw itching pair Equation 28 -Gate-referred flicker noise voltage where KF is a process parameter, Cox is the oxide capacitance, W and L are the width and length of the transistor and f the operating frequency 𝑉 𝑛 (𝑓) = 𝐾 𝑓 𝑊 𝐿 𝐶 𝑜𝑥 • In MOS transistors, flicker noise originates from electron tunneling through traps in the oxide to the gate causing fluctuations through carrier number and mobility, and consequently drain current flickering. The gate-referred noise voltage (Equation 28) increases with the reduction of operating frequency. In 28nm RF CMOS technology, it becomes predominant below 10-MHz which corresponds to the low-end of radar IF bandwidth. As radar solutions use thinner CMOS technologies to increase digital integration and transition frequency, flicker noise tends to increase.

Figure 56 -( 1 )Figure 57 - 2 •

 561572 Figure 56 -Switching pair considered in the flicker noise analysis

Figure 58 -

 58 Figure 58 -Mixer output direct flicker noise current (a) Time model of the noise pulse train with a sampling Dirac comb and the vn flicker noise; (b) Frequency model of the noise pulse train

( 2 )Equation 30 - 2 Equation 31 -

 230231 Figure 59 -(a) Switching pair with LO sine waveform; (b) Source follower model with a rectified LO sine waveform; (c) Rectified sinewave linked to LO waveform.

3 )

 3 Indirect m echanism due to gate noise The analysis of the two previous mechanisms of flicker noise generation suggests that a square-wave LO waveform would avoid the generation of flicker noise in an active mixer. Unfortunately, the indirect mechanism keeps on generating flicker noise due to the source follower behavior even with a square-wave drive. For this analysis, the signal at switches input can be decomposed on the superposition of the flicker noise voltage and the LO waveform which can be sinus or square wave. Without loss of generality, the Figure 60(a) features a LO square waveform on M1 and M2 transistors and a flicker noise voltage. These noise voltages are fully uncorrelated, the flicker noise of the overall switching cell is associated to M1 transistor without loss of generality.

Figure 60 -

 60 Figure 60 -(a) Switching pair with LO sine waveform and the vn gate noise; (b) Source follower model with modelized noise during ON state; (c) Tail voltage and currents due to vn noise

( 4 )

 4 C onclusion on flicker noise As presented in this section, the flicker noise is generated by three different mechanisms due to the LO waveform and the parasitic capacitance in the RF signal path. For long channel CMOS devices, the indirect mechanism is negligible, but it becomes predominant for millimeter-Wave applications in advanced RF-CMOS. previous analysis, a solution to reduce the generation of flicker noise is to reduce the width of noise pulse which requires to increase the LO slope and the power consumption of the LO generation circuit. Another solution is to design larger switches, this approach reduces the operating frequency of the mixer due to the increase of parasitics. The reduction of flicker noise at high frequency calls for circuit techniques. b. Flicker noise reduction techniques To overcome the issue of mixer noise at low IF, several techniques have been proposed in the literature to reduce the generation of flicker noise. This section proposes an overview of these techniques and discusses their interest in millimeter-Wave domain.

Figure 61 -

 61 Figure61-Quadrature active mixer with a 2•fLO extra-switches[START_REF] Pullela | Low Flicker-Noise Quadrature Mixer Topology[END_REF]

Figure 62 -Q

 62 Figure 62 -(a) Dynamic current injection in active mixer with signals waveforms; (b) Measurement results ofNFDSB (dB) regarding intermediate frequency (Hz) at 2GHz with and without noise reduction technique[START_REF] Darabi | A Noise Cancellation Technique in Active RF-CMOS Mixers[END_REF]

Figure 63 -

 63 Figure 63 -(a) Schematic of an active mixer with a negative impedance; (b) Measurement results of NFDSB (dB) regarding intermediate frequency (Hz) at 0.9GHz with and without noise reduction technique

Figure 64 -

 64 Figure 64 -(a) Switching pair with (b) LO square waveform with OFF state in accumulation generation; (c) Measurement result of NFdsb (dB) regarding Intermediate Frequency (Hz) at 60GHz with and without noise reduction technique

Figure 65 -Equation 34 -Figure 66 -Equation 35 -

 65346635 Figure 65 -Switching pair with a current bleeding in (a) schematic; (b) model

Figure 67 -

 67 Figure 67 -(a) Imaginary part and (b) real part of tail impedance over the 76 to 81GHz band

Figure 68 -Figure 69 -

 6869 Figure 68 -Noise figure regarding the intermediate frequency for the mixer with and without the flicker noise reduction techniques (post-layout simulation)

36 . 36 -Equation 37 -

 363637 equivalent model of the transconductance stage is presented in Figure 69(b). The transconductance core is composed of an auxiliary path with a CS transistor (M2) and a main path with a CG transistor (M1). The core input admittance Ycore is derived in Equation Equation Admittance of mixer core without access and input inductor 𝑌 𝑐𝑜𝑟𝑒 = 𝑌 𝐶𝐺 + 𝑌 𝐶𝑆 = 𝑗𝜔(𝐶 𝑔𝑠1 + 𝐶 𝑔𝑠2 ) + 1 + 𝑔 𝑚1 𝑍 𝑑𝑠1 𝑍 𝑑𝑠1 + 𝑍 𝐵 + 1 + 𝑔 𝑚2 𝑍 𝐶 𝑍 𝐶 + 𝑍 𝑑𝑔2 In section III.1.c, the impedance at node C (ZC) is assumed real at the operating frequency and closed to 40Ω. As the drain-to-gate impedance is significantly higher (Cgd2 = 2.7fF), the impedance of common source path is dominated by the gate-to-source capacitance (Cgs2). For common-gate path, the impedance at node B is also closed to 50Ω. Then the CG path input impedance is mainly controlled by gm1 and impacted by the gate-to-source capacitance (Cgs1). The core input admittance is then simplified to Equation 37. Simplified admittance of mixer core 𝑌 𝑐𝑜𝑟𝑒 = 𝑗𝜔(𝐶 𝑔𝑠1 + 𝐶 𝑔𝑠2 ) + 𝑔 𝑚1 Considering this core impedance, the inductor LIN is sized for input matching accounting for the pad model. Figure 70 proposes a representation on Smith chart of the different impedances considered in Figure 69(b). To ensure an input return loss, S11 lower than -10dB, the input impedance Zin must be located inside the dashed circle over the frequency band. The model of the circuit, presented on Equation 36, shown in Figure 69 leads to a similar input matching (green S11).

Figure 70 -Equation 38 -

 7038 Figure 70 -Smith chart of the input impedance from core impedance to measured impedance on pad on 76 to 81GHz band

Figure 71 -

 71 Figure 71 -Time domain behavior of mixing switches in post layout simulation (solid line) and square wave model (dashed line)

Figure 72 -Equation 40 -

 7240 Figure 72 -Noise cancelling principle in mixers during a LO cycle

Figure 73 -

 73 Figure 73 -Chip micrograph with measurement setup by Y method

  ), from 68 to 83GHz corresponding to a 20% bandwidth of the central frequency. The variation on return loss and central frequency, is mainly due to process variations which impact the common gate transconductance through the threshold voltage and the process parameters. This mixer achieves a sufficient matching on the radar band (76-81GHz). The calibration doesn't significantly impact the matching of the proposed mixer (Figure 74(b)). The measured S11 (Figure 74(c)) is higher than expected due to an underestimation of input pad parasitics. Some re-simulations have been down and show a similar behavior.

Figure 74 -

 74 Figure 74 -(a) PVT post-layout simulations of return loss; (b) PVT post layout simulations of return loss after calibration; (c) Measurement result and re-simulated result of return loss

Figure 75 -( 3 )Figure 76 -( 4 ) N oise figureFigure 77 -

 75376477 Figure 75 -(a) PVT post-layout simulation of conversion gain; (b) PVT post layout simulations of conversion gain after calibration; (c) Measurement result and re-simulated result of conversion gain

Figure 78 -Equation 41 -

 7841 Figure 78 -(a) PVT post-layout simulation of NFSSB (dB) versus IF at fRF = 78.5GHz; (b) PVT post layout simulations of NFSSB (dB) versus IF at fRF = 78.5GHz; (c) Measurement result and re-simulated result of NFSSB (dB) versus IF at fRF = 78.5GHz

  demonstrates a flicker noise reduction technique based on LO signal waveform shaping. It achieves a good noise performance at 60GHz but presents a low conversion gain with respect to the power consumption. The proposed mixer achieves a good performance trade-off compared to the state of the art. Besides it exhibits the smallest silicon area.

  roughly stable regarding the RF frequency for a high IF which is compatible with short range application. Nevertheless, it is sensitive to the intermediate frequency which is correlated to the target distance. As reported in Figure 15, low part of the IF spectrum corresponds to the closed targets. For those ones, the noise specification is relaxed since the input signal-to-noise ratio (SNR) is large. The Figure 79 proposes an evaluation of the achievable distance for a short-range radar module considering the analysis proposed in Chapter I.3.1. The grey line corresponds to the maximum measurable distance for various type of target. For large size targets, such as cars, it is possible to detect them up to 80-meters. For smaller targets, such as bikes and pedestrians, the RFFE limits the achievable range to respectively 40 meters and 15 meters. Hence the proposed downconverter can be exploited for a short-range use case.

Figure 79 -

 79 Figure 79 -Maximum achievable range (m) for various targets versus the intermediate frequency IF (MHz) at RF frequency of 78.5GHz for (a) measurement results; (b) worst case in post-layout simulations

, this chapter proposes the development of a 77GHz radar

  downconverter. Considering the requirement of high linearity and gain, an active real mixing is preferred. The topology is based on a noise canceling architecture which achieves a large voltage gain with balun operation and a good linearity, in terms of compression and intermodulation. The wideband behavior of a noise canceling configuration contributes to improve the circuit robustness to PVT variations. In order to validate the concept of noise canceling in frequency conversion, as well as various circuit techniques dedicated to improve the performances, a first demonstrator applications, namely RF blixer is developed and implemented in 28nm CMOS. The circuit exhibits competitive performance, reported in Table 13, regarding the state of the art. The RF mixer allows to demonstrate at circuit level: the concept of noise cancelation in mixer implementation, the reduction of the NF through path imbalance (beta factor), the improvement of the current efficiency in a transconductance stage with a current reuse configuration; and at device level: the biasing of MOS transistor in moderate inversion region to improve the figure of merit gm.fT/ID which represents the current efficiency accounting for frequency performance

  techniques experimented in the design of the RF mixer are then exported, when it is possible, in the implementation of the mm-Waves mixer. Notably the architecture exploits the same noise canceling configuration based on a common gate path and a common source path. Complementary approaches, specific to the application, are developed to reduce the flicker noise. After a comprehensive analysis of flicker noise generation in frequency down conversion process, two techniques are proposed to reduce the noise figure at low IF : -at circuit level -the configuration of current bleeding to reduce the DC component in the LO switching stage and a parallel inductor to cancel parasitic capacitances -at device level -the biasing of LO transistors in sub-threshold region to reduce the noise voltage at the gate The proposed downconverter, implemented in 28nm CMOS technology, achieves a wideband matching, a high gain for a moderate noise figure as reported in Table 13. Likewise the RF mixer, the mm-Waves mixer compares favorably with the state-of-art. It achieves among the best trade-off between performances accounting for the power consumption, and the silicon footprint. The analysis of radar RFFE performance featuring such active mixer figures out this solution can address short range radar applications. However, for a long-range radar module, it seems difficult for an active downconverter to achieve enough low noise figure to target the most distant targets. Chapter 3. 77-GHz low noise amplifier In Chapter I, stringent specifications are determined for RX-RFFE, especially regarding noise and linearity. Chapter II demonstrates the limitations of active downconverters to address long-range radar requirements, especially at low intermediate frequency (IF). To improve the performance at low IF we further investigate the development of an RX-RFFE combining a Low Noise Amplifier (LNA) and a passive mixer. This chapter focuses on the design of low noise amplifiers in advanced RF-CMOS featuring common-gate based noise-cancelling. The first section proposes to define the LNA specifications considering the different elements featuring the RX-RFFE. A single-to-differential noise-cancelling amplifier is proposed and simulated in section II. A new architecture called "complementary capacitor-cross-coupled amplifier" is introduced in section III. This LNA improves the overall current efficiency and the noise performance compared to conventional topologies. Finally, the section IV draws the conclusion about the proposed LNAs regarding the state-of-art.

Figure 80 -Figure 80 .

 8080 Figure 80 -Radar RX-RFFE typical architecture

  proposes an application of the Friis formula for the calculation of the demodulation chain NF depending on the LNA RF specifications. To achieve long range requirements, the noise figure of the overall RX-RFFE should not exceed 15dB. To address this constraint, and considering the characteristics reported in Table14for the RX-RFFE, the LNA would achieves more than 10dB of gain, and a NF below 7dB.

Figure 81 -Equation 43 -

 8143 Figure 81 -Noise figure of the overall chain regarding the noise figure and the gain of the LNA

  Chapter II, the common-gate based noise-cancelling topology for mixer is studied. It presents interesting performances regarding the bandwidth, the noise figure and the linearity. As discussed in Chapter 2.III, active mixers cannot perform low noise figure, especially at low IF. Nevertheless, the noise-cancelling principle has been widely investigated for RF amplification [26]. This section proposes to exploit this technique for the implementation of a 77-GHz low noise amplifier.

Figure 82 -Equation 44 -Equation 45 -

 824445 Figure 82 -Common-gate-based noise-cancelling amplifier in (a) schematic with m matching condition & (b) in schematic with gain imbalance condition

Combining Equation 47 and

 47 Equation 48, the overall input impedance of the noisecancelling amplifier is expressed in Equation 49. It includes the gate-source capacitances of M1 and M2, and the input parallel inductor L1, Figure 82(a), modeled by Zgs in Figure 82 (b). With a proper sizing of such devices, it is possible to cancel the imaginary part of Zin and to match a source impedance Rs.

Figure 83 -( 1 )

 831 Figure 83 -Layout of the proposed noise-cancelling LNA

Figure 85 -

 85 Figure 85 -PT post-layout simulation of: (a) conversion gain (Gv), 1dB compression point (ICP1)

Figure 86 -Figure 87 -

 8687 Figure 86 -PT post-layout simulation of NF

reports an evaluation of a downconverter 1 .Figure 88 ,

 188 Figure 88, to address it. The topology proposed in Figure 88 is technically a pseudo-differential circuit since the transistor sources are not tied to a current source. The reason is the parasitics of the current source has more drawbacks than benefits on circuit performances at mmWaves. For sake of simplicity we will further use the word differential instead of pseudo-differential for mmWaves circuits. The conditions for noise cancellation are the same as the ones described for the single balanced topology of section II considering a 100Ω input impedance. The input/output transformers are exploited as two folds: a) to isolate the circuit bias from

Figure 88 -

 88 Figure 88 -Differential noise-cancelling amplifier with input and output transformers (biasing not shown)

Figure 89 -

 89 Figure 89 -Capacitor-cross-coupled amplifier with input and output transformer (biasing not shown)

Figure 90 -

 90 Figure 90 -(a) RF signal and gain in cross coupled topology; (b) Noise cancellation in a cross coupled topology for the Mn+ noise

  P rinciple of C 4 LN A The section II demonstrates the interest of noise-cancelling technique in common-gate based architectures for the implementation of LNA in mm-Waves domain. It also highlights the limitations due to the inherent asymmetry of the original architecture. A fully differential implementation has been discussed in section III.1 to reduce the sensitivity of the balanced operation to PVT corners. A capacitive cross coupled arrangement of the differential architecture has been also considered to improve the current efficiency. This section focuses on a novel version of the capacitive cross coupled namely complementary capacitor-cross-coupled amplifier.

Figure 91 -Figure 91 .Figure 91 .

 919191 Figure 91 -Complementary capacitor-cross-coupled amplifier with an input transformer

Figure 92 -Figure 93 -

 9293 Figure 92 -Gain and RF signal in the cross coupled cell

Equation 52 -𝑍 2 ; 1 𝑛

 5221 equations, Equation 52 proposes a calculation of the impedance shown at secondary outputs and the associated voltages. As both secondary coils are identical, a transforming ratio (n) is considered between primary and secondary windings. This ratio may be considered for the sizing of the amplifier. Impedance and voltage at the secondary outputs considering a transforming ratio n 𝑉 𝑆1/2 = 𝑉 𝑝 • The back-end of the 28nm CMOS technology, illustrated in Figure 32, includes two thick copper layers (M7, M8) and a very thick aluminum (AP) layer which are exploited to implement the transformer as follows: the primary winding is laid out in AP layer and the two secondary windings are drawn in M7 and M8 layers. Figure 94 shows the layout of the transformer, its footprint is 62μm x 72μm.

Figure 94 -Figure 95 -

 9495 Figure 94 -Layout of the proposed transformer

Figure 95 .( 1 )Equation 53 -

 95153 Figure 95. For the sake of clarity, capacitive and inductive elements are neglected. As the PMOS and NMOS stages are symmetrical, their characteristics are assumed equivalent.

A m plifier transconductance Equation 55

 55 proposes an expression of the output current depending of the differential input voltage. If we assume a lossless transformer, the input RF power is fully transmitted through the transformer to the amplifier core.

Figure 96 -

 96 Figure 96 -Noise modelling of the C4 LNA

( 1 ) 2 FromEquation 60 -

 1260 Transistors noise Considering the noise of one transistor, for instance Mn+, the thermal noise current (𝑖 𝑛+ | 𝑛+ ) generates a noise voltage at node A (𝑣 𝐴 | 𝑛+ ), and through, the input transformer, a noise voltage at node B (𝑣 𝐵 | 𝑛+ ). Due to the transformer configuration, the differential noise voltage between A and B (𝑣 𝐴 | 𝑛+ -𝑣 𝐵 | 𝑛+ ) is also applied to the input of the complementary stage (Mp+, Mp-) as 𝑣 𝐶 | ,𝑛+ -𝑣 𝐷 | 𝑛+ . It leads to the overall expression given in Equation 58. Equation 58 -Noise output current expression of the M1 transistor 𝑖 𝑜𝑢𝑡 | 𝑛+ = (𝑖 𝑛+ -𝑖 𝑛-+ 𝑖 𝑝+ -𝑖 𝑝-)| 𝑛+ = 2𝑔 𝑚 (𝑉 𝐵 | 𝑛+ -𝑉 𝐴 | 𝑛+ + 𝑉 𝐷 | 𝑛+ -𝑉 𝐶 | 𝑛+ ) + 𝑖 𝑛+ | 𝑛+ with 𝑉 𝐴 | 𝑛+ -𝑉 𝐵 | 𝑛+ = 𝑉 𝐶 | 𝑛+ -𝑉 𝐷 | 𝑛+ = 𝑖 𝑜𝑢𝑡 | 𝑛+ • 𝑅 𝑆 2𝑛 Equation 58, we can derive the output current noise (𝑖 𝑜𝑢𝑡 | 𝑛+ ) due to Mn+ noise, Equation 59. Interestingly this output noise current (𝑖 𝑜𝑢𝑡 | 𝑛+ ) is half of the Mn+ noise current (𝑖 𝑛+ | 𝑛+ ). The C4 LNA performs a partial noise cancelation of the MOS thermal drain noise under input matching conditions. Equation 59 -Final expression of the output current of one transistor noise 𝑖 𝑜𝑢𝑡 | 𝑛+ = 𝑖 𝑛+ Noise output current expression of the source (RS) noise 𝑖 𝑜𝑢𝑡 | 𝑅 𝑆 = (𝑖 𝑛+ -𝑖 𝑛-+ 𝑖 𝑝+ -𝑖 𝑝-)| 𝑅 𝑆 = 2𝑔 𝑚 (𝑉 𝐵 | 𝑅 𝑆 -𝑉 𝐴 | 𝑅 𝑆 + 𝑉 𝐷 | 𝑅 𝑆 -𝑉 𝐶 | 𝑅 𝑆 ) with 𝑉 𝐴 | 𝑅 𝑆 -𝑉 𝐵 | 𝑅 𝑆 = 𝑉 𝐶 | 𝑅 𝑆 -𝑉 𝐷 | 𝑅 𝑆 = 𝑖 𝑜𝑢𝑡 | 𝑅 𝑆 • by the source is transferred to the amplifier core through the transformer. Equation 60 proposes an expression of the output current depending of the input voltages. Under input matching conditions, it can be simplified to Equation 61. Equation 61 -Final expression of the output current of the source (RS) noise 𝑖 𝑜𝑢𝑡 | 𝑅 𝑆 = 4 𝑔 𝑚 𝑣 𝑆 | 𝑅 𝑆 of noise contribution Equation 59 and 61 are used to define the C4 LNA noise factor F. Considering the matching conditions on transistor transconductance sizing, the noise factor does not depend on the topological factor (n) which makes this topology very interesting for the noise to power consumption trade-off. The minimum theoretical achievable noise figure is 1.25dB for a typical γ of 2/3. It is a reduction by a factor of 2.45 with respect to the theoretical NF of a conventional common-gate differential amplifier. This improvement corresponds to the partial noise-cancelling illustrated in Equation 61. In practice, the NF is larger accounting for all the sources neglected in this simplified noise analysis of the C4 LNA.

Figure 97 - 3 .Figure 98 -Figure 99 -( 1 )Figure 102 -Figure 103 -

 97398991102103 Figure 97 -Impact of transformer ratio on the figure of merit of the proposed amplifier

  reports an evaluation of a downconverter based on the proposed C4 LNA combined with a passive mixer in voltage mode. A current mode passive mixer would be better suited since the C4 topology is of transconductance amplifier type. However, for a fair comparison with the downconverter proposed in section II, the same mixer, further developed in Chapter IV, is exploited. The proposed downconverter presents a typical conversion gain of 11.5dB reduced to 10dB in worst case which is more than 3 dB higher than the downconverter of section II. The noise figure is now limited to 12dB in the worst case, it was 13.7dB for the previous downconverter. The downconverter linearity cannot be fairly compared due to the voltage operation of the mixer. Due to the advantages shown by the C4 LNA, this topology has been selected to build a RX-RFFE compatible with long range radar applications.

Figure 104 -

 104 Figure 104 -RX including C4 LNA with voltage mode mixer

  figures of merit are proposed: FOMI, Equation 63(a), only accounts for gain, NF, linearity and power consumption, and FOMII, Equation 63(b), which includes the silicon foot print. As discussed previously, conventional common-source amplifiers [5] cannot simultaneously achieve low noise, high linearity and enough gain with a low power consumption. Regarding multistages LNA, the two proposed circuits compares favorably in terms of noise figure, linearity and DC consumption, they exhibit a larger FOMI compared to other works. This trade-off is further increased considering the silicon area as illustrated with FOMII. Indeed, FOMII does not exceed 0.2 for the circuits of the state of the art, it is 5.3 for the NC LNA and 10.3 for the C4 LNA. The impact of the LNA performance on a downconverter has been also estimated. The two proposed LNA are combined with a voltage-mode passive mixer. The best combination with respect to the gain and noise figure, is the one based on the C4 LNA. For this reason, the next chapter investigates the development, and implementation, of a down converter with a

Figure 105 -

 105 Figure 105 -(a) Mixer core model with baseband impedance; LO switches behavior in time domain (b) and frequency domain (c)

Figure 106 -

 106 Figure 106 -Proposed Voltage-mode RX-RFFE in a complete demodulation chain

Figure 107 proposes the

  Figure 107 proposes the schematic of the cascaded high-pass filter with the baseband low noise amplifier VGA0. The first order high pass filter features a 15pF capacitor (Chpf) and a resistor (Rhpf) of 24.4kΩ, also used for gate biasing. For the baseband amplifier, a PMOS amplification (Mp) is selected to minimize the amplifier noise, the channel length is closed to 600nm to reduce the flicker noise contribution. For the DC current sources, a cascode approach is preferred to maximize the output impedance and recopy. The gain of the amplifier is set by the ratio between the load resistor (RL) and the feedback resistor (Rf). This feedback resistor is tunable to adjust the gain considering the various use cases. The amplifier consumes 5.7mW with a 1.5V supply voltage. The HPF-VGA0 is simulated in schematic and has not been implemented in the test vehicle. In Figure 108(a), a flat-band 10 dB gain is achieved from 2 to 100MHz, with an attenuation of 15dB at 20kHz. The bumper blocker is located at a frequency typically below 50kHz, the amplifier exhibits in this low band a high compression point, above +1.8Vrms, Figure 108(b). This performance allows the baseband section to address the overall linearity specification. Regarding the noise performance, Figure 108(c), a typical noise of 2nV/√Hz is achieved in the IF band, rising to 3 nV/√Hz in the worst hot case. The noise corner frequency of the proposed amplifier remains below 1MHz which does not impact the SNR of the IF signal.

Figure 108 -Figure 109 -

 108109 Figure 108 -Simulations of the baseband amplifier with (a) conversion gain in dB; (b) compression point in V; (c) input referred noise (nV/√Hz)

Figure 110 -Figure 111 -

 110111 Figure 110 -Layout of the millimeter-Wave front-end of the proposed voltage-mode downconverter

Figure 112 -Figure 113 -Figure 114 -

 112113114 Figure 112 -Conversion gain (Gc) in dB in process and temperature variations: (a) in RF band (76-81GHz) for a 10MHz IF; (b) in IF band (10kHz-100MHz) for a 78.5GHz

a noise figure below 14dB and a conversion gain higher than 15 .

 15 5dB in any case. These performances are included in the system model presented in chapter I to evaluate the RADAR performance, the TX section combines two transmitters achieving a +15dBm overall output power. Considering various target types, the proposed downconverter achieves a proper detection distance of more than 50 meters for pedestrians, 100 meters for bikes, 200 meters for cars and 1 km for trucks which is compatible with a long-range application. Nevertheless, voltage-mode mixing involves two limitations. First, the high real input impedance of the mixer is significantly impacted by parasitic capacitances leading to a reduced, and sliding, bandwidth in PVT corners. Secondly, the voltage mixing operation significantly impacts the linearity of the overall demodulator, leading to an ICP1 below -5dBm which is not compatible with radar specifications. The next section focuses on the RFFE design to address this issue.

Figure 115 -Figure 116 -Figure 117 -

 115116117 Figure 115 -Maximal range regarding the Intermediate Frequency for various targets in the worst case

Figure 118 -Figure 119 -

 118119 Figure 118 -Simulations of the baseband TIA with (a) TransImpedance in Ω; (b) 1dB compression point in mA; (c) input referred noise (nV/√Hz)

Figure 120 -( 1 )Figure 121 -

 1201121 Figure 120 -Layout of the millimeter-Wave front-end of the proposed current-mode downconverter

Figure 122 -

 122 Figure 122 -Conversion gain (Gc) in dB in process and temperature variations: (a) in RF band (76-81GHz) for a 10MHz IF; (b) in IF band (10kHz-100MHz) for a 78.5GHz

Figure 123 -

 123 Figure 123 -Noise figure (NF) in dB in process and temperature variations: (a) in RF band (76-81GHz) for a 10MHz IF; (b) in IF band (10kHz-100MHz) for a 78.5GHz

Figure 124 -JammerTable 24 -

 12424 Figure 124 -1dB compression point (ICP1) in dBm in process and temperature variations: (a) in RF band (76-81GHz) for a 10MHz IF; (b) in RF band (76-81GHz) for a 20kHz band; (c) in IF band (10kHz-100MHz) for a 78.5GHz

Figure 125 -Figure 126 -Table 25 -

 12512625 Figure 125 -Maximal range regarding the Intermediate Frequency for various targets in the worst case

  1) • 𝑃 𝐷𝐶 • 𝐴𝑟𝑒𝑎 The post-layout simulations, for typical case, of the two developed downconverters are summarized in Table 26 and compared to the state-of-the-art. Two figures of merit are proposed to compare these circuits: FOMI only accounts for noise, gain, linearity and power consumption, and FOMII includes the silicon footprint. The two proposed circuits present an interesting trade-off between noise and compression point. The current mode RX-RFFE achieves the best FOMI. With a moderate constrain on linearity, conventional architectures achieve lower noise figure and a large conversion gain. Regarding FOMII, the proposed circuits compares favorably with the state-of-the-art except [54] which has a reduced silicon footprint.

Chapter 5 .

 5 Conclusion To allow for full autonomy vehicles -i.e. a SAE level of 5-many sensors must be embedded from camera to lidar through RADAR systems. In this panel, radars present many advantages: a simultaneous measurement of distance and velocity, a high range and a high immunity to outdoor conditions. This flexibility allows various use cases from short range radars which create a safety cocoon around the car, to long range radars which allow an adaptive cruise control. Since few years, radars are based on fast chirp FMCW facilitating an unambiguous measurement of distance and velocity. Improving radar performances leads to two major trends for the receivers. First, the radar RF bandwidth increases to few gigahertz to improve the resolution. Secondly, the demodulated IF signal bandwidth also enlarges from 1MHz up to 40MHz. The development of the automotive radar market leads to select technologies capable of good RF performances, a high level of integration and a low production cost. Due to these considerations, a 28nm RF-CMOS technology is selected for this work.

Figure 127 -Figure 128 -

 127128 Figure 127 -RX-RFFE in a complete radar module

To address the linearity

  specification, a single stage LNA is preferred. Various arrangements of differential noise canceling LNA are evaluated, leading to the proposed C4-LNA which performances are summarized in Table 28. This topology designed at 77GHz compares favorably with the state of the art, especially in terms of performance to current trade-off. Passive mixing techniques is further investigated, with a comparison between voltage and current mode downconverter. Figure 129 presents a complete RX-RFFE with a millimeter-Wave Front-End and a baseband analog chain, including three stages of high-pass filters and VGA. The designed voltage mode mixer achieves a low noise figure, below 14dB in the worst case, a high gain, above 15dB but it suffers from a poor linearity, ICP1 below -7dBm. On the other hand, the current-mode downconverter presents a better trade-off between noise and linearity. Due to the high pass filtering, the baseband analog remains in voltage-mode and requires a TransImpedance Amplifier just after the mixer stage. The introduction of such additional block degrades the noise performance, +1dB, compared to voltage-mode downconverter. However, the current mode operation allows for an improved linearity since the ICP1 exceeds -5dBm. These results should be confirmed by the measurements of the manufactured circuits.

Figure 129 -

 129 Figure 129 -Architecture of the proposed downconverter including a TIA if the mixer works in current-mode

Regarding

  these state-of-the art, the proposed circuits presents significant improvements regarding RF performances. Even if flicker noise reduction techniques have been implemented, the active mixing topologies cannot fulfill long range radar specifications due to their high level of flicker noise. Finally, passive mixing topologies present an interesting tradeoff but requires advanced design techniques to improve the trade-off between linearity and noise. In this work, noise cancelling techniques have been experienced in millimeter-Wave to improve the RF performances of low noise amplifier with significant benefits, and the currentmode mixing operation demonstrates adequate performances. The research and work done for this thesis can be pursued by: a) improving the noise cancelling techniques for the development of 77GHz downconverters either at circuit level (LNA…) or architecture level b) investigating I/Q architectures and maybe multi-phase mixing c) the exploration of new frequency bands for emerging ADAS applications.

To address this purpose different technologies are proposed. The most popular, reported in Table 1, are: lidar, radar and secure vehicle-to- environment communication

(V2X)

. Since

these technologies provide a unique and different benefit, they are not concurrent but complementary [6].

		Driver Support	Automated Driving
		Features		Features
	0	1	2	4	5
	From Human	Steering, acceleration, deceleration Monitoring Driving environment	To driving system
		Fallback performance

Table 1 -

 1 Comparison of the proposed sensors for automated driving

	Technology	Lidar	R adar C am era measurement Object classification, sign reading	Safe connection to X (Traffic Lights, cars…)

Table 2 -

 2 ETSI Automotive Radar standards from

Table 3 and further discussed in the next sub-sections.

Table 3 -

 3 Typical LRR, MRR and SRR specifications[START_REF] Hasch | Millimeter-Wave Technology for Automotive Radar Sensors in the 77 GHz Frequency Band[END_REF]

		R adar type	LR R	M R R	SR R
		R Range (m)	10 to 250m	1 to 100m	0.15 to 30 m
	ΔR Range resolution (m)	0.5m	0.5m	0.1m
	δR Range accuracy (m)	0.1m	0.1m	0.02m
	Δv Velocity resolution (m/s)	0.6m/s	0.6m/s	0.6m/s
	δv Velocity accuracy (m/s)	0.1m/s	0.1m/s	0.1m/s
	ϕmax 3dB beamwidth in azimuth (°)	15°	40°	80°
	δϕ Azimuth angular resolution (°)	0.1°	0.5°	1°
	θmax 3dB beamwidth in elevation (°)	5°	5°	10°
	a.	Short R ange R adar (SRR)

Table 4 details the link between distance measurement performances key indicators and the signal characteristics. In modern ADAS applications, radars are expected to measure distant target with a high resolution and a good accuracy. For instance, a modern radar must be able to identity pedestrians over a long range and possibly close to a larger vehicle.

	Equation 1 -Range (R) measurement for Linear FMCW radar
	𝑅 =	𝐶 2	∆𝑡	𝐹𝑜𝑟 𝐿𝑖𝑛𝑒𝑎𝑟 𝐹𝑀𝐶𝑊 ⇔	𝑅 =	𝐶 2	𝑇 𝑐ℎ𝑖𝑟𝑝 𝐵𝑊 𝑐ℎ𝑖𝑟𝑝	𝑓 𝑏𝑒𝑎𝑡
	The

Table 4 -

 4 Target distance measurement performances (SNR: Signal-Noise-Ratio)

	P erform ance param eter	Equation	C onditions
	Maximum range Range resolution Range accuracy	𝑅 𝑚𝑎𝑥 = ∆𝑅 = 𝐶 2 𝐵𝑊 𝑐ℎ𝑖𝑟𝑝 𝑇 𝑐ℎ𝑖𝑟𝑝 𝑐 2 𝐵𝑊 𝑐ℎ𝑖𝑟𝑝 𝑓 𝑏𝑒𝑎𝑡,𝑚𝑎𝑥 𝛿𝑅 = 𝑐 3.6 𝐵𝑊 𝑐ℎ𝑖𝑟𝑝 √2 𝑆𝑁𝑅	Depending on maximal fbeat,max measurement As ∆𝑓 = 2 ∆𝑅 𝐵𝑊 𝑐ℎ𝑖𝑟𝑝 𝑐 𝑇 𝑐ℎ𝑖𝑟𝑝 1 and ∆𝑓 ≈ 𝑇 𝑐ℎ𝑖𝑟𝑝 Fraction of range resolution depending of SNR

Two major trends exist to improve the performance of the measured distance. First, RF chirp bandwidth is increased to improve range resolution and accuracy. For instance, a 1GHz bandwidth achieve a resolution of 15cm, a 4GHz bandwidth improves it to 4cm. Modern fast chirp modulation leads to reduce chirp time thus increases the maximum measurable frequency difference (fbeat). In consequence, RF wideband receivers with higher baseband Intermediate Frequency (IF) are needed.

Table 5 -

 5 Velocity measurement performances

	P erform ance		Equation	C onditions
	Maximum velocity	𝑣 𝑚𝑎𝑥 =	𝜆 4 𝑇 𝑐ℎ𝑖𝑟𝑝	Unambiguous measurement for |∆𝜙| < 𝜋
	Velocity resolution	∆𝑣 =	𝜆 2 𝑇 𝑐ℎ𝑖𝑟𝑝 𝑁	Unambiguous measurement for ∆𝜔 >	2𝜋 𝑁
	Velocity accuracy	𝛿𝑣 =	3.6 𝑇	𝜆

𝑐ℎ𝑖𝑟𝑝 𝑁 √𝑆𝑁𝑅 Fraction of velocity resolution depending in SNR

Table 3 ) and gain formula from [18]. On the other hand, the receiver needs a minimum input power called sensitivity noted PRX,min, to properly process the signal. The Equation 4 links the maximum achievable range to these parameters.

 3

	Equation 4 -Radar maximum achievable range
	𝑅 𝑚𝑎𝑥 (𝑚) = (	(4𝜋) 3 𝑃 𝑅𝑋,𝑚𝑖𝑛 𝑃 𝑇𝑋 𝐺 𝑇𝑋 𝐺 𝑅𝑋 𝜆 2 𝜎	) 0.25

Table 6 -

 6 Common radar cross section for specific targets[START_REF] Kärnfelt | 77 GHz ACC Radar Simulation Platform[END_REF]

Target Type σ (dB sm ) Pedestrian -10 Bike 7 Car Maximum {10 log (Range (m)) + 5; 20} Truck Maximum {20 log (Range (m)) + 5; 45} The MCU performs an FFT of the data with a resolution (RBW) as illustrated in Figure 18. To execute this signal processing, a minimum Signal-on-Noise Ratio (SNRmin) is required at ADC input. The SNRmin is linked to PRX,min by the Equation 5 which defines the receiver

		G TX (dB)
	Radar Cross Section σ (dBsm)	P TX (dBm)	TX	LO
	R max (m)
	Target	G RX (dB) P RX min	RF Demodulator	SNR min (dB)	ADC	Data	(RWB) MCU
		(dBm)	NF DEMOD
	Bumper

sensitivity from noise floor at -174dBm accounting the noise contribution of the RF demodulator represented by its noise figure NFDEMOD.

	Equation 5 -Sensitivity definition for a receiver
	𝑃 𝑅𝑋,𝑚𝑖𝑛 (𝑑𝐵𝑚) = -174𝑑𝐵𝑚 + 𝑁𝐹 𝐷𝐸𝑀𝑂𝐷 (𝑑𝐵) + 10 log(𝑅𝐵𝑊) + 𝑆𝑁𝑅 𝑚𝑖𝑛 (𝑑𝐵)

Considering these two equations, a top-level specification on the maximum range define the maximal NF of the RF demodulator as proposed in Figure 19. In short range use case, a 20 dB NFDEMOD is expected to target at 80m. In long range use case, the maximal measurable distance should be higher than 200 meters, such feature leads a NFDEMOD at 17dB. These noise specifications may appear as relaxed, but it should consider the noise distribution overall the all demodulator chain and the NF de-sensitization. These two elements are discussed is the following sections.

Table 7 sums up some of the important specifications of a radar RX-RFFE.

		RF bandwidth from 76 to 81 GHz
	Power			f beat ≈ 400 kHz to 40 MHz ≈ 10 kHz
			f LO f bump f RF1 f RF2	Frequency
	IF bump	IF 1	IF 2	IF bandwidth	Fundamental Frequencies
					2 nd Intermod products & Harm. distorsions
	Amplitude				3 st Intermod products & Harm. distorsions
					IF bandwidth

Table 7 -

 7 Specifications of receiver RF front-

dnd Specifications M in Typ M ax RF bandwidth (GHz) 76 81 IF bandwidth (MHz) 1 / 40 Noise figure (dB) For LRR 15 For SRR 17 ICP1 (dBm) -5 Passband ICP1 (dBm) -20 IIP3 (dBm) -26 LO-to-RF Isolation (dB) -35 Objectives M in Typ M ax Conversion Gain (dB) 10 Power consumption (mW) 20

Table 8 , they presented good performances in terms of gain, cut-off frequency or temperature behavior which are critical for 77GHz radar applications. Nevertheless, III-V technologies are expensive and do not offer a large scale of integration for digital processing. For these reasons, III-V technologies were reserved for luxurious car with an implementation based on discrete components, whereas semiconductor industry is moving to Silicon technologies for mass market products. The two historic market leaders (NXP and Infineon) have selected BiCMOS technology on Silicium-Germanium substrate for their products thanks to their good trade-off between integration level and RF performances. As shown in Figure 31, BiCMOS technologies can achieve high transition frequencies with good performances. Nevertheless, modern ADAS do not only require good RF performances but also a high level of integration and a low-cost production. On one hand, as discussed in 1.c, digital functions are included in radar chip and semiconductor manufacturers are considering the integration of the MCU and radar transceiver.

 8

Table 8 -

 8 Comparison of integrated technologies

	Technology	C M OS	B iC M OS (SiG e)	III-V (G

aA s) Transition frequency Medium High Very High B reakdow n voltage Low Medium High P ow er gain Medium High Very High Tem perature behavior Poor Good Good Integration level Very High Medium Very Low C ost Low Medium High These trends push the semiconductor industry is moving to consider CMOS technologies for the implement of the next generations of radar modules. As illustrated in Figure 31, recent advanced RF CMOS technologies are now compatible with 77GHz radar constraints, especially regarding the transition frequency which exceeds 300GHz for technology nodes below 50 nm. The circuits proposed in this manuscript are developed in a 28nm CMOS process. The technology back end is illustrated in Figure 32. The three tick upper metal layers (AP, M8, M7) are used for RF routing. The six thin lower metal layers (M1 to M6) are basically used for digital blocks.

Table 10 -

 10 State

	P aram eter	[28]	[30]	[31]	[32]	[33]	[34]	This w ork
	CMOS tech (nm)	65	28	28	65	65	65	28
	Architecture	Active IQ-mixer	LNTA IQ-mixer +BB	LNTA IQ-mixer +BB	Active IQ-mixer +BB	LNTA I-mixer	Active I-mixer	Active I-mixer
	VDD (V)	1.2	0.9	1.8	0.6	0.6	1.2	0.9	1.2
	Bandwidth (GHz)	0.8-7	0.6	0.8	1.3	0.8	1-10	0.8-4	0.8-6
	Conv. gain (dB)	18	53.3 +	43.4 +	57 +	22.8	12.8	10	15
	NF (dB)	5.5*	6.5*	7.5*	8.5*	8	7.5*	7	6.2
	IIP3 (dBm)	-3	-8	6	-6	-21	-5	0.5	-1
	Power (mW)	16	0.35	4.3	1.7	0.064	5	2	4
	Area (mm²)	0.01**	0.1	0.22	0.24	0.22	0.01	0.04

of the art of low power downconverters (* NFDSB, + including baseband, ** including pads)

  Equation 33 -Flicker noise current due tail capacitance and vn gate noise voltage

	𝑖 𝑛,𝑜 =	2 𝑇 𝐿𝑂	∫ 𝑇 𝐿𝑂 /2 0	𝑖 𝐶 𝑝 (𝑡)𝑑𝑡 =	2 𝑇 𝐿𝑂	∫ 𝑇 𝐿𝑂 /2 0	𝐶 𝑝	𝑑 𝑉 𝑠 (𝑡) 𝑑𝑡	𝑑𝑡	⇔	𝑖 𝑜,𝑛 (𝑓) =	𝐶 𝑝 𝜔 𝐿𝑂 𝜋	• 𝑉 𝑛 (𝑓)

Time (ps) Switches behavior LO period τ rf τ p Switches behavior in post simulations Square wave model of switches behavior

	Equation 39 -Expression of overall gain
	𝐺 𝑣 = (𝐺 𝑚𝐶𝐺 + 𝐺 𝑚𝐶𝑆 ) • 𝑅 𝐿 • {	2𝜋 𝜏 𝑝 𝑇 𝐿𝑂	𝑠𝑖𝑛𝑐 ( 𝜋 𝜏 𝑝 𝑇 𝐿𝑂	) • 𝑠𝑖𝑛𝑐 (	𝜋 𝜏 𝑟𝑓 𝑇 𝐿𝑂	)}

Table 11 -

 11 Noise summary of the proposed noise cancelling mixer at high intermediate frequency

	N oise source	N oise type	V alue (V ²/H z)	%
	Switches	Thermal Resistive	16.4•10 -18 4.9•10 -18	16.4 5
	Common Source transistor	Thermal Resistive	13.2•10 -18 5.6•10 -18	12.4 5.7
	Common-gate transistor	Thermal Resistive	5.1•10 -18 1.4•10 -18	5.2 1.4
	Load	Resistive	13.2•10 -18	13.4
	RF Port	Resistive	10.4•10 -18	10.6

Table 12 -

 12 State of the art of millimeter-wave mixers (*: Including pads; **: Including pads and baseband) for

	typical measurements

Regarding the Table 9 of front-end specifications, the present active downconverter is not suitable for long range applications. The Figure 77 shows that the noise figure remains

Table 13 -

 13 Measurement results of noise cancelling mixer for RF and millimeter-Wave applications

	P aram eters	R F M ixer	m m W M ixer
	Frequency (G H z)	0.4-6	76-81
	C onversion gain (dB )	15	12.5
	N F (dB) at 1M H z N F (dB) at 10M H z	6.2	20 13
	Linearity (dB m )	-1 (IIP3)	-2.5 (ICP1)
	P ow er (m W )

Table 14 -

 14 Overview of the main FoM of the considered blocks in the demodulation chain

	P aram eters	M ixer	V G A 0 (low noise) & H P F0	V G A 1 & H P F1	V G A 2 & H P F2
	Gain (dB)	-4	10	22	10
	NF (dB)	8	/	/	/
	Noise (nV/√Hz)	/	3	10	50
	Linearity	@20kHz @10MHz	+5 dBm	0.57 VRMS 0.14 VRMS	0.57 VRMS 0.45 VRMS	/ /
		1. LN A

specification: N oise Figure The main purpose of a LNA is to increase the level of the desired RF signal with a minimum degradation on its Signal-to-Noise Ratio. Considering the noise distribution on the demodulation chain expressed by Friis formula (Equation 42), the RF characteristics of the LNA are predominant for the overall noise performance. A large gain (GLNA) is supposed to hide the noise contribution of the following stage, and a reduced noise figure (FLNA) should lower the overall RX-RFFE noise figure.

Equation 42 -Friis formula on noise

Table 15 sums up the LNA specifications in terms of noise figure, linearity and gain accounting for the RX-RFFE characteristics reported in Table 9 to address long range radar (LRR) applications.

 15

Table 15 -

 15 Targeted performances for suitable LNA for radar applications

	Objectives	M in.	Typ.	M ax.
	Noise figure (dB) for LRR			7
	ICP1 (dBm) -For bumper management	-5
	IIP3 (dBm) -For signal intermodulation		-26
	Voltage Gain (dB)		10
	Power consumption (mW)		10

Table 16 presents an overview of the State of art of millimeter-Wave LNAs. These circuits usually feature several stages with conventional architectures such as common-source or common-gate topologies. Depending of the context, various trade-offs can be targeted. On one hand, some proposed LNAs [47] [48] optimize the NF of the first stage, then cascading a large gain stage limiting the overall linearity. One the other hand, some circuits [49] [50] achieve a high value of 1dB compression point by cascading 3 to 4 low gain stages thus degrading the NF performance. The nominal value of such NF, typically between 5.7 to 6.6 dB, does exhibit enough margin with our NFLNA specification, 7dB, to withstand PVT variations of advanced CMOS nodes. The 3-stage LNA proposed in [51] achieves good RF performances regarding gain, noise and linearity but it requires a very high-power consumption due to the number of cascaded stages. The 1-stage implementation [51] achieves a low noise figure, a high linearity with a limited power consumption, but a too low gain to ensure a RX-RFFE NF of 15dB.

 16

Table 16 -

 16 State of the art of 80GHz LNA for automotive radar applications (* Power gain)

	P aram eters	[47]	[48]	[49]	[50]	[51]	[51]
	Conditions	25	25	25	25	25	25
	Results	Meas.	Meas.	Meas.	Meas.	Meas.	Meas.
	Number of stages	2	2	3	4	3	1
	Tech. (nm)	CMOS 28nm	CMOS 28nm	CMOS 45nm	CMOS 90nm	CMOS 65nm	CMOS 65nm
	VDD (V)	0.9	2	1.4	1.2	1	1
	Frequency (GHz)	80	60	95	80	80	80
	Bandwidth (GHz)	12	15	23	23	14	14
	Gain (dB)	16.5*	13.8*	10.7*	7.6*	7.2*	2.1*
	NF (dB)	5.2	4	6	6.6	5.7	4.5
	ICP1 (dBm)	-15.5	-12.5	-3.2	-7	-3.75	0
	Power (mW)	26	24	52	11.3	70	22
	Area (mm²)	0.14	0.38	0.32	0.59	0.99	0.

33 The various CMOS low noise amplifiers proposed in Table 16 do not address the specifications of a modern 77GHz radars presented in Table 9. Technically the cascade of several stages is not a solution as it is difficult to achieve simultaneously a low noise figure and a high 1dB compression point at a moderate power consumption. On the other hand, a single stage LNA based on a basic common source, or common gate, configuration does not achieve enough gain. New LNA architectures are required to address our purpose.

For the auxiliary path, the expression of common-source amplifier input impedance is expressed in Equation 48.

	Equation 47 -Common-gate impedance
	𝑍 𝑖𝑛,𝐶𝐺 =	𝑍 𝑑𝑠1 + 𝑍 𝑑𝑔1 1 + 𝑔 𝑚,𝐶𝐺 𝑍 𝑑𝑠1	=	𝑔 𝑑𝑠1 + 𝑗𝜔(𝐶 𝑑𝑠1 + 𝐶 𝑑𝑔1 ) + 𝑌 𝐶𝐺 (𝑔 𝑚,𝐶𝐺 + 𝑔 𝑑𝑠1 + 𝑗𝜔𝐶 𝑑𝑠1 ) • ( 𝑗𝜔𝐶 𝑑𝑔1 + 𝑌 𝐶𝐺 )
	Equation 48 -Common-source impedance
	𝑍 𝑖𝑛,𝐶𝑆 =	𝑍 𝑑𝑠2 + 𝑍 𝑑𝑔2 1 + 𝑔 𝑚,𝐶𝑆 𝑍 𝑑𝑠2	=	𝑔 𝑑𝑠2 + 𝑗𝜔(𝐶 𝑑𝑠2 + 𝐶 𝑑𝑔2 ) + 𝑌 𝐶𝑆 (𝑔 𝑚,𝐶𝑆 + 𝑔 𝑑𝑠2 + 𝑌 𝐶𝑆 + 𝑗𝜔𝐶 𝑑𝑠2 ) • 𝑗𝜔𝐶 𝑑𝑔2

  Equation 49 -Overall input impedance of a common-gate-based noise-cancelling amplifier

	𝑍 𝑖𝑛 =	1 + 𝑔 𝑚2 𝑍 𝑑𝑠2 𝑍 𝑔𝑑2 + 𝑍 𝑑𝑠2	+	1 1 + 𝑔 𝑚1 𝑍 𝑑𝑠1 𝑍 𝑑𝑠1 + 𝑍 𝑑𝑔1 + 𝑗(𝜔(𝐶 𝑔𝑠1 + 𝐶 𝑔𝑠2 ) -	1 𝜔𝐿 1	)

Table 16 , this 1-stage LNA achieves a high linearity with an ICP1 of -5dBm at a reduced power consumption of 4.6mW. The typical noise figure is closed to 6dB, degraded to 9dB in worst case of PVT. According to Chapter 2.I.1, the noise-cancelling topology does not achieve lower NF than conventional common-source architectures but performs a good trade-off between RF performances and a wideband input matching.

 16

Table 17 -

 17 Performances

of the proposed LNA and a downconverter using this LNA with a voltage-mode mixer including typical case at 25°C and worst case in PVT P

aram eters LNA (Typical) LNA (Worst) RX (Typical) RX (Worst) Gain (dB) 14 8.7 9.6 6.3 Noise figure (dB) 6 9.4 11.4 13.7 ICP1

Table 18 -

 18 Schematic simulation in typical and hot case for automotive radar applications

Table 19 -

 19 Schematic simulation in typical and hot case for automotive radar applications

	P aram eters	LNA (Typical)	LNA (worst case)
	Conversion Gain (dB)	8.7	6.5
	Noise figure (dB)	5.3	6.8
	ICP1 (dBm)	-5.2	-5.6
	Power consumption (mW) / VDD (V)	<4 / 0.6V	<4 / 0.6V

With the expression of the output current Equation 55, it is possible to determine the overall transconductance of the amplifier (Gm,LNA) proposed in Equation 56.

	Equation 56 -Expression of the LNA transconductance
	𝐺 𝑚,𝐿𝑁𝐴 =	𝑖 𝑜𝑢𝑡 𝑣 𝑆	=	2 𝑛 • 1 + 𝑅 𝑆 2𝑔 𝑚 + 𝑔 𝑑𝑠 1 + 𝑍 𝐿± /𝑟 𝑑𝑠 𝑛 2 • 2𝑔 𝑚 + 𝑔 𝑑𝑠 1 + 𝑍 𝐿± /𝑟 𝑑𝑠	⇒ 𝐺 𝑚,𝐿𝑁𝐴 ≈	4𝑔 𝑚 𝑛 (1 + 2𝑅 𝑆 𝑔 𝑚 𝑛 2 )	= 𝑛/𝑅 𝑆
	Equation 55 -Output current expression
	𝑖 𝑜𝑢𝑡 = 𝑖 𝐴 -𝑖 𝐵 + 𝑖 𝐶 -𝑖 𝐷 =	2𝑔 𝑚 + 𝑔 𝑑𝑠 1 + 𝑔 𝑑𝑠 • 𝑍 𝐿	• (𝑉 𝐵 -𝑉 𝐴 + 𝑉 𝐷 -𝑉 𝐶 )
							with 𝑉 𝐴 -𝑉 𝐵 = 𝑉 𝐶 -𝑉 𝐷 =	𝑅 𝑆 2𝑛 2 • 𝑖 𝑜𝑢𝑡 +	𝑣 𝑆 𝑛

If the amplifier is exploited in a transconductance mode we can assume: rds >> ZL± and gm >> gds. Hence the transconductance is simplified to Equation 56 under matching conditions, defined in Equation 54. Interestingly the transimpedance gain (Gm,LNA) becomes proportional to the transformer ratio

  n.

	Equation 57 -Simplified transconductor gain under input matching conditions
	𝐺 𝑚,𝐿𝑁𝐴 ≈	4𝑔 𝑚 𝑛 (1 + 2𝑅 𝑆 𝑔 𝑚 𝑛 2 )	= 𝑛/𝑅 𝑆

Table 20 -

 20 Performances of the proposed LNA and a downconverter using this LNA with a voltage-mode mixer including typical case at 25°C and worst case in PVT

	P aram eters	LNA (Typical)	LNA (Worst)	RX (Typical)	RX (Worst)
	Conversion Gain (dB)	10.8	8.2	11.5	10
	Noise figure (dB)	4.2	7.2	9.4	12
	ICP1 (dBm)	-2.5	-3	-9	-9
	Power consumption (mW) / VDD (V)	4.2 / 1.5V	< 5 / 1.5V	< 5 / 1.5V	< 5 / 1.5V

Table 21 -

 21 State of the art of 80GHz LNA for automotive radar applications compared to post-layout simulations of the designed LNA in typical case (* Power gain; + Voltage gain)

	P aram eters	[48]	[49]	[50]	[51]	N C LN A	C 4 LN A
	Tech. (nm)	CMOS 28nm	CMOS 45nm	CMOS 90nm	CMOS 65nm	CMOS 28nm	CMOS 28nm
	Number of stages	2	3	4	1	1	1
	Bandwidth (GHz)	57-72	87-110	72-95	72-86	73-90	71-90
	Gain (dB)	13.8*	10.7*	7.6*	2.1*	14 +	10.8 +
	NF (dB)	4	6	6.6	4.5	6	4.2
	ICP1 (dBm)	-12.5	-3.2	-7	0	-5	-3
	Power (mW) / VDD	24	52	11.3	22	5	4.6
	(V)	/ 0.9V	/ 2V	/ 1.2V	/ 1.2V	/ 0.6V	/ 1.5V
	Area (mm²)	0.38	0.32	0.59	0.33*	0.01	0.01
	FOMI	0.008	0.011	0.012	0.032	0.053	0.12
	FOMII	0.02	0.033	0.02	0.096	5.3	10.3

each harmonic of LO. Considering a square waveform, the baseband current only contains odd harmonics. The baseband voltage vbb is then derived in Equation 66.

		s LO,2 (t)
	Z bb (ω) Equation 66 -Baseband voltage expression for differential mixer in time and frequency domains i bb+ (t) i RF+ (t) R SW ON state (t) OFF state (f) S LO,2 S LO
	v RF (t) (a)	v bb (t) Z bb (ω) 𝑣 𝑏𝑏 (𝑡) = 𝑖 𝑏𝑏 (𝑡) * 𝑧 𝑏𝑏 (𝑡) = ⇔ 𝑉 𝑏𝑏 (𝜔) = ∑ 𝑎 𝑛 • 𝐼 𝑅𝐹 (𝜔 -𝑛𝜔 𝐿𝑂 ) s LO,1 (t) s LO,1 (t) s LO,2 (t) Time f LO 3f LO 5f LO 7f LO T LO /2 +∞ i bb-(t) i RF-(t) Time s LO,1 (t) f LO 3f LO 5f LO 7f LO • 𝑍 𝑏𝑏 (𝜔) ON state OFF state R SW T LO /2 S LO (f) (b) (c) 𝑛=-∞ 𝑆𝑞𝑢𝑎𝑟𝑒 𝐿𝑂 ⇒ 𝑉 𝑏𝑏 (𝜔 𝐼𝐹 ) = ∑ +∞ 1 𝑛=-∞ 2𝑛 + 1 • 𝐼 𝑅𝐹 (𝜔 𝐼𝐹 -(2𝑛 + 1)𝜔 𝐿𝑂 ) • 𝑍 𝑏𝑏 (𝜔 𝐼𝐹 )	Frequency Frequency
		(2) Input R F voltage
	Equation 67 -Input RF voltage expression in time and frequency domain
		𝑣 𝑅𝐹 (𝑡) = 2𝑅 𝑆𝑊 • 𝑖 𝑅𝐹 (𝑡) + (𝑠 𝐿𝑂,2 (𝑡) -𝑠 𝐿𝑂,1 (𝑡)) • 𝑣 𝑏𝑏 (𝑡) 𝑉 𝑅𝐹 (𝜔) = 2𝑅 𝑆𝑊 • 𝐼 𝑅𝐹 (𝜔) + 1 2𝜋 (𝑆 𝐿𝑂,2 (𝜔) -𝑆 𝐿𝑂,1 (𝜔)) * 𝑉 𝑏𝑏 (𝜔)

To understand the difference between current and voltage mode operation, the input RF voltage (vRF) is derived in Equation 67. If we consider the input signal only includes components at the RF frequency, the input RF voltage is simplified to Equation 68.

	Equation 68 -Input RF voltage expression frequency domain
	+∞	+∞
	𝑉 𝑅𝐹 (𝜔) = 2𝑅 𝑆𝑊 • 𝐼 𝑅𝐹 (𝜔) + ∑ ∑ 𝑎 𝑚 • 𝑎 𝑛 • 𝐼 𝑅𝐹 (𝜔 -(𝑚 + 𝑛)𝜔 𝐿𝑂 ) • 𝑍 𝑏𝑏 (𝜔 -𝑚𝜔 𝐿𝑂 )
	𝑚=-∞	𝑛=-∞	+∞
	𝑉 𝑅𝐹 (𝜔 𝑅𝐹 ) = 𝐼 𝑅𝐹 (𝜔 𝑅𝐹 ) • (2𝑅 𝑆𝑊 • + ∑ |𝑎 𝑚 | 2 • 𝑍 𝑏𝑏 (𝜔 𝑅𝐹 -𝑚𝜔 𝐿𝑂 )
			𝑚=-∞

) (

3) Input R F im pedance

  Equation 69 -Input RF impedance expression and approximation in frequency domain𝑧 𝑅𝐹 (𝜔 𝑅𝐹 ) = 2𝑅 𝑆𝑊 + ∑ |𝑎 𝑚 | 2 • 𝑍 𝑏𝑏 (𝜔 𝑅𝐹 -𝑚𝜔 𝐿𝑂 )

	+∞
	𝑛=-∞ 𝑧 𝑅𝐹 (𝜔 𝑅𝐹 ) ≈ 2𝑅 𝑆𝑊 + |𝑎 0 | 2 • 𝑍 𝐵𝐵 (𝜔 𝑅𝐹 ) + |𝑎 1 | 2 • (𝑍 𝑏𝑏 (𝜔 𝑅𝐹 -𝜔 𝐿𝑂 ) + 𝑍 𝑏𝑏 (𝜔 𝑅𝐹 + 𝜔 𝐿𝑂 ))
	Combining Equation 67

and 68, the input RF impedance ZRF is derived in Equation 69. The baseband impedance ZBB is translated at RF frequency by virtue of LO switching.

Table 22 -

 22 Performances of the baseband analog blocks HPF-VGA

	P aram eters	H P F-V G A 0	H P F-V G A 1	H P F-V G A 2
	Conv. gain (dB)	10	22	10
	Noise (V/√Hz)	3	10	50
	ICP1	@20kHz	0.57	0.57	/
	(Vrms)	@10MHz	0.14	0.45	/

To properly process the signal, a dynamic range of 20 dB is needed with specific constraints on linearity. Due to this trade-off, the two cascaded VGA usually presents a high noise contribution as reported in Table 22. Thereby, a third low noise HPF-VGA (HPF-VGA0) is added before HPF-VGA1 and HPF-VGA2. As described in Table 22, it is designed to minimize the impact on analog chain noise with a reduced noise contribution of 3nV/√Hz and a moderate gain of 10dB.

Figure 107 -Schematic of a Variable gain amplifier including high-pass filter

The development of the LNA is based on the C4 topology presented in the Chapter 3.III. It is designed to simultaneously achieve a sufficient linearity, ICP1, to handle bumper reflection, to perform a low noise figure and a sufficient gain to attenuate the noise contribution of the following stages. The gain of the LNA would still be limited to avoid the mixer saturation. To drive the mixer a buffer stage is added on the LO path to hard switch the transistors. This buffer is based on C4 topology designed to achieve more than 8dB voltage gain, and a low output impedance. This LO driver is supplied by a 1.5 V and consumes 13mW, considering a 0dBm input LO power.

  Input RF voltage and current in the mixer considering an output load for LNA 𝐼 𝑅𝐹,𝑉𝑜𝑙𝑡-𝑀𝑖𝑥 (𝜔 𝑅𝐹 ) = 𝐼 𝑜𝑢𝑡,𝐿𝑁𝐴 (𝜔 𝑅𝐹 ) 1 + |𝑎 1 | 2 • (𝑍 𝑏𝑏 (𝜔 𝑅𝐹 -𝜔 𝐿𝑂 ) + 𝑍 𝑏𝑏 (𝜔 𝑅𝐹 + 𝜔 𝐿𝑂 )) 𝑍 𝑜𝑢𝑡,𝐿𝑁𝐴 (𝜔 𝑅𝐹 ) 𝑉 𝑅𝐹,𝑉𝑜𝑙𝑡-𝑀𝑖𝑥 (𝜔 𝑅𝐹 ) = 𝐼 𝑜𝑢𝑡,𝐿𝑁𝐴 (𝜔 𝑅𝐹 ) • (𝑍 𝑜𝑢𝑡,𝐿𝑁𝐴 (𝜔 𝑅𝐹 ) // 𝑍 𝑅𝐹,𝑉𝑜𝑙𝑡-𝑀𝑖𝑥 (𝜔 𝑅𝐹 ))

	Equation 71 -
				s LO,2
		v RF,Volt-Mix+	R SW	i bb+	Z bb
	I out,LNA+		i RF,Volt-Mix+
	Z out,LNA	C p,mix	Z RF,Volt-Mix	s LO,1 s LO,1	v bb
	I out,LNA-
			i RF,Volt-Mix-
		v RF,Volt-Mix -	R SW	i bb-	Z bb
				s LO,2

Table 23 -

 23 Simulations of Voltage-mode RX in typical, best and worst cases

	P aram eters	Specification	RX (Typical)	RX (Hot/Slow)	RX (Cold/Fast)
	VDD (V)	/	1.5	1.5	1.5
	Bandwidth (GHz)	76-81	75-89	77-90	73-82
	Conv. Gain (dB)	@IF=10MHz	> 15	20.3	15.5	22
	Noise figure (dB)	@IF=1MHz @IF=10MHz	/ < 15	16 10.1	21 15	14 8.1
	ICP1 (dBm)	@IF=20kHz @IF=10MHz	> -20 > -5	-9.1 -15.2	-7.1 -13.6	-9.8 -17.6
	Area (mm 2 )	/	0.09	0.09	0.09
	Power consumption (mW)	/	18	18	18

  2

Table 26 -

 26 State of the art of millimeter-Wave RX Front-End compared to post-layout simulations of the designed RX Front-End for automotive radar applications in typical case (* including pads)

P aram eters [43] [55] [56] [54] This w ork (V oltage) This w ork (C urrent)

	Results	Meas.	Meas.	Meas.	Meas.	Post-layout sim.	Post-layout sim.
	Tech. (nm)	CMOS 40 nm	CMOS 65nm	CMOS 40nm	CMOS 45nm	CMOS 28nm	CMOS 28nm
	VDD (V)	1.1	1	1.1	1.1	1.5	1.5
	Frequency (GHz)	77	77	60	60	77	77
	Bandwidth (GHz)	/	5	10	9	14	19
	Gain (dB)	30.8	11	30	26	20.3	20
	NF (dB)	@1MHz @10MHz	/ 9	/ 8	/ 5.5	/ 6	16 10.1	16.5 13.1
	ICP1 (dBm)	@20kHz @10MHz	-22.3	-15	-31	-22.5	-9.1 -15	-5 -15.5
	Power (mW)	143	22	35	21	18	23
	Area (mm²)	0.8*	0.231	0.7*	0.023	0.09	0.12
	FOMI	0.0002	0.001	0.0003	0.002	0.008	0.012
	FOMII	0.0003	0.004	0.0004	0.078	0.085	0.01

Table 28 -

 28 Summary of the performances of the RF noise cancelling demonstrator and the millimeter-Wave low noise amplifier designed in this thesis

						RF IN+	RF IN-
						Integrated 1:2
	vdd			vdd		transformer
	Z L			Z L			vdd
	IF+				IF-
	LO +		LO -		LO +	RF OUT+		RF OUT-
					Current reuse
	V b	Noise cancelling transconductance cell	Gm cell
	RF					BIAS
							DC current
	(a)		V b		(b)		source
			R F noise cancelling	m m W noise
	P aram eters		m ixer	cancelling LN A	m m W C 4 LN A
	Operating temperature	25	85	25	150	25	150
	Results		Meas.	Post-layout sim.	Post-layout sim.	Post-layout sim.
	Mixer topology		Passive		Passive	Passive
	Tech. (nm)		CMOS 28nm	CMOS 28nm	CMOS 28nm
	VDD (V)			1.2		0.6	1.5
	Frequency (GHz)			2.4		77	77
	Bandwidth (GHz)		5	5	16	16	15	15
	Gain (dB)		15	14	14	8.7	10.8	8.2
	NF (dB)		6.2	14	6	9.4	4.2	7.2
	ICP1 (dBm)		-1	-1	-5	-4.5	-2.5	-3
	Power (mW)		4	4	4.6	5	4.2	5
	Area (mm²)		0.04	0.04	0.024	0.024	0.011	0.011
	FOMI		0.35	0.041	0.12	0.025	0.28	0.061
	FOMII		8.8	1.0	4.8	1.0	25	5.3

Flicker noise reduction Switching cell Active mixing is first considered for its interesting trade-off between gain, noise and linearity. The noise cancelling is first demonstrated in an RF mixer whose performances are presented in Table 28. To reduce the impact of a large flicker noise, specific techniques are implemented in the mm-Waves version. Presented in Table 29, the 77GHz active mixer performing noise cancelation achieves in typical case a high gain, 12.5dB, a high linearity (ICP1), -2.5dBm and a moderate noise figure of 13dB. Considering PVT variations for automotive, the minimal gain is closed to 11dB, the linearity is higher than -7dBm while noise figure is below 18dB. Even with flicker noise reduction techniques, the noise figure is degraded by 8dB between 10MHz and 1MHz. This active down-converter demonstrates adequate performances for short range RADAR applications, but not enough low NF and linearity for a long-range use case as illustrated in Table 29. To address this issue a combination of a passive mixer with a LNA is further reconsidered in the following work.

Table 29 -

 29 Summary of the performances of the millimeter-Wave downconverter designed in this thesis

	P aram eters	m m W voltage m ode dow nconverter w ith C 4 LN A	m m W current m ode dow nconverter w ith C 4 LN TA	m m W noise cancelling active m ixer
	Conditions	25	150	25	150	25	150
	Results	Post-layout sim.	Post-layout sim.	Meas.	Post-layout sim.
	Tech. (nm)	CMOS 28nm	CMOS 28nm	CMOS 28nm
	VDD (V)	1.5		1.5		1.5
	Frequency (GHz)	77		77		77
	Bandwidth (GHz)	14	13	19	18	30	30
	Gain (dB)	20.3	15.5	20	16	13	12
	NF (dB)	@1MHz @10MHz	16 10.1	21 15	16.5 11	20 15.5	20 13.5	22 16
	ICP1 (dBm)	@20kHz @10MHz	-9.1 -15.2	-7.1 -13.6	-4 -14	-3 -16	-2.5	-4
	Power (mW)	18	18	23	23	16	16
	Area (mm²)	0.09	0.09	0.12	0.12	0.04	0.04
	FOMI	0.008	0.003	0.015	0.004	0.007	0.003
	FOMII	0.085	0.031	0.015	0.033	0.18	0.064

Table 30 -

 30 State of the art of millimeter-Wave downconverters

	P aram eters	[42]	[45]	[55]	[43]	[56]	[54]
	Results	Meas.	Meas.	Meas.	Meas.	Meas.	Meas.
	Mixer topology	Active	Active	Passive	Passive	Passive	Passive
	Tech. (nm)	SiGe Bip.	CMOS 65nm	CMOS 65nm	CMOS 40nm	CMOS 40nm	CMOS 45nm
	VDD (V)	3.3	1	1	1.1	1.1	1.1
	Frequency (GHz)	77	77	77	77	60	60
	Bandwidth (GHz)	15	28	5	/	10	9
	Gain (dB)	21.5	9.5	11	30.8	30	26
	NF (dB)	@1MHz @10MHz	/ 10.8	/ 9.2	10 9	8	/ 5.5	/ 6
	ICP1 (dBm)	@20kHz @10MHz	-5	-3.8	-22.3	-15	-31	-22.5
	Power (mW)	70	15	143	143	35	21
	Area (mm²)	0.53*	0.27	0.231	0.8	0.7	0.023
	FOMI	0.005	0.011	0.001	0.0002	0.0003	0.002
	FOMII	0.009	0.042	0.004	0.0003	0.0004	0.08

The post-layout simulations with temperature and process variations, Figure 101, show that the amplifier achieves a 10dB conversion gain in a typical case. In comparison to other presented topologies, the central frequency in PVT presents a small variation below 5GHz while the 3dB bandwidth is around 20GHz for each case corresponding to a 25% ratio between the bandwidth and the central frequency. On the other hand, the variation of the voltage gain is limited to 2.5dB, between 8.3dB and 10.9dB. Illustrated in Figure 102, the 1dB compression point is typically around -2.5dBm with small variations of ±0.5dB. This high linearity is fully compatible with the specification for radar applications. 27. Besides this thesis studies various mixing approach to determine the best suited for radar applications. First, active mixers achieve good performance trade-off but suffer from high flicker noise. To address this issue, passive mixers are then studied, achieving better trade-off. To overcome this drawback, an 80GHz current mode mixer featuring noise and linearity improvement is developed. K eyw ords: radar, millimeter-Wave, low noise, amplifier, mixer, receiver